System and method for generating signal waveforms in a CDMA cellular telephone system

A SYSTEM AND A METHOD FOR GENERATING SIGNAL WAVE FORMS IN A CDMA CELL PHONE SYSTEM

The invention relates to cellular telephone systems. More particularly, the present invention relates to a new and improved system and method for communicating information in a mobile cellular telephone system or a satellite mobile communication system using wide-spectrum communications signals.

The use of code division multiple access modulation (CDMA) technology is one of many techniques that enable communication where a large number of system users are present. Other multiple access communication system techniques are also known, such as time division multiple access (TDMA), frequency division multiple access (FDMA), and AM modulation systems, such as amplitude-computed single-band (ACSSB). However, CDMA wide (t. spread spectrum modulation techniques have significant advantages over these other multi-use communication system modulation techniques. The use of CDMA technology in multi-use communication systems has been described in U.S. Pat. No. 4,901,307, issued Feb. 13, 1990, under the name SPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS. applicant for this application, and each publication incorporated herein by reference.

The patent just mentioned discloses a multi-use technology where a large number of mobile system users, each with a transceiver, communicate with satellite repeaters or ground base stations (referred to as cellular stations, or cellularly) using code division multiple access (CDMA) wideband signaling. When using CDMA communications, the frequency spectrum can be reused several times, allowing for increased system user capacity. The use of CDMA technology leads to

20095859 prh 04 -04- 2011 spectral efficiency that can be achieved with other multiple access technologies.

The satellite channel typically has a sway called the Rician sway. Accordingly, the received signal includes a direct component summed several times with the reflected component, which follows Rayleigh hover statistics. The power ratio of the straight and reflected component is typically in the order of 6 to 10 dB, depending on the antenna characteristics of the mobile unit and the environment of the mobile unit.

When comparing a satellite channel to a terrestrial channel, the latter experiences a signal fluctuation, which typically includes a Rayleigh flicker component without a direct component. Thus, the terrestrial channel represents a more severe sway environment than the satellite channel, where the Rician sway is the predominant sway.

The Rayleigh wobble features of your Earth channel gna aI are due to a signal reflected in many different aspects of the physical environment. As a result, the signal arrives almost simultaneously at the receiver of the mobile unit in various directions with different propagation delays. In the UHF frequency bands commonly used for cellular radio communications including cellular cellular telephone systems, significant phase differences may occur with signals propagating along different paths. The possibility of a devastating summation of these signals exists, which deepens the swaying as the opportunity arises.

Land canal shielding is very strongly dependent on the physical location of the moving unit. A small change in the location of the mobile unit alters the physical delays of all signal paths, which further causes a different phase of each path. Thus, the motion of the moving unit in the environment can cause a rather rapid fluctuation phenomenon. For example, 850 MHz cellular

20095859 prh 04 -04- 2011 In the radio frequency band, this sway can typically be one sway per second and the vehicle speed miles per hour. Fluctuations of this magnitude can be extremely destructive to terrestrial channel signals and result in poor quality connection. However, additional transmitter power can be used to solve the sway problem. However, such power increases affect both the communications device, increased power consumption, and increased system interference.

The CDMA modulation technique described in US 4,901,307 offers many advantages over the narrowband modulation technique used in communications systems using satellite or terrestrial repeaters. The terrestrial channel causes particular problems for any communication system, especially with respect to multipath signals. The use of CDMA technology allows to overcome specific problems of the terrestrial channel by mitigating the adverse effects of multipath, e.g., flutter, while taking advantage of its benefits.

In a CDMA cellular telephone system, the same frequency band is used for communication in all cells. The CDMA waveform characteristics that provide processing gain are also used to discriminate between signals in the same frequency band. Further, high-speed pseudo-noise (PN) modulation allows the separation of many different paths, provided that the difference in path paths exceeds the PN gain or one / bandwidth. If a 1 MHz PN return frequency is used in a CDMA system, then full bandwidth processing gain equal to the ratio of spread bandwidth to system data rate may be used against paths that differ by more than one microsecond in path delay desired. One microsecond path delay differential equals 1,000 feet

20095859 prh 04 -04- 2011 differential route trip. The urban environment typically provides differential path delays greater than one microsecond and delays of up to 10-20 microseconds have been observed in some areas.

In narrowband modulation systems, such as the analogue FM modulation used with conventional telephone systems, the existence of multiple paths results in severe multipath fading. However, in broadband CDMA modulation, different paths can be discriminated in the demodulation process. This discrimination greatly reduces the severity of the multipath flicker. Multipath fluttering is not completely eliminated using CDMA discrimination because of the occasional occurrence of paths with delayed differents less than the PN block duration of a particular system. Signals with path magnitudes of this magnitude cannot be discriminated in the demodulator.

Therefore, it is desirable for the system to provide diversity to further reduce the effects of turbulence. Diversity is one approach to mitigate the adverse effects of fluctuations. There are three main types of diversity:

time, frequency, and space.

The time characteristic can best be achieved by using repetition, time interleaving and error detection and coding, which is a form of repetition. The present invention uses each of these techniques as a form of time characteristic.

CDMA, being a broadband signal by nature, provides a form of frequency diversity by spreading the signal energy over a wide bandwidth. Therefore, the frequency-selective oscillation affects only a small portion of the CDMA signal bandwidth.

Space or path diversity is achieved by providing multiple signal paths via simultaneous links from a mobile station to two or more

20095859 prh 04 -04- 2011 via cell site. Further, path multiplicity or multipath can be achieved by utilizing the multipath environment through wide-spectrum processing by allowing the signal to arrive at various propagation delays to be received and processed separately. Examples of multipath are described in a co-pending U.S. patent application entitled SOFT HANDOFF IN A CDMA CELLULAR TELEPHONE SYSTEM, No. 07 / 433,030, filed November 7, 1989, now patent US-5,101,501, filed March 31, 1992, and in the co-pending US patent, DIV. IN A CDMA CELLULAR TELEPHONE SYSTEM, NO. No. 07 / 432,552, also filed November 7, 1989, now U.S. Patent No. 5,109,390, issued April 28, 1992, each having the same holder as the present application.

The unfavorable effects of the wobble can be further controlled to some extent in the CDMA system by controlling the transmitter power. A system for controlling the power of a cell site and a mobile station is described in a co-pending U.S. patent application entitled METHOD AND APPARATUS FOR CONTROLLING TRANSMISSION POWER IN A CDMA CELLULAR MOBILE TELEPHONE SYSTEM, No. 07 / 433,031, filed November 7, 1989; and the holder of which is also the applicant for this application.

The use of coherent modulation and demodulation in bidirectional cell-to-satellite communication was considered by the CDMA technique disclosed in US 4,901,301. Accordingly, the use of a pilot carrier signal as a coherent phase reference in a satellite-to-mobile link and a cell-to-mobile link has been described. However, in the terrestrial cellular environment, the severity of multipath fluctuation, and consequently channel phase spreading, prevents the use of coherent demodulation technology over

20095859 prh 04 -04- 2011 from the media-to-cell link. The present invention provides means for overcoming the disadvantages of multicast in a mobile-to-cell link using non-coherent modulation and demodulation techniques.

The use of relatively long PN sequences, where each channel is assigned a different PN sequence, was further contemplated by the CDMA technique disclosed in US 4,901,301. The cross-correlation between the different PN sequences and the autocorrelation of the PN sequence for all non-zero time offsets have an average value of zero, which allows for discrimination of different user signals upon reception.

However, such PN signals are not orthogonal. Although the cross-correlations average to zero, in a short period of time, such as the information bit time, the cross-correlation follows the binomial distribution. In this way, the signals interfere with each other much in the same way as they would with wide bandwidth Gaussian noise at the same power spectrum density. Thus, other users' signals, or interference noise between them, will eventually limit the available capacity.

The existence of multipath may result in route spreading in a PN-CDMA broadband system. If two or more paths are available with a path delay difference greater than one microsecond, then two or more PN receivers can be used separately to receive these signals. Since these signals typically indicate independence in multipath wobbling, i.e., they are not usually wobbled together, the outputs of the two receivers can be multipathed. Therefore, a loss in performance occurs only when both receivers experience a flutter at the same time. Therefore, one feature of the present invention is to provide two or more PN receivers with a multipath combiner. Multi

20095859 prh 04-04-2011 In order to exploit the existence of road signals, to overcome the sway, it is necessary to use a waveform that permits multipath combining operations.

It is therefore an object of the invention to provide the generation of PN sequences which are orthogonal to the PN sequences to reduce mutual interference, thereby allowing for greater user capacity and supporting path diversity to overcome the flutter.

The implementation of wide-spectrum communications technology, particularly CDMA technology, in a cellular communications environment provides features that greatly enhance system reliability and capacity over other communications systems. CDMA technology, as previously mentioned, allows for the elimination of problems such as flutter and interference. Further, CDMA technology favors greater frequency reuse, thereby allowing for a significant increase in system users.

The present invention is a novel and improved method and system for generating PN sequences that provide orthogonality between users so that mutual interference is reduced, allowing for higher throughput and better link performance. For orthogonal PN codes, the cross-correlation is zero over a predetermined time period, resulting in no interference, interference, between the orthogonal codes, provided that the code time frames are time-aligned with each other.

In the exemplary embodiment, the signals are communicated between the cell-site and the mobile stations using direct-sequence wide-band communication signals. In the cell-to-mobile link, pilot, synchronization, paging, and voice channels are specified. Information communicated on cell-to-mobile link channels is generally encoded, interleaved, two-phase exchange key (BPSK) modulated with an orthogonal covering each

The BPSK symbol together with the Transverse Phase Key (QPSK) distribution of the masked symbols.

In the mobile-to-cell link, usage and audio channels are defined. The information communicated in the mobile-to-cell link channels is generally encoded, interleaved, and orthogonal signaled with QPSK propagation.

The features, objects and advantages of the present invention will become more apparent from the following detailed description taken in conjunction with the drawings, in which Figure 1 is a schematic view of an exemplary CDMA cellular telephone system;

Figure 2 is a block diagram of a cell site apparatus implemented in a CDMA cellular telephone system · Figure 3 is a block diagram of a cell site receiver;

Fig. 4 is a block diagram of an exemplary timing diagram of a cell site transmitting module from a detector; Figure 5 illustrates synchronization channel symbol synchronization;

Fig. 6 is an exemplary distortion channel timing together with;

Fig. 7 is an exemplary timing diagram of a tachogonal coverage timing diagram of a whole cell-to-mobile link;

is a block diagram of a mobile switching center image

20095859 prh 04-04-2011 hardware;

the picture is a block diagram of a mobile unit,

CDMAsuunniteltu

CDMA communication in a cellular telephone system;

Fig. 10 is a block diagram of a mobile station receiver;

Fig. 11 is a block diagram of a mobile transmitting modulator;

Fig. 12 is an exemplary timing diagram of a mobile-to-cell link for variable data rate in burst transmission; and

20095859 prh 04-04-2011 Figure 13 is an exemplary timing diagram of the entire mobile-to-cell link timing.

In a CDMA cellular telephone system, each cell site has a plurality of modulator-demodulator units or wide-spectrum modems. Each modem includes a digital wide spectrum transmit modulator, at least one digital wide spectrum data receiver, and a paging receiver. Each modem at the cell site is designated to a mobile unit, i.e., a mobile station, to enable communication with the designated mobile station.

The soft handoff system is used in a CDMA cellular telephone system where the new cell-site modem is assigned to the mobile station while the old cell-site modem continues the call service. When the mobile station is in the transition area between two cellular stations, the call may be switched back and forth between the cellular stations as determined by the signal strength. Because the mobile unit always communicates via at least one cell-site modem, there is less interruption effect on the mobile unit or service. The mobile unit uses a plurality of receivers to assist in the handover process, in addition to the multitasking function, to reduce the effects of wobble.

In a CDMA cellular telephone system, each cell site transmits a pilot carrier signal. If a cell is divided into sectors, each sector has a separate pilot signal within it within the cell. The mobile units use this pilot signal to provide system initial synchronization and to provide rough time, frequency, and phase tracking of signals transmitted by the cell site. In addition, each cell site transmits broad-spectrum modulated information, such as cell station identification information, system timing information, mobile paging information, and various other control signals.

20095859 prh 04 -04- 2011

The control signal transmitted by each cell site has the same spreading code but different code phase offset. The phase shift allows the control signals to be separated from one another, resulting in the separation of cell sites or sectors from which they originate. Using the same pilot signal code allows the mobile unit to find system timing synchronization through a single search through all pilot signal code steps. The strongest pilot signal determined by the correlation process for each code step is immediately identifiable. The strongest pilot signal detected corresponds to the pilot signal transmitted by the nearest cell site.

When acquiring the strongest pilot signal, i.e., initializing the mobile unit with the strongest pilot signal, the mobile unit looks for another carrier to be received by all users of the system in the cell. This carrier, referred to as a synchronization channel, transmits a broadcast message containing system information for use by the media in the system. The system information identifies the cell site and the system, in addition to carrying information that enables synchronization of long PN codes, interleaving frames, vocoders, and other system timing information used by the mobile unit without additional search. A second channel, called a paging channel, may be arranged to send messages to the mobile stations to indicate that a call has been received and to respond with channel assignments when the mobile station initiates a call.

The mobile unit continues to examine the received pilot carrier signal code in code transitions corresponding to pilot signals transmitted by the adjacent cell site sector. This investigation is performed to determine whether the pilot signal from the adjacent sector or cell is becoming stronger

20095859 prh 04 -04- 2011 as a pilot signal, which is first determined to be the strongest. If, when the call is in this inactive mode, the pilot signal from the adjacent sector or cell site becomes stronger than the pilot cell sector or. a pilot signal transmitted by the cell site, the mobile unit assumes the strongest pilot signals and the corresponding sector and cell station synchronization and paging channel.

Once the call has been initiated, a pseudo-noise (PN) code address is assigned to be used during this call. The code address may either be designated by the cell station or be predefined based on the identifier of the mobile unit. After the call is initiated, the mobile station continues to examine the pilot signal of the cell site through which the communication is established, in addition to the pilot signal of adjacent sectors or cells. Examination of the pilot signal continues to determine whether one of the pilot signals transmitted by the adjacent cell station or sector becomes stronger than the pilot signal transmitted by the cell station with which the mobile unit is in communication. When the pilot signal from a cell station in an adjacent cell or cell sector becomes stronger than the pilot signal transmitted by the current cell or cell sector, it indicates to the mobile unit that a new cell has been reached and that handover should be initiated.

Figure 1 shows an example of a telephone system in which the present invention has been applied. The system illustrated in Figure 1 employs wide-spectrum modulation technology for communication between mobile units of the system or between mobile phones and cellular stations. Cellular systems in large cities can have hundreds of cellular stations that serve hundreds of thousands of mobile phones. The use of wide spectrum technology, especially CDMA, makes it easy to increase the user capacity of systems of this size

20095859 prh 04 -04- 2011 compared to conventional FM modulation cellular systems.

Figure 1 shows a system controller and a switch 10, also referred to as a mobile switching center (MTSO), and including an interface and processing circuit for providing system control to cellular stations. Controller 10 also controls the routing of calls from the public switched telephone network (PSTN) to a suitable cell site for transmission to a suitable mobile communication unit. The controller 10 also controls the routing of calls from the mobile units through at least one cell site to the PSTN. Controller 10 can control calls between mobile stations through suitable cell sites, since such mobile units typically do not communicate directly with one another.

Controller 10 may be connected to cellular stations in various ways, such as on dedicated telephone lines, dedicated telephone lines optical fiber links or microwave communication links. Figure 1 shows two such cellular stations 12 and 14 and mobile units 16 and 18, each of which includes a cellular telephoneix. The cell stations 12 and 14 described and shown in the drawings are expected to serve the entire cell. However, it is to be understood that a cell may be geographically divided into sectors, whereby each sector is treated as a different coverage area. Accordingly, handovers are made between sectors of the same cell as described herein for multiple cells, while multiplicity can be achieved between sectors as well as between cells.

In Figure 1, arrows 20a-20b and arrows 22a-22b, respectively, define possible communication links between the cell site 12 and the mobile units 16 and 18. Similarly, arrows 24a-24b and 26a-26b respectively define possible communication links between the cell site 14 and the mobile units 16 and 18. Cell stations 12 and 14 transmit nominally using the same power.

20095859 prh 04 -04- 2011

Cell site service areas or cells are geographically designed such that the mobile unit is normally located closer to one cell site, and within a single cell sector if the cell is subdivided into sectors. When the mobile unit is idle, i.e. no calls are being processed, the mobile unit continuously monitors pilot signal transmissions from each nearby cell site, and suitably from a single cell station in which the cell is sectored. As shown in Figure 1, cellular stations 12 and 14 send pilot signals to mobile unit 16 from the outside or forward via communication links 20a and 26b. The mobile unit determines which cell it is in by comparing the strength of the pilot signals transmitted from cell stations 12 and 14.

In the example shown in Figure 1, the mobile unit 16 may be considered closest to the cell site 12. When the mobile unit 16 initiates a call, a control message is sent to the closest cell site, cell station 12. Upon receiving the call request message, cell station 12 The system controller 10 then connects the call via the PSTN to the desired recipient.

If the call is received in the PSTN, the system controller 10 transmits the call information to all cell stations in the area. The cellular stations, in turn, send a paging message to each of the desired mobile recipients called within the respective coverage area. When the intended mobile unit hears a paging message, it responds with a control message sent to the nearest cellular station. This control message gives a signal to the system controller that this particular cell site is communicating with the mobile unit. The controller 10 then routes the call through this cell site to the mobile unit. If the mobile unit 16 moves out of coverage of the original cell site, cell station 12

20095859 prh 04-04-2011, so an attempt is made to continue the call by routing the call through another cell site.

With regard to cellular telephone systems, the Federal Communications Commission (FCC) has allocated a total of 25 MHz for cell-to-cell links and 25 MHz for cell-to-cell links. The FCC has split its award between two service providers, one wired in the service area and the other selected by lottery. Due to the order in which the concessions were made, the 12.5 MHz allocated to each carrier in each direction of the link is further subdivided into two subbands. For wire carriers, the subbands are both 10 MHz and 2.5 MHz wide. For wireless carriers, the subbands are both 11 MHz and 1.5 MHz wide. Thus, a signal bandwidth of less than 1.5 MHz would fit into any subband while a bandwidth of less than 2.5 MHz could be fitted to all but one subband.

To maintain maximum flexibility in the allocation of CDMA technology to the available cellular frequency spectrum, the waveform used in a cellular telephone system should have a bandwidth less than 1.5 MHz. A good next choice would be a bandwidth of about 2.5 MHz, which allows full flexibility for wire cellular carriers and almost full flexibility for wireless cellular carriers. While the use of a wider bandwidth has the advantage of providing increased multi-path discrimination, there are disadvantages of higher hardware costs and less flexibility in the allocated bandwidth.

In a wide spectrum cellular telephone system such as that shown in Figure 1, the preferred implementable waveform structure comprises a direct sequence pseudo-noise wide spectrum carrier. The PN sequence return frequency is selected at 1.2288 MHz in a preferred embodiment. This

20095859 prh 04-04-2011 a particular return frequency is selected such that the resulting bandwidth, after about 1.25 MHz filtering, is about one-tenth of the total bandwidth allocated to a single cellular service carrier.

Another consideration when choosing the exact bit rate is that it is desired that the return frequency be accurately divisible by the baseband data rates used in the system. It is also desirable for the divider to have the power of two. In a preferred embodiment, the baseband data rate is 9600 bits per second, resulting in a selection of 1.2288 MHz, 128 times 9600, PN return frequency.

In the cell-to-mobile link, the binary sequences used to spread the spectrum are constructed from two different types of sequences, each with different properties to give different functions. There is an outdoor code, divided by all the signals in a cell or sector, that is used to discriminate between multipath signals. The outdoor code is also used to discriminate between signals transmitted from different cells or sectors to mobile units. There is also an internal code used to discriminate between user signals transmitted by a single sector or cell.

In a preferred embodiment of the signals transmitted by the cell site, the carrier waveform structure uses a sinusoidal carrier which is quadrature-modulated with a pair of binary PN sequences, which gives an external code transmitted by a single sector or cell. The sequences are generated by two different PN generators having the same sequence length. One sequence biphasic modulates the carrier single-phase channel (Channel I) and the other sequence biphases the carrier transverse phase (Q Channel). The resulting signals are summed to form a composite phase carrier.

Although values ​​of logical zero and logic one are usually used to represent binary sequences, the signal voltages used in the modulation process are + V volts for logic one and -V volts for logic zero. For biphasic modulation of a sine wave signal, a zero-volt average sinusoid is multiplied at + V or -V by the binary sequences using a multiplier circuit. The resulting signal can then be band-limited by a band-pass filter. It is also known to low pass filter a binary sequence stream before multiplying by a sinusoidal signal, thereby changing the order of operations. A four-phase modulator includes two biphasic modulators, each operated by a different sequence and with sinusoidal signals used in biphasic modulators, with a 90 degree phase difference.

In a preferred embodiment, the transmitted signal. the carrier sequence length is selected to be 32768 pieces. Sequences of this length can be generated by a modified maximum length linear sequence generator by inserting a zero bit into a 32767 length piece sequence. The resulting sequence has good cross-correlation and autocorrelation properties. Good cross-correlation and autocorrelation properties

20095859 prh 04 -04- 2011 are essential to prevent interference between pilot carriers transmitted by different cells.

A sequence of this short length is desirable to minimize the access time of mobile stations when they first enter the system without information on system timing. With unknown timing, the entire sequence length must be searched to find the correct timing. The longer the sequence, the longer the yield search will take. Although shorter sequences than 32768 could be used, it will be appreciated that when the length of the sequence is reduced,

20095859 prh 04-04-2011 the confirmation will decrease. As the processing gain decreases, the control of the multipath disorder and interference from other cells and sources is also reduced, perhaps to unacceptable levels. Thus, there is a desire to use the longest sequence that can be obtained within a reasonable time. It is also desirable to use the same code polynomials in all cells so that a mobile unit that does not know which cell it is in when it initially acquires synchronization can obtain full synchronization by retrieving a single code polynomial.

To simplify the pacing process, all cells in the system are synchronized to each other. In the exemplary embodiment, cell synchronization is performed by synchronizing all cells to a common time reference, the Navstar Global Positioning System, which itself is synchronized to Universal Coordinated Time (UTC).

Signals from different cells are separated by providing time shifts in the basic sequences. Each cell is assigned a time shift of different base sequences that is different from its adjacent ones. In a preferred embodiment, the 32768 repetition period is divided into 512 time offsets. The 512 transitions are 64 pieces apart. Each sector of each cell of a cellular system is also assigned a different transition for use in all its transmissions. If the system has more than 512 sectors or cells, then transitions can be reused in the same way as frequencies are reused in the current analog FM cellular system. Other structures could use a different number than 512 transitions. With reasonable care in detecting pilot signal transitions, it would never be necessary for nearby neighbor cells to use time shifts for nearby neighbors.

All signals transmitted by a cell or one sector share the same external PN codes for I and Q channels

20095859 prh 04 -04- 2011 for. The signals are also propagated by an internal orthogonal code generated using Walsh functions. The signal assigned to a particular user is multiplied by the outer PN sequences and a particular Walsh sequence, or by the sequence of Walsh sequences transmitted by the system controller during the call. The same core code is used on both I and Q channels, resulting in a modulation that is effectively biphasic for the core code.

It is well known that a set of n orthogonal binary sequences, each of which has a power of n, n of any 2, can be generated, see Digitai Communications with Space Applications, S. Golomb W. et al., Prentice-Hall, Inc, 1964, p. 45-64. In fact, orthogonal binary sequence sets for most lengths of four times and less than two hundred are also known. One class of such sequences that is easy to generate is called the Walsh function, also known as Hadamard matrices.

The Walsh function of an order number n can be determined reversibly as follows:

W (n) = · W (n / 2), W (n / 2) ® • W (n / 2), W '(n / 2) · where W' represents the logical complement of W, and W (1 ) = | 0 | .

Therefore,

<td rowspan="2">W (2)</td><td rowspan="2"> ©</td><td rowspan="2"> 0, 0,</td><td colspan="2"> 0 ®</td>

<td> 1 ®</td><td>and</td>

<td></td><td></td><td> 0,</td><td> 0, 0</td><td> 0 ·</td>

<td>W (4)</td><td> = ©</td><td> 0,</td><td>1, o,</td><td> 1 »</td>

0, 0,

1,

<td></td><td> &amp;</td><td> 0,</td><td>I,</td><td>I,</td><td> 0</td><td></td><td></td><td></td><td></td><td></td>

<td colspan="2">W (8) is as follows:</td><td></td><td></td><td></td><td></td><td></td><td></td><td></td><td></td><td></td>

<td></td><td> 9</td><td> 0,</td><td> 0,</td><td> 0,</td><td> 0,</td><td> 0,</td><td>No,</td><td> 0,</td><td> 0</td><td> •</td>

<td></td><td>s</td><td> 0,</td><td>I,</td><td> 0,</td><td> 1,</td><td> 0,</td><td> 1,</td><td> 0,</td><td> 1</td><td></td>

<td></td><td> ©</td><td> 0,</td><td> 0,</td><td>I,</td><td>I,</td><td> 0,</td><td>No,</td><td>I,</td><td> 1</td><td></td>

<td>W (8) =</td><td>s</td><td> 0,</td><td>I,</td><td>I,</td><td> 0,</td><td> 0,</td><td> 1,</td><td>I,</td><td> 0</td><td> . ©</td>

<td></td><td></td><td> 0,</td><td> 0,</td><td> 0,</td><td> 0,</td><td> 1,</td><td> 1/</td><td>I,</td><td> 1</td><td> ©</td>

<td></td><td></td><td> 0,</td><td>I,</td><td>No,</td><td> 1,</td><td>I,</td><td>No,</td><td> 1,</td><td> 0</td><td> •</td>

<td></td><td></td><td> 0,</td><td> 0,</td><td> 1,</td><td> 1,</td><td> 1,</td><td>I,</td><td> 0,</td><td> 0</td><td></td>

<td></td><td>s</td><td> 0,</td><td> 1,</td><td> 1,</td><td> 0,</td><td> 1,</td><td> 0,</td><td> 0,</td><td> 1</td><td></td>

<td>Walsh sequence</td><td></td><td></td><td>is</td><td></td><td></td><td>exc</td><td>i</td><td></td><td>Wa</td><td>ilsh</td>

From the rows of Chapter n function matrices.

The Walsh function contains n sequences, each of which is n bits long.

The Walsh function of the sequence number (as well as any other orthogonal function) has the property that n between the code symbols is zero between all the different sequences provided with each other.

each sequence of cross-correlation within a set is that the sequences are time-aligned This can be seen by noting that it differs from every other sequence in exactly half of its bits. It should also be noted that there is always one sequence containing only zeros and that all other sequences contain half of the ones

20095859 prh 04 -04- 2011 and half zeros.

Walsh sequences can be reused by neighbor cells and sectors because the external PN codes used in neighboring cells and sectors are different. Due to the different propagation times of the signals between a particular mobile station and two or more different cells, it is not possible to satisfy the time equalization condition required for Walsh function orthogonality for both cells simultaneously. So it is necessary to rely on the external PN code to discriminate the mobile unit from different cells.

20095859 prh 04 -04- 2011 between the signals. However, all signals transmitted by a cell are orthogonal to each other and thus do not cause interference to each other. This eliminates most interference in most locations, allowing for greater capacity.

Further, the system imagines the audio channel to be a variable rate channel whose data rate can be changed from a data block to a data block with the minimum increment required to control the data rate in use. The use of variable data rates reduces mutual interference by eliminating unnecessary transmissions when there is no usable speech to transmit. The vocoders use algorithms to generate a variable number of bits in each vocoder block according to variations in speech activity. During active speech, the vocoder may produce 20 ms data blocks containing 20, 40, 80, or 160 bits, depending on the activity of the speaker. It is desirable to transmit data blocks at a fixed time by varying the transmission rate. It is also desirable not to require signaling bits to inform the receiver how many bits are transmitted.

The blocks are further encoded using a cyclic redundancy check code (CROC) which appends to the block an additional set of parity bits that can be used to determine whether the data block is correctly decoded. CROC check codes are generated by dividing the data block by a predefined binary polynomial. The CRCC consists of all or part of the residual bits of the allocation process. The CRCC is checked at the receiver by re-generating the same residue and checking to see if the received residual bits are the same as the re-generated bits.

In the disclosed invention, the receiving decoder decodes the block as if it contains 160 bits, and then again as if it contains 80 bits, etc., until all possible block lengths are tested

20095859 prh 04-04- 2011 Tent. The CRCC is calculated for each experiment decoding. If one of the test decoders leads to the correct CRCC then the data block is accepted and passed to the vocoder for further processing. If no test decoding produces a valid CRCC, the received signals are fed to a system signal processor, where other process functions can alternatively be performed.

In a cellular transmitter, the power of the transmitted waveform is varied when the data rate of the block is changed. The highest data rate uses the highest carrier power. When the data rate is less than the maximum, the modulator, in addition to the Power Reduction, repeats each set of encoded data symbols a number of times necessary to achieve the desired transmission rate. For example, at the lowest transmission rate, each encoded symbol is repeated four times.

In a mobile station transmitter, peak power is kept constant, but the transmitter is closed for 1/2 or 1/4 or 1/8 time depending on the number of bits transmitted in the data block. The transmitter turn-on times are pseudorandom according to the assigned user code of the mobile station.

CELL-TO-MOBILE LINK

In a preferred embodiment, the size n of the Walsh function is set to sixty-four (n = 64) on the cell-to-mobile link. Therefore, each of the up to sixty-four different transmitted signals is assigned a unique orthogonal sequence. The forward error correction coded (FEC) symbol stream for each voice conversation is multiplied by its given Walsh sequence. The Walsh-coded / FEC-coded symbol stream for each audio channel is then multiplied by an external PN-coded waveform. The resultant applied symbol streams are then added together to form a composite waveform.

The resulting composite waveform is then modulated into a sinusoidal carrier waveform, bandpass filtered, converted to a desired operating frequency, amplified and radiated by an antenna system. Various embodiments of the invention may reverse the order of some of the procedures just described to generate a signal transmitted by a cell site. For example, it may be desirable to multiply each audio signal channel with an external PN coded waveform and perform filtering before summing all the channel signals emitted by the antenna. It is known in the art that the order of linear functions can be reversed to achieve different implementation advantages and different structures.

The waveform structure of the present invention for cellular service utilizes a pilot carrier approach to a cell-to-mobile link, as described in US 4,901,307. All cells transmit pilot carriers with a sequence of 32768 lengths, but to prevent interference between different cells.

The pilot carrier uses the same timing transitions all-zero

A Walsh sequence, i.e., a Walsh sequence consisting of just zeros present in all Walsh function sets. The use of the all-zero Walsh sequence for all cell pilot carriers allows pilot20095859 prh 04 -04-2011 carrier search. to ignore Walsh functions until the PN code synchronization is achieved. The Walsh frame is locked to the PN code pulse by the length of the Walsh frame, which is a PN sequence length factor. Therefore, provided that the PN address cell address transitions are multiples of 64 bits (or Walsh frame length), the Walsh frame is implicitly known from the outer PN code timing cycle.

All cells in the service area are equipped with precise synchronization. In a preferred embodiment, the GPS receiver in each cell synchronizes with the local one

20095859 prh 04 -04-2011 Universal Coordinated Time (UTC) for waveform timing. The GPS system allows time synchronization to better than 1 microsecond. Precise cell synchronization is desired to allow easy handover of calls between cells as mobile stations move from one cell to another while a call is in progress. If the neighboring cells are synchronized, then the mobile unit will have no difficulty in synchronizing with the new cell, thereby allowing for a soft handover.

The pilot carrier is transmitted at a higher power level than the typical audio carrier to provide a higher signal-to-noise ratio and interference margin for this signal. The higher power pilot carrier enables high-speed initial search and very accurate pilot carrier carrier tracking with a relatively wide bandwidth tracking circuit. The carrier phase obtained by pilot carrier tracking is used as a reference for the carrier phase to demodulate the carriers modulated by the user information signals. This technique allows many user carriers to distribute a common pilot signal to a carrier phase reference value. For example, in a system transmitting a total of fifteen simultaneous audio carriers, a pilot carrier could be provided with transmission power equal to four carriers.

In addition to the pilot carrier, the cell site transmits another carrier intended to receive all system users in the cell. This carrier, called the synchronization channel, also uses the same 32768-length PN sequence for spectrum spreading but with a different predetermined Walsh sequence. The synchronization channel transmits a broadcast message containing system information for use by the mobile stations in the system. System information identifies the cell site and system and

20095859 prh 04 -04- 2011 carries information that allows the use of long PN codes to synchronize mobile information signals without additional search.

A second channel, called a paging channel, may be arranged to send messages to the mobile stations indicating that the call has been received by them and to respond with channel transfers when the mobile station initiates the call.

Each audio carrier transmits a digital presentation of speech to the call. The analog speech waveform is digitized using standard digital telephone technology and then compressed using a vocoding process at a data rate of about 9600 bps. This data signal is then convolutionally encoded at r = 1/2, forced length K = 9, repetition, and interleaved to provide error detection and correction operations that allow the system to operate at a much lower signal-to-noise and interference ratio.

Convolutional coding, repetition and interleaving techniques are well known.

The resulting encoded symbols are multiplied by the assigned Walsh sequence and then multiplied by the outer PN code. This process results in a PN sequence rate of 1.2288 MHz or 128 times 9600 bps data rate. The resulting signal is then modulated onto an RF carrier and summed with pilot and setup carriers together with other audio carriers. Summing can be accomplished at multiple points in processing, such as the IF frequency or the baseband either before or after the PN sequence multiplication.

Each audio carrier is also multiplied by a value that sets its transmitted power relative to the power of the other audio carriers. This power control feature permits the assignment of power to those links which require higher power due to the relatively disadvantaged position of the intended recipient.

20095859 prh 04-04-2011

The mobile stations are provided with means for reporting the received signal-to-noise ratio to allow power to be set to a level that provides sufficient performance without loss. The orthogonality of Walsh functions is not disturbed by using different power levels for different sound carriers, provided that time alignment is maintained.

Figure 2 is a block diagram illustrating an embodiment of a cell site apparatus. The cellular station uses two receiver systems, each with a different antenna and an analog receiver for multipath reception. In both receiver systems, the signals are similarly processed until the signals undergo a multipath combining process. The elements within the dotted lines correspond to the elements corresponding to the communication between the cell site and one mobile unit. The output of the analog receivers is also arranged on other elements used for communicating with other mobile units.

In Figure 2, the first receiver system includes an antenna 30, an analog receiver 32, a candidate receiver 34 and a digital data receiver 36. The first receiver system may also include an alternative digital data receiver 38. The second receiver system includes an antenna 40, analog receiver 42, applicant receiver 44 and digital data receiver 46.

The cell site also includes a cell site control processor 48. The control processor 48 is coupled to the data receivers 36, 38 and 46 together with the applicant receivers 34 and 44. The control processor provides, among other functions, functions such as signal processing; generating a timing signal; power control; and handover, multipath, multipath control, and system control processor interface with the MTSO (Figure 8). Walsh sequence indication as well

20095859 prh 04 -04-2011 transmitter and receiver assignment is also provided by control processor 48.

Each receiver system is connected by data receivers 36, 38 and 46 to a multipath combiner and decoder circuit 50. Digital link 52 is coupled to receive an output of the multipath combiner and decoder circuit 50. Digital link 52 is also coupled to control processor 48, cell-site transmit modulator 54 and MTSO digital switch. Digital link 52 is used to communicate signals from the MTSO and the MTSO (FIG. 8) to the cell-site transmit modulator 54 and the circuit 50 under control of the control processor 48.

The signals transmitted by the mobile station are direct sequence wide spectrum signals which are modulated by a PN sequence clocked to a predetermined frequency which is 1.2288 MHz in a preferred embodiment. This clock frequency is selected to be an integer multiple of 9.6 Kbps baseband data rate.

Signals received by antenna 30 are provided to analog receiver 32. Details of receiver 32 are further illustrated in Figure 3. Signals received by antenna 30 are provided to a down mixer 100 consisting of an RF amplifier 102 and mixer 104. The received signals are input to the RF amplifier where Mixer 104 is provided with a second input which is output of frequency synthesizer 106. In the mixer 104, the amplified RF signals are converted to an IF frequency by mixing with a frequency synthesizer output signal.

The IF signals are then output from the mixer 104 to the heel filter (BPF) 108, which is typically a Surface Acoustic Wave (SAW) filter with a bandwidth of 1.25 MHz where it is bandpass filtered. The filtered signals are output from BPF 108 to IF amplifier 110 where the signals are amplified. The amplified IF signals are output to the IF

20095859 prh 04-04-2011 from amplifier 110 to analog-to-digital converter (A / D) 112, where they are digitized with a clock frequency of 9.8304 MHz, which is exactly 8 times the PN return frequency. Although the (A / D) converter 112 is shown as part of the receiver 32, it could instead be part of the data and paging receivers. The digitized IF signals are output from the A / D converter 112 to the data receiver 36, the alternative data receiver 38, and the applicant receiver 34. The signals output from the receiver 32 are I and Q channel signals, as discussed below. Although Fig. 3 shows that the A / D converter is a single device and the I and Q signals are divided later, it is imagined that the channel division can be done before digitization by two separate A / D converters arranged to digitize the I and Q channels. RF-IF baseband mixing and analog-to-digital conversion for I and Q signals are known in the art.

The applicant receiver 34 is used at the cell site to sense the time domain of the received signal to ensure that the digital data receiver 36, and the data receiver 38, if used, monitor and process the strongest time domain signal. The applicant receiver 64 provides a signal to the cell site control processor 48 which provides control signals to the data receivers 36 and 38 to select a suitable received signal for processing.

Signal processing at cell-site data receivers and applicant receivers is different in many respects than signal processing with similar elements in a mobile unit. In an internal, i.e. inverse or mobile-to-cell link, the mobile unit does not transmit a pilot signal that can be used for coherent comparison purposes in signal processing at the cell site. The mobile-to-cell link is characterized by non-coherent modulation

20095859 prh 04 -04- 2011 and as demodulation systems using orthogonal signaling of the 64j system.

In the 64 system orthogonal signaling process, symbols transmitted by the mobile unit. coded into 2⁶of 64, different binary sequences. The selected set of sequences is known as Walsh functions. The optimal reception function for m-system signal coding of the Walsh function is the so-called. Fast Hadamard Transform (FHT).

Referring still to Figure 2, the applicant receiver 34 and the digital data receivers 36 and 38 receive signals output from the analog receiver 32. In order to encode the wide spectrum signals transmitted to the particular cell site receiver through which the mobile unit communicates, proper PN sequences must be transmitted. Further details on generating signals for the mobile unit are described below.

As shown in Figure 3, the receiver comprises two PN generators, PN generators 120 and 122, which generate two different short code PN sequences of the same length. These two PN sequences are common to all cell site receivers and to all mobile units with respect to the modulation system outer code, as will be described in more detail below. The PN generators 120 and 122 thus provide the output sequences PN, respectively_Z and PN_Q. PN_Xand PN_qsequences are referred to as in-phase (I) and transverse (Q) channel PN sequences.

Two PN sequences, PN_r and PN_Q are generated by various 15th degree polynomials added to produce sequences of 32768 lengths rather than 32767 that would normally be produced. For example, insertion may occur in the form that one zero is added to a line of fourteen zeros, which occurs once in every maximum-length 15th order linear sequence. In other words, one state of the PN generator would be repeated during sequence generation. So

20095859 prh 04 -04- 2011 modified sequence contains one 15-digit line and one line with 15 zeros.

In the exemplary embodiment, receiver 36 also includes a long code PN generator 124 which generates a PN jj sequence corresponding to a PN sequence generated from a mobile unit by a mobile unit in a link. The PN generator 124 may be a maximum length linear sequence generator which generates a user PN code that is very long, e.g., degree 42, time-shifted by an additional factor, such as a mobile unit address or user ID, to discriminate among other users. Thus, the signal received by the cell site is modulated by both the long code PN j sequence and the short code PN j_xand PN_Qsequences. Alternatively, a non-linear encryption generator, such as an encryption device that uses a data encryption standard (DES) to encrypt a 64-character universal time representation using a user-specific key, may be used in place of the PN generator 124.

ΡΝ_σsequence output from PN generator 124 Exclusive-OR is implemented in PN_Xand PN_Qsequences in Exclusive OR gates 126 and 128, respectively, of PN sequences_X, and PN_Q, to achieve this.

Sequences PN_X, and PN_Q, is provided to PN-QPSK correlator 130 together with I and Q channel signals as output from receiver 32. The correlator

0 is used to correlate the I and Q channels_Σ, - and

PN_Q1sequences. I and Q channel outputs 132 and 134 respectively

Correspondingly, the correlators 130 of the correlator 130 are provided to the collectors as the symbol data is assembled as a 4-return input.

The outputs of the collectors 132 and 134 are provided as inputs Fast

Hadamard Transform (FHT). to the processor 136. The FHT processor 148 produces a set of 64 coefficients for each of the 6 symbols. The 64 coefficients are then multiplied by the weighting function generated in the control processor 48. The weighting function is linked

20095859 prh 04 -04- 2011 demodulated signal strength. The weighted data output from FHT 136 is provided to the multipath combiner and decoder circuit 50 (Figure 2) for further processing.

The second receiver system processes the received signals in the same manner as described for the first receiver system of Figures 2 and 3. The weighted 64 symbol output from the receivers 36 and 46 is provided to the multipath combiner and decoder circuit 40. The circuit 50 includes an adder which summs the 64 weighted coefficients from the receiver 36 to the 64 weighted coefficients from the receiver 46. The resulting 64 coefficients are compared with each other to determine the maximum coefficient. The magnitude of the comparison result, one of 64 coefficients, is used to determine a set of decoder weights and symbols for use in a Viterbi algorithm decoder implemented in circuit 50.

The Viterbi algorithm decoder included in the circuit 50 is of a type capable of decoding data encoded in a mobile unit with a constrained length K = 9 and a code rate r = 1/3. The Viterbi decoder is used to determine the most probable information bit sequence. Periodically, nominally 1.25 ms, a signal quality estimate is obtained and transmitted as a power control command of the mobile unit together with the data to the mobile unit. Further details of this quality evaluation are described in more detail in the aforementioned parallel patent application. This quality estimate is the average signal-to-noise ratio over a time interval of 1.25 ms.

Each data receiver follows the timing of the signal it receives. This is accomplished by the known technique of correlating the received signal with a slightly early local reference PN and correlating the received signal with a slightly late

20095859 prh 04 -04- 2011 by tailu-PN. The difference between the two correlations is, on average, zero unless there is a timing error. Conversely, if there is no timing error, this difference in the magnitude and error indication and the receiver timing are adjusted accordingly.

The cellular station further comprises an antenna 62 coupled to a GPS receiver 64. The GPS receiver processes signals received from the satellites 62 from the satellite navigation system of the Navstar Global Positioning System to obtain timing signals that indicate Universal Coordinated Time (UTC). The GPS receiver 64 provides these timing signals to the control processor 48 for timing synchronization at the cell site, as previously described.

In Figure 2, an alternate digital data receiver 38 may be included to improve system performance. The structure and operation of this receiver is similar to that described for the data receivers 36 and 46. Receiver 38 can be used in a cellular position to provide additional multimode emails. This additional data receiver, alone or in combination with others, may track and receive other possible delay paths of signals transmitted by the mobile unit. Alternative additional digital data receivers, such as receiver 38, provide additional multimode emails that are extremely useful in cellular stations located in densely built-up urban areas where multiple opportunities exist for multipath signals.

Signals from the MTSO are coupled to a suitable transmission modulator via digital link 52 under control of control processor 48. The transmission modulator 54, under the control of the control processor 48, broad-spectrum modulates the data for transmission to the intended receiving mobile unit. For more details on the broadcast mode

20095859 prh 04 -04- 2011 The structure and operation of the stator 54 will be described below with reference to Figure 4.

The output of the transmission modulator 54 is provided in a transmission power control circuit 56 where control of the transmission processor 48 can be used to control the transmission power. The output of the circuit 56 is provided in an adder 57, where it is summed with the output of the transmission modulator / transmit power control circuits directed to other cellular mobile stations. The output of the adder 57 is provided in a transmit power amplifier circuit 58, which has an output to the antenna 60 for transmitting mobile units within the cell service area. Figure 2 further illustrates pilot / control channel generators and transmit power control circuitry 66. Circuitry 66 generates a pilot signal, a synchronization signal, and a paging signal in control of the control processor and controls their power for connection to circuit 58 and output to antenna 60.

Figure 4 is a block diagram of an exemplary embodiment of a cell site transmitter. The transmitter includes. A pair of PN sequence generators used to generate an external code. These PN generators generate two different PN sequences, i.e. PN_Xand PN_Qsequences as described with reference to Figure 3. However, these PNs_Xand PN_Qsequences are time-delayed according to the sector or cell address.

In Figure 4, the transmitter circuit of Figure 3 is shown in more detail with pilot, synchronization, and audio channel signals. The transmitter circuit includes two PN generators, PN Generators 196 and 198, which generate PN_Zand PN_Qsequences. PN generators 196 and 198 respond to an input signal corresponding to a control processor sector or cell address signal to provide a predetermined time delay to the PN sequences. These time-delayed ΡΝ_Σand PN_Qsequences are relative to in-phase (I) and transverse-phase (Q) channels. Although only two PN generators are described to generate a PN_rand PN_Qsequences similar

20095859 prh 04 -04- 2011 for cell station or sector channels, it will be appreciated that many other PN generator systems can be implemented. For example, in a non-sectored cell, a pair of PN generators may be provided for each pilot, synchronization, paging, and audio channel to produce, in synchronization, the PNj and PN used in the outdoor code._Qsequences. This may be advantageous PN_Xand PN_Qsequences to avoid division into a large number of circuits.

In a preferred embodiment, the Walsh function coding of the channel signals is used as the internal code. In the exemplary numerology described herein, a total of 64 different Walsh sequences are available, with three of these sequences dedicated to pilot, synchronization, and paging functions. In pilot, synchronization, and paging channels, the input data is convolutionally coded and then interleaved, as is well known in the art. Further, convolutionally encoded data is also provided with repetition before interleaving, as is well known.

The pilot channel does not include any data modulation and is characterized by an unmodulated wide-band signal used by all users of a given cell site or sector for acquisition and tracking purposes. Each cell, or if it is divided into sectors, each sector has a unique pilot signal. However, rather than using different PN generators for the pilot signals, it has been found that a more efficient way to generate different pilot signals is to use transitions in the same base sequence. Using this technique, the mobile unit sequentially retrieves the entire sequence and tunes to the transition that produces the strongest correlation. With this base sequence offset, the offsets must be such that there is no need to disrupt or override the cells or sectors adjacent to the pilots.

20095859 prh 04 -04- 2011

Therefore, the pilot sequence must be long enough that different sequences can be generated by transitions in the base sequence to support a large number of pilot signals in the system. Further, the difference or transitions must be large enough so that there is no interference with the pilot signals. Accordingly, in the exemplary embodiment of the present invention, the length of the pilot sequence is chosen to be 2¹⁵. The sequence is generated starting from the sequence 2 ^ -1 with an extra zero attached to the sequence when a particular state is detected. In the exemplary embodiment, 512 different pilot signals are selected with the offsets in the base sequence having a 64-bit offset with a corresponding reduction in the number of different pilot signals.

When generating the pilot signal, Walsh zero (W_o) sequence consisting of zeros is used as if not to modulate the pilot signal, which is ΡΝ_Χand PN_Qsequences. Walsh zero {W_o) is therefore multiplied by ΡΝ_Σ~. and PN_Qsequences in Exclusive OR gates. Thus, the resulting pilot signal contains only PN_T-. and PN_Qsequences. With all cell sites and sectors having the same PN sequence for the pilot signal, the distinguishing feature at the initiation of cell-to-sector transmission is the sequence phase.

For the transmit modulator and power control circuitry section 66 for the pilot channel, the Walsh generator (W_o) 200 generates a signal that corresponds to the all-zero function, as just explained. The timing for generating the Walsh function is arranged by the control processor, as is the case with the Walsh function generators at the cell site and the mobile unit. The output of generator 200 is provided as input to Exclusive OR gates 202 and 204.

The second input of the Exclusive OR port 202 receives the PNj signal, while the Exclusive OR port

20095859 prh 04 -04- 2011 the second input of port 204 receives a PN signal. PNj and PN_Qsignals, respectively, Exclusive OR are implemented with the output of generator 200 and respectively input as Finite Impulse Response (FIR) filters 206 and 208. The filtered signals are output from FIR filters 206 and 208, which are provided in a transmit power control circuit 21 The signals provided to the gain control elements 210 and 212 are adjusted in response to the input signals from the control processor (not shown). The signals output from the gain control elements are provided to the transmission power amplification circuit 58, the detailed structure and operation of which are described below.

The synchronization channel information is encoded and then multiplied in the Exclusive OR gates by a predetermined Walsh sequence. In the exemplary embodiment, the selected Walsh function is (W₃₂) sequence consisting of the 32nd sequence followed by 32 zeros. The resulting sequence is multiplied by PN j and PN_Qsequences in Exclusive OR gates. In the exemplary embodiment, the synchronization channel data information is provided to the transmission modulator, typically at 1200 bps. In the exemplary embodiment, the synchronization channel data is preferably convolutionally coded at a rate r = 1/2 at a forced length K = 9 each code symbol repeated twice. This coding rate and length are common to all encoded forward link channels, i.e., synchronization, paging, and audio channels. In the exemplary embodiment, a shift register structure is used for code generators G1 = 753 (octal) and G2 = 561 (octal). In the exemplary embodiment, the symbol rate for the pacing channel is 4800 sps, i.e. one symbol is 208 microseconds or 256 PN chips.

The code symbols are interleaved by a convolutional interleaver, which in the exemplary embodiment has a period of 40

20095859 prh 04-04-2011 ms. Preliminary parameters of the interleaver are I = 16 and J = 48. Further information on interleaving can be found in Data Communication, Network and Systems, Howard W. Sams &amp; Co., 1987, pp. 343-352. The effect of the convolutional interleaver is to decompose untrusted channel symbols such that any two symbols in the adjacent sequence of 1-1 or less symbols are separated by at least J + 1 symbols at the delimiter output. Similarly, any two symbols in the adjacent sequence having J1 symbols are separated by at least 1 + 1 symbols at the delimitation output.

In other words, if I = 16 and J = 48, then in a 15-character string, the symbols are transmitted separated by 885 microseconds, which gives a time characteristic.

The synchronization channel symbols of a particular cell or sector are tied to a corresponding pilot signal for that cell or sector. Figure 5 illustrates the timing of two different pilot channels (N) and (N + 1) separated by a 64-bit shift. Fig. 5 is merely an exemplary timing diagram of the pilot and synchronization channels in the example, with the actual pilot channel blocks and synchronization channel symbols not shown. Each synchronization channel initiates a new pair of code symbols (c_x, c'J in the first code symbol il la (c_x), due to two code repetitions, moves in absolute time with an amount equal to the corresponding pilot.

As depicted in Figure 5, the N-pilot channel initiates a new interleaving cycle, pilot synchronization, at time t_x. Similarly, the N + 1 pilot channel initiates a new interleaving cycle or pilot synchronization at time t, which occurs 64 times later than time t_x. The pilot cycle is 26.67 ms in the exemplary embodiment, which corresponds to 128 synchronization channel code symbols or 32 synchronization channel information bits. The synchronization channel symbols are interleaved with a convolutional interleaver of 26.67 ms. Once the mobile unit has received a pilot

20095859 prh 04 -04- 2011 signal, it has instant synchronization channel selection.

The synchronization channel symbols are covered with a predetermined Walsh sequence to give orthogonality to the signal. In the synchronization channel, one code symbol carries four masking sequences, i.e., one code symbol per four iterations of 32-to-32 zeros as shown in Figure 6. As shown in Figure 6, one logical one represents 32 first Walshpals and one logical zero represents 32. zero Walsh pieces. However, the orthogonality of the synchronization channel remains even though the synchronization channel symbols vary in absolute time depending on the pilot channel, since the synchronization channel offsets are integers multiple of the Walsh frame.

In the exemplary embodiment, the synchronization channel messages are variable in length. The message length is an integer multiple of 80ms, which corresponds to 3 pilot cycles. The synchronization channel information bits include cyclic override confirmation bits (CRC) for error detection.

Figure 7 is a timing diagram showing the timing of the entire exemplary system. There are 75 pilot cycles over a two-second period. In Figure 7, the N-pilot and synchronization channels correspond to a sector or cell that uses a non-shifted pilot such that the pilot synchronization signals are aligned exactly with the UTC time. As such, the pilot synchronization, i.e. the initial state, aligns exactly with a common 1 pulse / s (pps) signal.

In all cases where the offset pilot is used, the PN phase offset corresponding to the pilot offset is used. In other words, pilot synchronization (initial) and synchronization channel messages are staggered with respect to 1 pps signals. The synchronization messages carry this phase shift information such that

20095859 prh 04-04-2011 The mobile unit may adjust its timing accordingly.

As soon as the synchronization channel message is correctly received, the mobile station has the ability to immediately synchronize with either the paging channel or the voice channel. In pilot synchronization, corresponding to the end of each synchronization message, a new 40 ms interleaving cycle begins. At this point, the mobile unit begins to delimit either the code repetition or (c, c,) the first pair of code symbols, with decoder synchronization achieved. The delimiter write address is initialized to 0 and the read address to J, memory delimiter synchronization is achieved.

The synchronization channel messages carry information depending on the state of the 42 bit length PN generator for communication with the mobile unit to the voice channel. This information is used in the digital data receivers of the mobile unit to synchronize the corresponding PN generators. For example, in FIG. 7, the synchronization channel message N + 1 contains a 42 bit area indicating state, the state X which the long code PN generator corresponding to the sector or cell audio channel will have at a predetermined later time, such as 160 ms later. After successful decoding of the synchronization channel message, the mobile unit loads X into the long code PN generator at the right time. The long code PN generator of the mobile unit is thus synchronized to allow the user messages to be re-edited.

With respect to the transmission modulator and power control circuitry 66 for the synchronization channel, the synchronization channel information is input from the control processor to the encoder 214. The synchronization channel data is convolutionally encoded by the encoder 214 in an exemplary embodiment as described above. The encoder 214 also provides repetition of the encoded symbols, in the case of a synchronization channel, the encoded symbols are repeated. The output symbols of the encoder 214 are provided to the interleaver 215, which gives the convolutional

20095859 prh 04 -04- 2011 sen. The interleaved symbols are output from the interleaver 215 as input to the Exclusive OR port 216.

The Walsh generator 218 generates a signal corresponding to the Walsh (W₃₂), which is provided as a second input to the Exclusive OR port 216. The synchronization channel symbol stream and the Walsh (W₃₂) sequence Exclusive-OR is implemented on Exclusive-OR port 216 and its result is provided as input to each of the exclusive OR ports 220 and 222.

The second input of exclusive OR port 220 receives a PN_xThe second input of the signal and the Exclusive OR port 222 receives the PN_Qsignal. PN_Xand PN_Qsignals are exclusively OR implemented exclusively with the output of the TA gate 218 and are provided as inputs to the Finite Impulse Response (FIR) filters 224 and 226, respectively. The filtered signals are output from the FIR filters 224 and 226 arranged in a digital transmission power control circuit. variable gain control elements 228 and 230. The signals provided to the gain control elements are digitally adjusted in response to the input digital signals from the control processor (not shown). The signals output from gain control elements 228 and 230 are provided to transmit power gain circuit 58.

The search channel information is also coded by repeating, interleaving and then multiplying by a predetermined Walsh sequence, the resulting sequence is then multiplied by PN_Xand PN_Qsequence. The paging channel data rate for a particular sector or cell is assigned in a specific area in a synchronization channel message. Although the data rate of the synchronization channel is variable, in the exemplary embodiment it is set to be fixed for each system at one of the example data rates: 9.6. , 4.8, 2.4 and 1.2 kbps.

With respect to the paging channel 'transmission modulator and power control circuit, the paging channel information is input from the control processor to the encoder 232. The encoder 232 is a

20095859 prh 04 -04- 2011 in a character application, a convolutional encoder which also provides symbol repetition according to the channel data rate. The output of encoder 232 is provided to interleaver 233, where symbols are convolutionally interleaved. Output. the interleaver 233 is provided as input to Exclusive OR gate 234. Although the paging channel data rate varies, the code symbol rate is kept constant at 19.2 ksps with code repetition.

The Walsh generator generates a signal corresponding to a predetermined Walsh sequence, which is provided as a second input to the Exclusive OR port 234. The symbol data and the Walsh sequence to the Exclusive OR port 234 is provided as an input to each of the ORs 23 and 240.

The second input of the exclusive OR port 238 receives the PNj signal and the second input of the exclusive OR port 240 receives the PNj signal_Qsignal. PN_Tand PN_Qsignals are exclusively OR implemented with an output of exclusively TAI port 234 and output as respective inputs of Finite-Impulse-Response (FIR) filters 242 and 244. The filtered signals are output from FIR filters 242 and 244 provided in a transmission power control circuit including amplifiers 246 and 248. The signals provided to gain control elements 246 and 248 are adjusted in response to input digital signals from the control processor (not shown). The signals output from the gain control elements are provided to the transmit power gain circuit 58.

The data for each audio channel is encoded by repeating, interleaving, editing, multiplying by a predetermined Walsh sequence (W₁ - wp and then multiplying by PN_Tand PN₀sequence. The Walsh sequence intended for use in a particular channel is assigned by the system controller during the call setup time in the same way as channels are assigned to calls in the analogue FM cellular system. In the foregoing, 41

20095859 prh 04-04-2011 In the character application, 61 different Walsh sequences are available for audio channels.

In the exemplary embodiment of the present invention, the audio channel uses a variable data rate. The purpose of variable data rate is to reduce the data rate when there is no voice activity, thereby reducing the interference this particular audio channel generates for other users. Such a vocoder produces data at four different data rates based on voice activity on a 20 ms frame basis. Exemplary data rates are 9.6 kbps, 4.8 kbps, 2.4 kbps, and 1.2 kbps. Although the data rate varies on a 20 ms basis, the code symbol rate is kept constant at code repetition of 19.2 ksps. Accordingly, the code symbols are repeated 2, 4, and 8 times for the respective data rates of 4.8 kbps, 2.4 kbps, and 1.2 kbps.

Because variable speeds are designed to reduce interference, at lower speeds, code symbols have less energy. For example, for example data rates of 9.6 kbps, 4.8 kbps, 2.4 kbps, and 1.2 kbps, code symbol energy (E_s) is E, respectively_b/ 2, E_b/ 4, E_b/ 8 and E_b/ 16, where E_b is the information bit energy for a transmission rate of 9.6 kbps.

The code symbols are interleaved with a convolutional interleaver such that code symbols having different energy levels are converted by the interleaver's operation. In order to track what energy level the code symbol should have, a symbol is assigned to each symbol specifying its data rate for scaling purposes. After orthogonal Walsh coverage and PN application, the transverse channels are digitally filtered as a Finite Impulse Response (FIR) filter. The FIR filter receives a signal corresponding to the symbol energy level to effect energy scaling according to the data rate. The I and Q channels are scaled by factors of 1, 1/2, 1/2 or 1/2 ^ 2. In one embodiment, the vocoder would give the data rate entry a 2-bit number format42

20095859 prh 04 -04- 2011 sa FIR filter to control filter scaling factor.

In Figure 4, the circuitry of two exemplary audio channels is shown in the audio channels (i) and (j). The audio channel (i) data is an input from a vocoder (not shown) to a transmission modulator 54 (Figure 3). The transmission modulator 54 includes an encoder 250_iZ limit 251_i; exclusiveTAI ports 252_iZ 255., 256₁ and 258_iZ· PN generator

253 ^ · and Walsh generator (WJ 254 ..

The audio channel (i) data is the input to the encoder 250_iZ where in the exemplary embodiment it is convolutionally encoded by code symbol repetition according to the input data rate. The encoded data is then provided to the interleaver 251_iZ where, in the exemplary embodiment, it is convolutionally interleaved. The interleaver 2517 receives from the vocoder associated with the audio channel (i) a 2-bit data rate title which is interleaved with the symbol data to identify the data rate to the FIR filters. The data rate item is not being sent. In the mobile unit, the decoder checks all possible codes. The interleaved symbol data is output from the interleaver 251_£ at an example rate of 19.2 ksps to the 252nd input of the Exclusive OR port.

In the exemplary embodiment, each voice channel signal is converted to provide greater certainty to the cellular mobile station during transmissions. Although such conversion is not required, it contributes to the security of communications. For example, conversion of audio channel signals may be accomplished by PN encoding the audio channel signals with a PN code determined by the mobile unit address of the user ID. This conversion can use ΡΝ_σsequence or encryption system as described for a particular receiver in a cell-to-cell communication. Accordingly, a separate PN generator may be provided for this function, as shown in the figure. Although the transformation is described with reference to the PN sequence, the transformation can be carried out

20095859 prh 04 -04- 2011 other methods, including known methods.

Referring again to FIG. 4, signal conversion of the audio channel (i) may be accomplished by providing a PN generator 253 which receives the assigned mobile unit address from the control processor. PNG Generator 253_x generates a unique PN code that is given as a second input to Exclusive OR port 252_χ. Output 252 of the exclusive OR port 255 is provided at one of the inputs of the exclusive OR port 255.

The Walsh generator (Wj 254. generates, in response to a function selection signal. And a timing signal, a signal corresponding to a predetermined Walsh sequence. The value of the function selection signal can be determined at the address of the mobile unit. is implemented by Exclusive OR port 255_iand the result is provided as input to each of the exclusive TAI ports 256 and 258₁. PN Generator 253. together with all other PN generators and Walsh generators at the cell site, output at 1.2288 MHz. Note that PN generator 253 includes a decimator that outputs at 19.2 kHz at Exclusive OR port 255.

Exclusive OR port 256_i the input receives the PN_x258, the second input of the Signal and the Exclusive OR port receives the PN_Qsignal. PN_Xand PN_Qthe signals Exclusive OR are implemented with the output of Exclusive OR gate 252, and output as inputs to the Finite Impulse Response (FIR) filters 260 and 262, respectively. The input symbols are filtered from the convolutional interleaver 251 according to the incoming input data rate designation (not shown). The filtered signals are output from the FIR filters 260_i and 262, which are arranged in a transmission power control circuit 56 which includes

20095859 prh 04-04-2011 gain adjustment elements 264_i and 266 ^ to gain control elements 264_Α and 266_i the given signals are gain adjusted in response to the input digital signals from the control processor (not shown). The signals output from the gain control elements are provided to the transmit power gain circuit 58.

In addition to the audio bits, the forward link channel carries power control information. The power control bit rate in the exemplary embodiment is 800 bps. The cell-site receiver, which demodulates the mobile-to-cell signal from a particular mobile station, generates power control information which is included in the cellular mobile station in an audio channel assigned to that mobile station. Further details of the power control feature are described in the aforementioned parallel patent application.

The power control bits are incorporated into the output of the convolutional interleaver by a technique called code symbol punching. In other words, whenever a power control bit is required to be transmitted, the two code symbols are replaced by two identical code symbols having the polarity provided by the power control information. Further, the power control bits are transmitted at an energy level corresponding to an energy level of 9600 bps.

An additional binding condition assigned to the power control information stream is that the location of the bits must be randomized in the mobile-to-cell channels. Otherwise, the full energy power control bits would generate interference peaks at regular intervals, thereby reducing the detectability of such bits.

Figure 4 further shows an audio channel (j) identical in function and structure to those of the audio channel (j). It is contemplated that there are many more audio channels (not shown) with the total of 61 audio applications shown.

20095859 prh 04 -04- 2011

Referring to the Walsh generators of Figure 4, the Walsh functions are a set of orthogonal binary sequences that are readily generated in a known manner. An interesting feature of the Walsh function is that each of the 64 sequences is completely orthogonal to all other sequences. Thus, in this sequence the sequence pair differs exactly as many bit positions as they are similar, i.e. 32 out of 64 symbols. When information is encoded for transmission by Valsal sequences, the receiver is able to select any of the Walsh sequences as the desired carrier signal. Any signal energy encoded for other Walsh sequences will be rejected and will not result in interference with the desired Walsh sequence.

In the exemplary embodiment of the cell-to-mobile link, the synchronization, paging, and voice channels, as previously mentioned, are convolutionally encoded at a constrained length K = 9 and at a code rate r = 1/2, i.e., two encoded symbols are produced and transmitted for each bit of information transmitted. In addition to convolutional coding, convolutional interleaving of symbol data is also used. It is further contemplated that repetition is also used in conjunction with convolutional coding. In a mobile unit, the optimal decoder for this type of code is a soft-ended Viterbi algorithm decoder. The standard structure can be used for decoding purposes. The resulting decoded information bits are input to the digital carrier of the mobile unit and to the bandwidth device.

Referring again to Figure 4, circuit 58 includes a series of digital-to-analog (D / A) converters for converting to digital - infomation PN_Zand PN_Qbroadcast data for pilot, pacing, paging and audio channels in analog format. Pilot channel PNj spreading data is output from gain control element 210 to D / A converter 268. Digitized data is output from D / A converter 268 sum

20095859 prh 04-04-2011 to maime 284. Likewise, outputs from respective gain control elements for synchronization, paging, and audio channel PNj spreading data, i.e. gain control elements 228, 246, and 264_£ - 264_JZ is output to D / A converters 272, 276, and 28CL to 280, respectively, where the signals are digitized and output to adder 284. PN_Qspreading data for pilot, pacing, paging and audio channels as output from gain control elements 221, 230, 248 and 266_x - 266 ^, is given to D / Amuunts 270, 274, 278 and 282, respectively_i - 282_JZ wherein the signals are digitized and fed to adder 286.

Summa 284 totals PN_zspreading data for pilot, synchronization, paging and audio channels and adder 286 PN_Qdistribution data for the same channels. The summed I and Q channel data are fed together to the local oscillator (LO) frequency signals Sin (2nft) and Cos (2nft) mixers 288 and 290, where they are mixed and supplied to adder 292. The LO frequency signals 3ϊη (2πίϋ) and Cos (2Kft) are given suitable frequency sources (not shown). These mixed IF signals are summed in adder 292 and output to mixer 294.

The mixer 294 mixes the summed signal with the RF frequency signal provided by the frequency synthesizer 296 to provide a frequency-up conversion for the RF frequency band. The RF signal output from the mixer 294 is pass-filtered by the band-pass filter 298 and output to the RF amplifier 299. The amplifier 299 amplifies the frequency-band-limited signal according to an input gain control signal from the transmit power control 56 (Figure 3).

It is to be understood that the illustrated embodiment of transmit power gain circuit 58 is for presentation purposes only, with many transforms being able to add, mix, filter and amplify the signal, as is known in the art.

20095859 prh 04-04- 2011

The cell site control processor 48 (Figure 3) is responsible for assigning digital data receivers and transmission modulators to a particular call. The control processor 48 also monitors the progress of the call, the quality of the signals, and initiates decompression when the signal is lost. The cellular station communicates with the MTS0 via link 52, where it is connected by a standard telephone wire, optical fiber, or microwave link.

Figure 8 shows in block diagram form the equipment used in the MTSO. The MTSO typically includes a system controller or control processor 300, a digital switch 302, a multipath combiner 304, a digital video encoder 306 and a digital switch 308. Although not shown, additional multipath connectors and digital video encoders are coupled between digital switches 302 and 308.

When cellular mode is active, the call is handled by two cellular stations. Accordingly, the signals arrive at the MTSO from more than one cell site with nominally the same information. However, due to fluctuation and interference in the internal and reverse link from the mobile unit to the cell sites, the signal from one cell site may be of better quality than the signal from another cell site.

Digital switch 302 is used to route the information stream corresponding to a particular mobile unit from one or more cell stations to a multipath combiner 304 or a corresponding multipath combiner determined by a signal from a system control processor 300. When the system is not in cellular multipath mode, the multipath combiner can either be bypassed or input the same information at each input port.

A plurality of serially connected multipath connectors and vocoders are arranged in parallel, nominally one for each call to be processed. The multipath combiner 304 compares signal quality indicators by monitoring Informant bits from two or more cells

20095859 prh 04 -04- 2011 station signal. The multipath combiner 304 selects those bits corresponding to the highest quality cell position on a frame-by-frame basis for information to be output to vocoder 306.

The vocoder 306 converts the digitized audio signal format to a standard 64Kbps PCM phone format, analog, or any other standard format. The resulting signals are transmitted from vocoder 306 to digital switch 308. Controlled by system control processor 300, the call is routed to PSTN.

The audio signals from the PSTN for the mobile units are provided to digital switch 308 for connection to a suitable digital video encoder, such as vocoder 306, under the control of the system control processor 300. Vocoder 306 encodes the incoming digitized audio signals and outputs the resulting information bit stream directly to digital switch 302. Controlled by the system control processor, digital switch 302 directs the encoded data to the cell site (s) to which the mobile station is communicating. Although it has been previously described that the information transmitted to the MTSO is analog audio, it is still conceivable that the digital information may also be communicated in the system. To ensure compatibility in the system, the correct structure of the data must be taken care of.

If the mobile unit is in a handover mode that communicates with multiple cell sites or in a multi-cell mode, digital switch 302 routes the calls to suitable cell stations for transmission to a receiving cellular unit for use with a suitable cell site transmitter. However, if the mobile unit communicates with only one cell site or not in the cellular multipath mode, the signal is directed to only one cell site.

20095859 prh 04 -04- 2011

System control processor 300 provides control to digital exchanges 302 and 306 for routing data to and from the MTSO. The system control processor 300 also orders call transfer to cell sites and vocoders in the MTSO. Further, the system control processor 300 communicates with each cell site control processor. with the transfer of certain calls between the MTS0 and the cell-site, and the assignment of PN codes to the calls. It will further be understood that when shown in Figure 8, digital switches 302 and 306 are shown as two separate switches, however, it will be understood that this function may be performed by a single physical switching unit.

When the cellular multipath mode is enabled, the mobile unit uses the paging receiver to identify and obtain the strongest multipath signal from each of the two cellular stations. Digital data receivers are controlled by the applicant receiver and the control processor to demodulate the strongest signals. When the number of receivers is less than the number of cell sites transmitting information in parallel, multipath capability is possible. For example, with only one data receiver and two cell stations transmitted, the applicant monitors the pilots from each cell site and selects the strongest signal to demodulate the receiver. In this embodiment, the selection can be made as frequently as each vocoder frame, or approximately every 20 milliseconds.

The system control processor is responsible for assigning digital data receivers and modulators at the cell site to handle certain calls. Thus, in the cell-to-mobile link, the system control processor controls the assignment of Walsh sequences used to send a particular call to the mobile station at the cell site. In addition, the system control processor controls the receiver's Walsh sequences and PN

20095859 prh 04-04- 2011 codes. In the mobile-to-cell link, the system control processor also controls the mobile unit PN codes for the call. Addressing information is therefore transmitted from the MTSO to the cell site and from there to the cell to the mobile station. The system control processor also monitors the progress of the call, the signal quality, and begins decompression when the signal is lost.

MOBILE TO CELL LINK

In the mobile-to-cell link, the channel characteristics dictate that modulation techniques must be modified. In particular, the pi-carrier used in the cell-to-mobile link is no longer usable. The pilot carrier must be stronger than the audio carrier to provide good phase comparison for data modulation. With a cellular station transmitting many simultaneous audio carriers, all of the audio carriers can share a single pilot signal. Therefore, the power of the pilot signal per audio carrier is quite small.

However, the mobile-to-cell link usually has only one audio carrier per mobile. If the pilot were used, it would require significantly more power than a sound carrier. This situation is clearly undesirable because the capacity of the entire system would be greatly reduced due to interference caused by a larger number of high power pilot signals. Therefore, modulation capable of efficient demodulation without a pilot signal must be used.

In a mobile-to-cell channel that is spoiled by a Rayleigh wobble leading to a rapidly changing channel phase, coherent demodulator techniques, such as the Costas loop that derives the phase from the received signal, are not usable. Other techniques, such as differential

20095859 prh 04 -04- 2011 Herentic PSKs can be used but fail to provide the desired level of signal-to-noise ratio.

Thus, a form of orthogonal signaling, such as binary, quaternary or m-system signaling, should be used. The exemplary embodiment employs 64 system orthogonal signaling technology using Walsh functions. The demodulator for the orthogonal signaling of the m system requires channel coherence only for the duration of the transmission of the m system symbol. In the exemplary embodiment, this is only for two bits.

The message coding and modulation process begins with a convolutional encoder with a forced length K = 9 and a code rate r of 1/3. At a nominal data rate of 9600 bits per second, the encoder produces 28800 binary symbols per second. These are grouped into characters containing 6 symbols each, at a rate of 4800 characters per second, with 64 possible characters. Each character is encoded into a 64-length Walsh sequence containing 64 bits or returns. The 64-system Walsh return frequency is 307,200 bits per second in the exemplary embodiment.

The Walsh pieces are then covered or multiplied by a PN sequence running at 1.2288 MHz. Each mobile unit is assigned a unique PN sequence for this purpose. This PN sequence may be assigned only for the duration of the call or permanently assigned to the mobile unit.

The indicated PN sequence is referred to herein as the user PN sequence. The user PN sequence generator runs at 1.2288 MHz clock and produces four PN chunks for each Walsh chunk.

Finally, a pair of 32768 short PN sequences are generated. In the exemplary embodiment, the same sequences are used as in the cell-to-mobile link. The Walsh chunk sequence covered by the user PN is masked or multiplied

20095859 prh 04 -04- 2011 with each of two short PN sequences. The two resulting sequences are then biphasic modulating a quadratic pair of sinusoids and summing to a single signal. The resulting signal is then bandpass filtered, converted to a final RF frequency, amplified, filtered and radiated by the antenna of the mobile unit. As described with reference to a cell-to-mobile station, the order of the filtering, gain, conversion, and modulation operations may be reversed.

In the exemplary embodiment, two different phases of the user PN code could be generated and used to modulate the two carrier phases of the quadrature waveform, eliminating the need to use 32768 length sequences. In yet another embodiment, the mobile-to-cell link could only use two-phase modulation, which also eliminates the need for short sequences.

The cell site receiver provides short PN sequences for each signal and a user PN sequence for each active received mobile signal. The receiver correlates the received signal energy with each encoded waveform in separate correlators. Each of the outputs of the correlator is then processed separately to demodulate 64 system coding and convolutional coding using a Fast Hadamard Transform processor and a Viterbi algorithm decoder.

In another alternative modulation system in a cell-to-cell link, the same modulation system would be used as in a cell-to-cell link. Each mobile station would use a pair of 32768 sector codes as an outdoor code. The code would use the 64-bit Walsh sequence assigned to the mobile station for use in that sector. Nominally, the same Walsh sequence would be assigned to the mobile station from the mobile station-

<td colspan="3"> 53</td>

<td>to cell,</td><td>please</td><td>used in the cell-</td>

<td>the mobile station.</td><td></td><td></td>

<td>Mentioned above</td><td></td><td>orthogonal PN coding</td>

<td>system limits</td><td colspan="2">modulation]</td>

20095859 prh 04 -04- 2011 the available bandwidth spread back frequency divided by 64 for maximum speed, or 19200 Hz for the numbers used in the exemplary embodiment. This would prevent the m-system coding, with a high numerical value, from being used, as described for the exemplary embodiment. Alternatively, however, the convolutional code for rate r = 3, forced length K = 9, could be used with differential binary phase keying modulation of the encoded binary symbols. In a cellular position, a demodulator could generate a phase reference value in the short term using the technique described in Andrew J. Viterbi and Audrey M. Viterbi, Nonlinear Estimation of PSK Modulated Carrier with Application to Burst Digital Transmission, Vol IT-29, Vol. 4 ,. July 1983. For example, the phase comparison value could, on average, be only 4 symbols, which does not require more channel coherence than the above 64 system.

However, the performance of the alternative system just described is inferior to that of the preferred embodiment under severe Rayleigh wobble and multipath conditions. However, in certain environments where flutter and multipath are less severe, such as satellite-to-mobile in a channel and certain terrestrial channels, the performance of an alternative system could be better than that of an inexpensive application. This can occur because the benefit of making mobile signals orthogonal to each other can overcome the loss of detection efficiency of the DPSK system.

To satisfy the time equalization requirement for orthogonal Walsh functions in the alternative mobile-to-cell link, each cellular receiver

20095859 prh 04 -04- 2011 defines a time error on the nominal timing of each received signal. If any received signal leaves in timing, then the corresponding cell modulator and transmitter sends a command to this mobile station to advance its transmission timing in small increments. Conversely, if the timing of the received signal is ahead of the nominal timing, the command to transmit is delayed by a small increment. The timing adjustments are in the order of 1/8 PN or 101.7 nanoseconds. The commands are transmitted at a relatively low speed, in the order of 10 to 50 Hz, and consist of a single bit embedded in the digital audio data stream.

During soft handoff, the mobile unit receives signals from two or more cells. Because the mobile unit can only equalize its timing in response to one of the cell timing control commands, the mobile station normally shifts its timing in response to commands received from the strongest cell being received. The signal transmitted by the mobile unit is thus time-aligned with the cell with which it has the best route. Otherwise, the result is a greater interference with other users.

If each of the cellular receivers receiving the mobile signal performs the aforementioned time error measurement and correction transmit operation, then all signals received by the mobile station will normally be received at approximately the same timing, resulting in reduced interference.

Fig. 9 is a block diagram of an exemplary CDMA telephone. The mobile unit CDMA telephone includes an antenna 430 coupled via a diplexer 432 to an analog receiver 344 and a transmission power amplifier 436. The antenna 430 and the diplexer 432 are of conventional design and allow simultaneous transmission and reception through a single antenna. an

20095859 prh 04 -04- 2011 term 430 collects the transmitted signals and outputs them through diplexer 432 to analog receiver 434. Receiver 434 receives RF frequency signals from diplexer 432, typically in the 850 MHz frequency band, for amplification and IF frequency downmixing. This conversion process is accomplished using a standard structure frequency synthesizer that allows the receiver to be tuned to any frequency within the receiving cell frequency of the whole cellular telephone band within the new band. The signals are also filtered and digitized for output to digital data receivers 540 and 542 and to applicant receiver 544.

Details of receiver 434 are further shown in Figure 10. Signals received from antenna 430 are provided to a downstream mixer 500 which includes an RF amplifier 502 and a mixer 504. The received signals are output to an RF amplifier 502, where they are amplified and output as an input to mixer 504. which is the signal output from the frequency synthesizer 506. The amplified RF signals are converted in the mixer 504 to the IF frequency by mixing with the output signal of the frequency synthesizer.

The IF signals are output from mixer 504 to bandpass filter (BPF) 508, which is typically a Surface Acoustic Wave (SAW) filter having a passband of about 1.25 MHz, where they are bandpass filtered. The properties of the SAW filter are selected to match the waveform of the signal transmitted by the cell site. The signal transmitted by the cell site is a direct sequence wide spectrum signal modulated by a PN sequence clocked at a predetermined frequency of 1.2288 MHz in the exemplary embodiment. This clock rate is selected to be a multiple of 9.6 kbps baseband data rate.

The filtered signals are output from BPF 508 as an input to a variable gain IF amplifier 510,

20095859 prh 04 -04- 2011 where the signals are again amplified. The amplified IF signals are output from the IF amplifier 510 to an analog to digital (A / D) converter 512, where the signals are digitized. The IF signal is converted to a digital signal at a clock frequency of 9.8304 MHz in the exemplary embodiment, which is exactly 8 times the PN return frequency. Although the (A / D) converter 512 is shown as part of the receiver 534, it could instead be part of the data and paging receivers. The digitized IF signals are output from the (A / D) converter 512 to the data receivers 440 and 442, and the applicant receiver 444.

The receiver 434 also performs a power control function to adjust the transmission power of the mobile unit. The automatic gain control circuit 514 (AGC) is also connected to the output of the IF amplifier 510. In response to the level of the amplified IF signal, the AGC circuit 514 provides a response signal to the gain control input of the IF amplifier 510. The receiver 434 also uses an AGC circuit 514 to generate an analog power control signal, which is provided to the transmit power control circuit 438.

9, the digitized signal output from the receiver 434 is provided to the digital data receivers 440 and 442 and the applicant receiver 444. It will be understood that a cheap, low power mobile unit may have only one data receiver, while high power units may have two or more multipath receivers.

The digitized IF signal may include signals from many ongoing calls together with pilot carriers transmitted by the co-location or all neighboring cell-sites. The purpose of the receivers 440 and 442 is to correlate the IF samples with the correct PN sequence. This correlation process yields a quantity known as processing gain, which improves the signal-to-interference ratio of a signal suitable for the correct PN sequence while not amplifying other signals. The correlation output is then observed

7

20095859 prh 04 -04- 2011 synchronously using a pilot carrier from the nearest cell station as a carrier phase reference. The result of this detection process is a sequence of encoded data signals.

One feature of the PN sequence used in the present invention is discrimination against multipath signals. When a signal arrives at a mobile station receiver after traveling more than one route, there is a difference in signal reception time. This difference in reception time corresponds to the difference in distance divided by the propagation speed. If this time difference exceeds one microsecond, then the correlation process discriminates between paths. The receiver can choose whether to follow an earlier or a later route. If two receivers, such as receivers 440 and 442, are provided, then two independent routes can be tracked and processed in parallel.

Applicant receiver 444 is guided by control process 446 for continuous time domain investigation around the nominal time of the received pilot signal of the cell station to other multipath pilot signals from the same cell station and to pilot signals transmitted by other cell stations. Receiver 444 measures the desired waveform at any reception time other than the nominal time. Receiver 444 compares the signal strength in the received signals. Receiver 444 provides a signal strength signal to control processor 44 indicating the strongest signals.

Processor 446 outputs control signals to data receivers 440 and 442, which each process a different strongest signal. Occasionally, the pilot signal transmitted by the second cell site has a higher signal strength than the current cell station signal strength. Then, the control processor 446 generates a control message to send to the system controller via the current cell site, requesting a cell transfer to the cell

20095859 prh 04 -04- 2011 to the position corresponding to the strongest pilot signal. Receivers 440 and 442 can therefore handle calls through two different cell sites.

During the soft handoff operation, the mobile unit receives signals from two or more cells. Because the mobile unit can only equalize its timing in response to one of the cell timing control commands, the mobile unit will normally shift its timing in response to the commands it receives from the strongest cell. The signal transmitted by the mobile unit is thus time-aligned with the cell with which it has the best route. Otherwise, there will be a greater interference with other users.

Further details of an exemplary receiver, such as a data receiver 440, are shown in greater detail in Figure 10. The data receiver 440 includes PN generators 516 and 518 which generate PN_Tand PN_Qsequences in a manner corresponding to those generated by the cell site. The timing and sequence control signals are provided to PN generators 516 and 518 from the control processor 446. The data receiver 440 also includes a Walsh generator 520 which provides a suitable Walsh function for communicating with this mobile unit at the cell site. The Walsh generator 520 generates, in response to timing signals from the control processor (not shown) and a function selection signal, a signal corresponding to the assigned Walsh sequence. The cellular station transmits a function selection signal to the mobile unit as part of the call construction message. PN_Zand PN_Qsequences are output from PN generators 516 and 518 as inputs to Exclusive OR ports 522 and 524, respectively. The Walsh generator 520 outputs to each of Exclusive OR ports 522 and 524, where the signals Exclusive OR are implemented and sequences PN are output._X, and PN_Q,.

20095859 prh 04 -04- 2011

Sequences PN_r and PN_Q, are provided to receiver 440, where they are supplied to PN-QPSK correlator 526. PN correlator 526 may be constructed in the same manner as PN correlators of digital station cellular receivers. PN correlator 526 correlates the received I and Q channel data ΡΝ_Σ, - and PN_Q, and provides a correlated I and Qchannel data output to the respective despreaders 528 and 530. The despreaders 528 and 530 retain the input information for one symbol over a 64-bit period. The hopper outputs are provided to a phase inverter 532, which also receives a pilot phase signal from the control processor 446. The phase of the received symbol data is inverted according to the phase of the pilot signal as determined by the applicant receiver and the control processor. The output from the phase inverter 532 is I channel data provided to the deinterleaver and decoder circuit.

The control processor 446 also includes a PN generator 534, which generates a user PN sequence in response to the mobile unit address or user ID entered. The PN sequence output from PN generator 534 is provided to the multipath combiner and decoder circuit. Since the cellular mobile station signal is modified by the mobile PN unit's address PN sequence, the output of the PN generator 534 is used to counter-edit the signal transmitted by the cell site for the purpose of this mobile station, in the same manner as the cell site receiver. In particular, the PN generator provides an output PN sequence to the de-interleaver and decoder circuit, where it is used to counter-edit the custom user data. Although the modification is described with reference to the PN sequence, it is conceivable that other modification techniques, including known modification techniques, may be used.

The outputs of the receivers 440 and 442 are provided to the multipath combiner and decoder circuit 448. The circuit 448

20095859 The multi path combiner circuit included in prh 04 -04- 2011 simply adjusts the timing of the two received symbols to equalize each other and adds them together. This addition process can be performed by multiplying two currents by a number corresponding to the relative signal strengths of the two currents. This function can be considered a maximum ratio multipath combiner. The resulting composite signal stream is then decoded using a forward sports detection (FEC) decoder, also included in circuit 448. A common digital baseband equipment is a digital vocoder system. The CDMA system is designed to accommodate different vocoder structures.

The baseband circuitry 450 typically includes a digital vocoder (not shown), which may be of variable rate type as described in the aforementioned parallel patent application. The baseband circuitry 450 further serves as an interface to a handset or any other type of terminal. The baseband circuit 450 adapts to various vocoder structures. The baseband circuit 450 provides output information signals to the user in accordance with the information provided therein by the circuit 448.

In the mobile-to-cell link, the user's analog audio signals are usually provided via a handset as input to the baseband circuitry 450. The baseband circuitry 450 includes an analog-to-digital converter (A / D) (not shown) that converts the analog signal to a digital format. The digital signal is supplied to a digital vocoder where it is encoded. The vocoder output is provided to a forward error correction (FEC) encoding circuit (not shown) for error correction. The error correction coding performed in the exemplary embodiment is a convolutional coding system. The digitized coded signal is output from the baseband circuit 450 to the transmission modulator 452.

20095859 prh 04-04- 2011

First, the transmission modulator 452 Walsh encodes the transmission data and then modulates the encoded signal to a PN carrier signal having a PN sequence selected according to the assigned address function for the call. The control processor 446 determines the PN sequence from the call setup information transmitted by the cell site and decoded by the receivers 440 and 442 and the control processor 446. Alternatively, the control processor 446 may determine the PN sequence with the cell site by preordering. Control processor 446 provides PN sequence information to transmit modulator 452 and receivers 440 and 442 for call decoding.

The output of transmit modulator 452 is provided to transmit power control circuit 438. The signal transmit power is controlled by an analog power control signal provided by receiver 434. The control bits transmitted by the cell stations in the form of a power control command are In response to this command, control processor 446 generates a digital power control signal to be provided to circuit 438. Further information on the relationship between the receivers 440 and 442, control processor 446, and transmit power control 438 is further described in the above-referenced patent application.

The transmit power control circuit 438 outputs a power-controlled modulated signal to the transmit power amplifier circuit 436. The circuit 436 amplifies and converts the IF signal to the RF frequency by mixing the frequency synthesizer with the output signal, which tunes the signal to the correct output frequency. Circuit 436 includes an amplifier that amplifies power to a final output level. The intended transmission signal is output from circuit 436 to diplexer 432. Diplexer 432 couples the signal to antenna 340 for transmission to cellular stations.

20095859 prh 04 -04- 2011

The control processor 446 is also capable of generating control messages, such as cellular multipath mode requests and cell-site communications termination commands. These commands are provided to the transmission modulator 452 for transmission. Control processor 446 responds to data received from data receivers 440 and 442 and applicant receiver 444 to make decisions about handoff and multipath combining.

With respect to transmission by the mobile unit, the analog audio signal of the mobile station user is first transmitted through a digital vocoder. The vocoder output is then sequentially convolutional forward error correction (FEC) coded, the 64 system orthogonal signal encoded and modulated to the PN carrier signal. The 64 system orthogonal signal is generated by a Walsh function encoder. The encoder is controlled by collecting six consecutive binary symbol outputs from the convolutional FEC encoder. The six binary together determine which of the 64 possible Walsh sequences will be transmitted. The Walsh sequence is 64 bits long. Thus, the Walsh bit rate must be 9600 * 3 * (1/6) * 64 - 307200 Hz for 9600 bps data rate.

The mobile-to-cell link utilizes a short PN sequence for all audio carriers in the system, while user address encoding is performed using the user's PN sequence generator. The user PN sequence is assigned exclusively to the mobile station for at least the duration of the call. User PN Sequence Exclusive OR is implemented with generic PN sequences that are 32768 inserted maximum length linear shift register sequences. Of the resulting binary signals, each biphasic modulates a transverse carrier, is summed to form a composite signal, is baseband filtered and converted to an IF frequency output. In the exemplary embodiment, a portion of the filtering process is performed thereafter

20095859 prh 04 -04- 2011 is actually a finite impulse response (FIR) digital filter that operates on a binary sequence output.

The modulator output is power controlled by signals from the digital control processor and the analog receiver, converted to RF operation by mixing with a frequency synthesizer, which tunes the signal to the correct output frequency, and then amplified to the final output level. The transmission signal is then applied to the diplexer and antenna.

Figure 11 illustrates a preferred but still exemplary embodiment of a transmission module 452 of a mobile unit. Data is provided in digital form from a user digital baseband circuit to an encoder 600, where in an exemplary embodiment it is convolutionally coded. The output of encoder 600 is provided to interleaver 602, which in the exemplary embodiment is a block interleaver. The interleaved symbols are output from the block interleaver 602 to the Walsh encoder 604 of the transmission modulator 452. Walsh encoder 604 uses the input signals to generate a code sequence output. The Walsh sequence is provided at one input of Exclusive OR port 606.

The transmission modulator 452 further includes a PN generator 608 which receives the output of the mobile unit as an input in determining the PN sequence. PN generator 608 generates a user-specific 42-bit sequence as described with reference to Figures 3 and 4. An additional feature of PN generator 608, common to all user PN generators and not previously described, is a masking technique for generating an output PN user sequence. For example, a 42-bit mask is provided to the user with each bit of the 42-bit mask being Exclusively OR implemented with the shift register set forming the PN generator with the bit output from each register. Mask and Shift Register Bit Exclusive OR Implementation Output64 Exclusive OR implementation implemented together to form

PN generator output to be used as a user PN sequence.

The output PN sequence of PN generator 608, the sequence PN ', is provided to Exclusive OR port 606. Walsh symbol data and PN_ sequence to Exclusive OR is implemented in Exclusive OR port 606 and provided as input to each Exclusive OR port 610 and 612.

Transmit modulator 452 further includes ΡΝ generators 614 and 616 which generate PN_Xand PN_Qsequences. All mobile units use the same PN_Xand PN_Qsequences. These PN sequences are exemplary zero transitions used in cell-to-mobile communication. Only owned by -OR gates. 610 and 612 give the second input PN, respectively_Xand PN_Qsequences en as output from PN generators 614 and 616. Sequences_Χ and PN_q Exclusive OR is implemented at the respective exclusive TA ports and output is provided to transmit power control 438 (Figure 9).

In the exemplary embodiment, the mobile-to-cell link uses a rate r = 1/3 convolutional code with a forced length K = 9. The code generators are G₂ = 557 (octal), G₂ = 663 (octal), G₃ = 711 (octal). As with the cell-to-mobile link, code20095859 prh 04 -04-2011 repetition is used to adapt to four different data rates produced by the vocoder on a 20 ms frame basis. Unlike the cell-to-mobile link, the repeated code symbols are not transmitted to the air at low power levels, but only one reproduction group code symbol is transmitted at the nominal power level. Finally, in the exemplary embodiment, code repetition is used merely as a means of adapting the variable data rate system to the interleaving and modulation structure, as will be described in the following paragraphs.

A block interleaver of 20 ms duration, exactly one vocoder frame, is used

20095859 prh 04 -04- 2011 in the master-cell link. The number of code symbols at 20 ms, assuming a data rate of 9600 bps and a code rate of r = 1/3, is 576. The N and B parameters, N is the number of rows in the overlap line, and B is the number of columns and 32 and 18, respectively. The code symbols are written as rows in the interleaver memory matrix and read as columns.

The modulation format is 64 system orthogonal signaling. In other words, the interleaved code symbols are grouped into groups of six to roll one of the 64 orthogonal waveforms. The 64 time orthogonal waveforms are the same Walsh functions used as overlay sequences in the cell-to-mobile link.

The data modulation time interval is 208.33 μβ and is called the Walsh symbol interval. At 9600 bps, 208.33 με corresponds to 2 information bits and 6 code symbols, respectively, at a code symbol rate of 28800 sps. The Walsh symbol interval is divided into 64 equal-length time intervals, called Walsh chunks, each of which takes 208.33 / 64 = 3.25 μβ. The Walsh frequency is thus 1 / 3.25 μβ = 307.2 kHz. Because

The PN spread rate is symmetric on two links, i.e. 1.2288 MHz, so there are 4 PN pieces per Walsh piece.

In total, three PN generators are used in the mobile-to-cell link path. A user-specific 42-bit PN generator and a pair of 15-bit I and Q channel PN generators. After the user-specific spreading operation, the QPSK signal is spread, as was done in the cell-to-mobile link. Unlike the cell-to-mobile link, where each sector or cell was identified by unique 2¹⁵length sequences, here all mobile units use the same I and Q PN sequences. These PN sequences are null transition sequences used for cell-to-mobile

20095859 prh 04 -04- 2011

link, also called pilot sequences.

Code repetition and energy scaling are used in the cell-to-mobile link to adapt to the varying speeds produced by the vocoder. The mobile-to-cell link uses a different system based on Hurst's transmission.

The vocoder produces four different data rates, i.e., 9600, 4800, 2400 and 1200 bps on a 20 ms frame basis, such as a cell-to-mobile link. The information bits are encoded by a convolutional encoder of r = 1/3 and the code symbols are repeated 2, 4, and 8 times at three lower data rates. Thus, the code symbol rate is kept constant at 28800 sps. After the encoder, the code symbols are interleaved with a block interleaver that takes exactly one vocoder frame or 20 ms. The convolutional encoder generates a total of 576 code symbols every 20 milliseconds, some of which may be repeated.

The sequence of code symbols transmitted is shown in Figure 12. Note that the vocoder frame, 20 ms, is subdivided into 16 slots, each of which takes 1.25 ms. The numerology of the cell-to-cell link is such that each slot has 36 code symbols at 28800 sps or 6 Walsh symbols at 4800 sps respectively. At 1/2 speed, i.e., 4800 bps, the slots are grouped into 8 groups, each with 2 slots. At 1/4 speed, i.e. 2400 bps, the slots are grouped into 4 groups each having 4 slots, and finally at 1/8 speed, i.e. 2400 bps, slots are grouped into 2 groups each comprising 8 slots.

Figure 12 illustrates an exemplary symbol burst transmission scheme. For example, at a rate of 1/4, i.e. 2400 bps, during the fourth slot of the first group, the fourth and eighth rows of the interleaver memory matrix are read by columns and transmitted sequentially.

20095859 prh 04 -04- 2011

The slot position for the data to be transmitted must be randomized to reduce interference.

The timing of the mobile-to-cell link is shown in Figure 13. Figure 13 expands over the timing diagram of Figure 7 to include mobile-to-cell channels, voice and input channels. Here are the steps to synchronize a cell-to-cell link:

1. Successfully decoding synchronization message, CRC check;

2. Loading the long PN shift register in the space received with the synchronization message; and

3. The pilot code offset is compensated if received from a sector using the migrated pilot.

In this case, the mobile station has full synchronization, i.e. PN synchronization and real time synchronization, and can start transmitting on either the input channel or the audio channel.

In order to initiate a call, the mobile unit must be equipped with signaling capabilities in order to make a call to another user of the system via the cell site. The revenue technique conceived in the mobile-to-cell link is the slit-ALOHA. An exemplary transmission bit rate on the reverse channel is 4800 bps. The inbound channel package includes an introduction followed by information.

In the exemplary embodiment, the preamble length is an integer multiple of a 20 ms frame and is a sector / cell parameter received by the mobile station in one of the paging channel messages. Because cellular receivers use preambles to solve propagation delays, this system allows the preamble length to vary based on the cell radius. The PN code of the users in the inbound channel is either pre-arranged or transmitted to the mobile units in the paging channel.

20095859 prh 04-04-2011

The modulation is fixed and constant over the duration of the introduction. The orthogonal waveform used in the introduction is W_o, that is, any zero Walsh function. It should be noted that a string containing only zeros at the input of the convolutional encoder produces the desired waveform W_o.

A paging channel data packet can consist of one or two frames of 20 msec. Input channel coding, interleaving and modulation is exactly the same as for an audio channel at 9600 bps. In the exemplary embodiment, the sector / cell requires a 40 msec preamble of the mobile units and the input channel message type requires a single data frame. Let N be_p the number of preamble frames, where k is the number of 20 msec used from a predetermined time start point. Then, the mobile stations are allowed to start transmitting on the paging channel only when the equation: (k, N_p+2) = 0 is realized.

For other communication applications, it may be desirable to rearrange the various elements of error correction coding, orthogonal sequence coding, and PN coding to better suit the application.

For example, in satellite mobile communication, where the signals are relayed between large hub stations and mobile terminals by one or more earth orbiting satellites, it may be desirable to use coherent modulation and demodulation techniques in each direction of the link, since the channel is much more phase-coherent than terrestrial. In such an embodiment, the mobile station modulator would not use m-system coding as described above. Instead, two-phase or four-phase modulation of the forward error correction symbols could be used with conventional coherent demodulation, with the carrier phase removed from the received signal using Costas loop technique. In addition, orthogonal Walsh function multiplexing may be used, as described e.g.

20095859 prh 04 -04- 2011 to the mobile-to-mobile link. As long as the channel phase remains reasonably coherent, this modulation and demodulation system allows operation at a lower Eb / No ratio than m-system orthogonal signaling, resulting in greater system capacity.

In another embodiment, it may be advantageous to encode the speech waveform directly into the RF waveform instead of using the vocoder and FEC technique. When the use of vocoder and FEC technology results in high link efficiency, the complexity of its implementation is high resulting in additional cost and high power consumption. These disadvantages can be particularly disadvantageous in a portable pocket phone where battery consumption and cost are important. In the conventional digital telephone transmission, the speech waveform is represented in the digital format as 8 bit speech samples at a sampling frequency of 8 kHz. The CDMA system could encode 8-bit images directly to carrier phase angles. This would eliminate the need for a vocoder or an FEC encoder / decoder. It would also require a somewhat higher signal-to-noise ratio for good operation, resulting in lower capacity. In one alternative, 8-bit speech samples could be directly encoded into carrier amplitudes. In yet another alternative, speech waveforms could be encoded into carrier phases and amplitudes.

The foregoing description of preferred embodiments is intended to enable any person skilled in the art to make or use this invention. Various modifications to these applications will be apparent to those skilled in the art, and the general principles defined herein may be applied to other applications without the use of the inventive ability. Therefore, the present invention is not intended to be limited to the applications disclosed herein, but is to be understood in the broadest sense in accordance with the principles and novel features contained herein.