Automatic frequency control for a wireless communication system with multiple subcarriers

The present application claims priority to provisional U.S. Application Ser. No. 60/660,914, entitled “Automatic Frequency Controller,” filed Mar. 11, 2005, assigned to the assignee hereof and incorporated herein by reference.

BACKGROUND

I. Field

The present disclosure relates generally to communication, and more specifically to automatic frequency control (AFC) for wireless communication.

II. Background

In wireless communication, a transmitter modulates data onto a radio frequency (RF) carrier signal to generate an RF modulated signal that is more suitable for transmission. The transmitter then transmits the RF modulated signal via a wireless channel to a receiver. The transmitted signal may reach the receiver via one or more signal paths, which may include a line-of-sight path and/or reflected paths. The characteristics of the wireless channel may vary over time due to various phenomena such as fading and multipath. Consequently, the transmitted signal may experience different channel conditions and may be received with different amplitudes and/or phases over time.

The receiver receives the transmitted signal, downconverts the received signal with a local oscillator (LO) signal, and processes the downconverted signal to recover the data sent by the transmitter. The receiver typically performs frequency control (e.g., frequency acquisition and tracking) to estimate the frequency error in the LO signal and to correct this frequency error. This frequency error may be due to various factors such as receiver circuit component tolerances, temperature variations, and Doppler effect due to movement by the receiver and/or transmitter. The frequency control may be challenging if the requirements on frequency accuracy are stringent.

There is therefore a need in the art for techniques to expeditiously and reliably perform frequency control for wireless communication.

SUMMARY

Techniques for performing frequency control in a wireless communication system with multiple subcarriers are described herein. The multiple subcarriers may be obtained with Orthogonal Frequency Division Multiplexing (OFDM), Single-Carrier Frequency Division Multiple Access (SC-FDMA), or some other modulation technique.

In one aspect, techniques are described for performing frequency control in a system that transmits a pilot along with OFDM symbols. Frequency acquisition is performed based on a received pilot, which may be time division multiplexed with the OFDM symbols. Frequency tracking is performed based on received OFDM symbols. For frequency acquisition, an initial frequency error estimate may be derived based on the received pilot, and an automatic frequency control (AFC) loop may be initialized with the initial frequency error estimate. For frequency tracking, a frequency error estimate may be derived for each received OFDM symbol, and the AFC loop may be updated with the frequency error estimate. Frequency error in input samples is corrected by the AFC loop with the initial frequency error estimate as well as the frequency error estimate for each received OFDM symbol.

In another aspect, techniques are described for deriving a frequency error estimate for a received OFDM symbol. A variable number of samples of the received OFDM symbol are selected (e.g., based on the received OFDM symbol timing) for use for frequency error estimation. In an embodiment, the start of an FFT window is determined based on the timing of the received OFDM symbol. The samples to use for frequency error estimation are then selected from among the samples within the FFT window and for a cyclic prefix of the received OFDM symbol. A frequency error estimate is then derived based on the selected samples.

Various aspects and embodiments of the invention are described in further detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

The features and nature of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout.

<figref idrefs="DRAWINGS">FIG. 1</figref> shows a block diagram of a base station and a terminal.

<figref idrefs="DRAWINGS">FIG. 2</figref> shows an exemplary super-frame structure.

<figref idrefs="DRAWINGS">FIG. 3</figref> shows a block diagram of a receiver and a demodulator.

<figref idrefs="DRAWINGS">FIG. 4</figref> shows a block diagram of an AFC unit.

<figref idrefs="DRAWINGS">FIG. 5</figref> shows a block diagram of an initial frequency error estimator.

<figref idrefs="DRAWINGS">FIG. 6</figref> shows a block diagram of an arctangent unit.

<figref idrefs="DRAWINGS">FIG. 7</figref> shows OFDM symbol copies received via multiple signal paths.

<figref idrefs="DRAWINGS">FIG. 8</figref> shows a block diagram of a frequency error estimator.

<figref idrefs="DRAWINGS">FIG. 9</figref> shows a block diagram of a phase accumulator and a phase rotator.

<figref idrefs="DRAWINGS">FIG. 10</figref> shows a process for performing frequency control.

<figref idrefs="DRAWINGS">FIG. 11</figref> shows an apparatus for performing frequency control.

<figref idrefs="DRAWINGS">FIGS. 12 and 13</figref> show a process and an apparatus, respectively, for deriving a frequency error estimate.

<figref idrefs="DRAWINGS">FIGS. 14 and 15</figref> show a process and an apparatus, respectively, for determining an initial frequency error estimate.

<figref idrefs="DRAWINGS">FIGS. 16 and 17</figref> show a process and an apparatus, respectively, for performing frequency control with phase compensation for changes in circuit blocks.

DETAILED DESCRIPTION

The word “exemplary” is used herein to mean “serving as an example, instance, or illustration.” Any embodiment or design described herein as “exemplary” is not necessarily to be construed as preferred or advantageous over other embodiments or designs.

The frequency control techniques described herein may be used for various communication systems such as cellular systems, broadcast systems, wireless local area network (WLAN) systems, satellite positioning systems, and so on. The cellular systems may be Code Division Multiple Access (CDMA) systems, Time Division Multiple Access (TDMA) systems, Frequency Division Multiple Access (FDMA) systems, Orthogonal Frequency Division Multiple Access (OFDMA) systems, Single-Carrier FDMA (SC-FDMA) systems, and so on. The broadcast systems may be MediaFLO systems, Digital Video Broadcasting for Handhelds (DVB-H) systems, Integrated Services Digital Broadcasting for Terrestrial Television Broadcasting (ISDB-T) systems, and so on. The WLAN systems may be IEEE 802.11 systems, Wi-Fi systems, and so on. These various systems are known in the art.

The frequency control techniques described herein may be used for systems with a single subcarrier as well as systems with multiple subcarriers. Multiple subcarriers may be obtained with OFDM, SC-FDMA, or some other modulation technique. OFDM and SC-FDMA partition a frequency band (e.g., the system bandwidth) into multiple (K) orthogonal subcarriers, which are also called tones, bins, and so on. Each subcarrier may be modulated with data. In general, modulation symbols are sent on the subcarriers in the frequency domain with OFDM and in the time domain with SC-FDMA. OFDM is used in various systems such as MediaFLO, DVB-H and ISDB-T broadcast systems, IEEE 802.11a/g WLAN systems, and some cellular systems. For clarity, the techniques are described below for a broadcast system that uses OFDM, e.g., a MediaFLO system.

<figref idrefs="DRAWINGS">FIG. 1</figref> shows a block diagram of a base station <b>110</b> and a terminal <b>150</b> in a broadcast system <b>100</b>. For simplicity, base station <b>110</b> and terminal <b>150</b> are each equipped with a single antenna. A base station is typically a fixed station and may also be called a base transceiver system (BTS), an access point, a Node B, and so on. A terminal may be fixed or mobile and may also be called a mobile station, a user equipment, a mobile equipment, and so on. Terminal <b>150</b> may be a cellular phone, a personal digital assistant (PDA), a wireless modem, a wireless communication device, a handheld device, a subscriber unit, and so on.

At base station <b>110</b>, a transmit (TX) data processor <b>120</b> processes (e.g., encodes, interleaves, and symbol maps) traffic data and generates data symbols. A pilot processor <b>122</b> generates pilot symbols. As used herein, a data symbol is a modulation symbol for data, a pilot symbol is a modulation symbol for pilot, and a modulation symbol is a complex value for a point in a signal constellation, e.g., for PSK or QAM. A modulator <b>130</b> multiplexes the data symbols and pilot symbols, performs OFDM modulation on the multiplexed data and pilot symbols, and generates OFDM symbols. A transmitter (TMTR) <b>132</b> processes (e.g., converts to analog, amplifies, filters, and frequency upconverts) the OFDM symbols and generates an RF modulated signal, which is transmitted via an antenna <b>134</b>.

At terminal <b>150</b>, an antenna <b>152</b> receives the RF modulated signal from base station <b>110</b> and provides a received RF signal to a receiver (RCVR) <b>160</b>. Receiver <b>160</b> conditions (e.g., filters, amplifies, frequency downconverts, and digitizes) the received RF signal and provides received samples. A demodulator <b>170</b> performs OFDM demodulation on the received samples and provides data symbol estimates, which are estimates of the data symbols sent by base station <b>110</b>. A receive (RX) data processor <b>172</b> processes (e.g., symbol demaps, deinterleaves, and decodes) the data symbol estimates and provides decoded data. In general, the processing at terminal <b>150</b> is complementary to the processing at base station <b>110</b>.

Controllers/processors <b>140</b> and <b>180</b> direct the operation of various processing units at base station <b>110</b> and terminal <b>150</b>, respectively. Memories <b>142</b> and <b>182</b> store program codes and data for base station <b>110</b> and terminal <b>150</b>, respectively.

<figref idrefs="DRAWINGS">FIG. 2</figref> shows an exemplary super-frame structure <b>200</b> that may be used for broadcast system <b>100</b>. In the embodiment shown in <figref idrefs="DRAWINGS">FIG. 2</figref>, the transmission timeline is partitioned into super-frames, with each super-frame having a particular time duration, e.g., approximately one second. Each super-frame includes a field <b>212</b> for a time division multiplexed (TDM) pilot, a field <b>214</b> for overhead/control information, and a field <b>216</b> with N frames for traffic data, where N≧1. A super-frame may also include different and/or additional fields not shown in <figref idrefs="DRAWINGS">FIG. 2</figref>.

In the embodiment shown in <figref idrefs="DRAWINGS">FIG. 2</figref>, the TDM pilot is composed of S identical pilot sequences, with each pilot sequence containing L time-domain samples, where S&gt;1 and L&gt;1. The TDM pilot may be generated by (1) performing an L-point inverse fast Fourier transform (IFFT) on L pilot symbols to obtain a pilot sequence with L time-domain samples and (2) repeating the pilot sequence S times. The TDM pilot may be used for signal detection, frame synchronization, initial frequency error estimation, coarse time synchronization and/or other purposes.

The overhead information may convey the identity of a base station transmitting the overhead information, where and how data channels are sent in the frames of a super-frame, and/or other information. The data channels are sent in the N frames and at frequency and time locations indicated by the overhead information. Each data channel may carry any type of data such as video, audio, tele-text, data, video/audio clips, and so on. Terminal <b>150</b> may be interested in receiving one or more specific data channels from base station <b>110</b>. Terminal <b>150</b> may ascertain where each desired data channel is sent, e.g., based on the overhead information and/or the data sent on the data channel. Terminal <b>150</b> may go to sleep much of the time to conserve battery power and may wake up periodically to receive the desired data channel(s).

Each frame carries multiple (M) OFDM symbols. An OFDM symbol may be generated by (1) performing a K-point IFFT on K modulation symbols to obtain K time-domain samples for a data portion of the OFDM symbol and (2) copying the last C samples of the data portion to form a cyclic prefix for the OFDM symbol. The data portion is also referred to as a useful portion, a transformed symbol, and so on. Windowing/filtering may also be performed on the cyclic prefix and the data portion. An OFDM symbol may contain K+C samples without windowing or possibly more than K+C samples with windowing.

In an embodiment, K=4096, C=512, and each OFDM symbol contains 4608 time-domain samples prior to windowing. In an embodiment, L=128, S=36, and the TDM pilot contains 36 identical pilot sequences of length 128. Other values may also be used for K, C, L and S.

<figref idrefs="DRAWINGS">FIG. 2</figref> shows a specific super-frame structure. The frequency control techniques described herein may be used for other frame and super-frame structures.

<figref idrefs="DRAWINGS">FIG. 3</figref> shows a block diagram of an embodiment of receiver <b>160</b> and demodulator <b>170</b> at terminal <b>150</b>. Within receiver <b>160</b>, a low noise amplifier (LNA) <b>312</b> amplifies the received RF signal and provides an amplified RF signal. A filter <b>312</b> filters the amplified RF signal to pass signal components in the band of interest and to remove out-of-band noise and undesired signals. A downconverter <b>314</b>frequency downconverts the filtered RF signal with an LO signal from an LO generator <b>324</b> and provides a downconverted signal. The frequency of the LO signal is selected such that the signal component in an RF channel of interest is downconverted to baseband or near-baseband. A lowpass filter <b>316</b> filters the downconverted signal to pass the signal components in the RF channel of interest and to remove noise and undesired signals. An amplifier <b>316</b> amplifies the filtered baseband signal and provides an output baseband signal. An analog-to-digital converter (ADC) <b>318</b> digitizes the output baseband signal and provides received samples r(k) to demodulator <b>170</b>. The received samples are typically complex-value samples having inphase (I) and quadrature (Q) components.

A reference oscillator (Ref Osc) <b>322</b> generates a reference signal having a precise frequency fref. Reference oscillator <b>322</b> may be a voltage controlled crystal oscillator (VCXO), a temperature compensated crystal oscillator (TCXO), a voltage controlled TCXO (VC-TCXO), a voltage controlled oscillator (VCO), or some other type of oscillator. LO generator <b>324</b> receives the reference signal and generates the LO signal at the desired RF frequency. A clock generator <b>326</b> also receives the reference signal and generates a sampling clock for ADC <b>318</b>. LO generator <b>324</b> and clock generator <b>326</b> may each be implemented with VCOs, phase locked loops (PLLs), dividers, and so on, as is known in the art.

<figref idrefs="DRAWINGS">FIG. 3</figref> shows a specific design for receiver <b>160</b>. In general, a receiver may implement a super-heterodyne architecture or a direct-to-baseband architecture. In the super-heterodyne architecture, the received RF signal is downconverted in multiple stages, e.g., from RF to an intermediate frequency (IF) in one stage, and then from IF to baseband in another stage. In the direct-to-baseband architecture, which is shown in <figref idrefs="DRAWINGS">FIG. 3</figref>, the received RF signal is downconverted from RF directly to baseband in one stage. The super-heterodyne and direct-to-baseband architectures may use different circuit blocks and different LO frequencies. A receiver may also perform signal conditioning with one or more stages of amplifier, filter, mixer, and so on. A receiver may include different and/or additional circuit blocks not shown in <figref idrefs="DRAWINGS">FIG. 3</figref>.

Within demodulator <b>170</b>, an automatic gain control (AGC) unit <b>330</b> multiplies the received samples r(k) with a variable gain Gagc and provides input samples x(k) having the desired magnitude. An AFC unit <b>340</b> estimates frequency error in the input samples, removes the estimated frequency error from the input samples, and provides output samples y(k) having the estimated frequency error removed. A fast Fourier transform (FFT) unit <b>350</b> performs a K-point FFT on K output samples for each received OFDM symbol and obtains K frequency-domain received symbols for the K subcarriers. FFT unit <b>350</b> provides received symbols for traffic data to a data detector <b>352</b> and provides received symbols for pilot to a channel estimator <b>354</b>. Channel estimator <b>354</b> derives channel estimates for the wireless channel between base station <b>110</b> and terminal <b>150</b> based on the received symbols for pilot. Data detector <b>352</b> performs data detection (e.g., equalization or matched filtering) on the received symbols for traffic data with the channel estimates and provides data symbol estimates

An AGC controller <b>332</b> determines the magnitude of the received samples r(k) and provides the variable gain Gagc used by AGC unit <b>330</b> to obtain the desired magnitude for the input samples x(k). AGC controller <b>332</b> also provides one or more gain control signals to one or more circuit blocks (e.g., LNA <b>312</b>, downconverter <b>314</b> and/or amplifier <b>316</b>) within receiver <b>160</b>. The gain control signal(s) maintain the magnitude of the received samples r(k) within a suitable range. An AFC controller <b>342</b> receives the output of AFC unit <b>340</b> and generates a frequency control signal for reference oscillator <b>322</b>. A time tracking unit <b>344</b> detects for the start of a super-frame (e.g., based on the TDM pilot) and also determines the start of each received OFDM symbol.

<figref idrefs="DRAWINGS">FIG. 4</figref> shows a block diagram of an embodiment of AFC unit <b>340</b> within demodulator <b>170</b> in <figref idrefs="DRAWINGS">FIG. 3</figref>. Within AFC unit <b>340</b>, the input samples x(k) are provided to a sample buffer <b>408</b>, a phase rotator <b>410</b>, and an initial frequency error estimator <b>420</b>. Sample buffer <b>408</b> stores the input samples and/or output samples for subsequent processing. Estimator <b>420</b> derives an initial frequency error estimate Δ{circumflex over (f)}_init(e.g., based on the TDM pilot) whenever directed and provides the initial frequency error estimate to one input of a multiplexer (Mux) <b>446</b>. Phase rotator <b>410</b> rotates each input sample x(k) by a phase value θ_kfrom a phase accumulator <b>412</b> and provides a phase-rotated output sample y(k). The output samples have much of the frequency error removed once frequency lock is achieved. A frequency error estimator <b>430</b> derives frequency error estimates Δ{circumflex over (f)}_m, e.g., based on received OFDM symbols. The frequency error estimates are indicative of the residual frequency error in the output samples. A frequency lock detector <b>432</b> determines whether frequency lock is achieved.

A loop filter <b>440</b> filters the frequency error estimates Δ{circumflex over (f)}_mand provides an average frequency error Δ{circumflex over (f)}_err, which is indicative of the frequency error in the input samples. Within loop filter <b>440</b>, a multiplier <b>442</b> multiplies the frequency error estimates Δ{circumflex over (f)}_mwith a loop gain α. A summer <b>444</b> sums the output of multiplier <b>442</b> with the output of a frequency register <b>448</b>. Multiplexer <b>446</b> receives the output of summer <b>444</b> at another input and provides either the output of summer <b>444</b> or the initial frequency error estimate Δ{circumflex over (f)}_init. Frequency register <b>448</b> stores the output of multiplexer <b>446</b> and provides the average frequency error Δ{circumflex over (f)}_err. Phase accumulator <b>412</b> accumulates the average frequency error in each sample period and provides the phase value for each input sample.

Phase rotator <b>410</b>, frequency error estimator <b>430</b>, loop filter <b>440</b>, and phase accumulator <b>412</b> form an AFC loop that estimates and corrects frequency error in the input samples. In an embodiment, the AFC loop operates as follows. When the terminal first wakes up or first tunes to the broadcast system, estimator <b>420</b> derives an initial frequency error estimate Δ{circumflex over (f)}_initthat captures much of the frequency error between the base station and the terminal. Frequency register <b>448</b> stores the initial frequency error estimate. Phase accumulator <b>412</b> computes the phase shift in each sample period due to the frequency error from register <b>448</b>. Phase rotator <b>410</b> rotates each input sample by the phase shift from phase accumulator <b>412</b>. Thereafter, for each received OFDM symbol, estimator <b>430</b> derives a frequency error estimate Δ{circumflex over (f)}_mbased on the output samples for that OFDM symbol. The frequency error estimate Δ{circumflex over (f)}_mis scaled by the loop gain α and accumulated by frequency register <b>448</b> via summer <b>444</b> and multiplexer <b>446</b>. Hence, frequency register <b>448</b> and the AFC loop are initialized with the initial frequency error estimate and are thereafter updated by the frequency error estimate from each received OFDM symbol.

In the embodiment described above, phase rotation is performed on each input sample, and the AFC loop is updated in each OFDM symbol period. The AFC loop may also be updated at other rates. In general, the AFC loop may be updated whenever a frequency error estimate is available. For example, the AFC loop may be updated after receiving an OFDM symbol, after receiving a burst of data, at the end of a frame, and so on. The AFC loop may also be operated in different modes, e.g., an acquisition mode and a tracking mode, as described below.

The input samples for the broadcast system may be expressed as: <br /><i>x</i>(<i>k</i>)=<i>s</i>(<i>k</i>)·<i>e</i>^{j2π·Δf·k·T}^{<sub2>s</sub2>}^+φ<i>+n</i>(<i>k</i>),  Eq (1)<br /> where s(k) is a sample transmitted in sample period k, x(k) is an input sample for sample period k, n(k) is the noise for input sample x(k), Δf is a frequency error, φ is an arbitrary phase, and T_sis one sample period.

The TDM pilot contains S identical pilot sequences, as shown in <figref idrefs="DRAWINGS">FIG. 2</figref>. Hence, the transmitted samples are periodic during the TDM pilot, and s(k)=s(k+L). In this case, a correlation on the input samples may be expressed as: <br /><i>x</i>*(<i>k</i>)·<i>x</i>(<i>k+L</i>)=|<i>s</i>(<i>k</i>)|²<i>·e</i>^{j2π·Δf·L·T}^{<sub2>s</sub2>}<i>+ñ</i>(<i>k</i>),  Eq (2)<br /> where ñ(k) is the post-processed noise. Equation (2) indicates that the frequency error Δf may be isolated by correlating input sample x(k) with delayed input sample x(k+L).

A delayed correlation may be performed for each pilot sequence as follows:

<maths id="MATH-US-00001" num="00001"><math overflow="scroll"><mtable><mtr><mtd><mrow><mrow><msub><mi>C</mi><mi>l</mi></msub><mo>=</mo><mrow><munderover><mo>∑</mo><mrow><mi>i</mi><mo>=</mo><mn>1</mn></mrow><mi>L</mi></munderover><mo>⁢</mo><mrow><mrow><msubsup><mi>x</mi><mi>l</mi><mo>*</mo></msubsup><mo>⁡</mo><mrow><mo>(</mo><mi>i</mi><mo>)</mo></mrow></mrow><mo>·</mo><mrow><msub><mi>x</mi><mi>l</mi></msub><mo>⁡</mo><mrow><mo>(</mo><mrow><mi>i</mi><mo>+</mo><mi>L</mi></mrow><mo>)</mo></mrow></mrow></mrow></mrow></mrow><mo>,</mo></mrow></mtd><mtd><mrow><mi>Eq</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mn>3</mn><mo>)</mo></mrow></mrow></mtd></mtr></mtable></math></maths><br /> where x_l(i)=x(i+l·L+k_s) is the i-th input sample for the l-th pilot sequence, <ul><li id="ul0001-0001" num="0000"><ul><li id="ul0002-0001" num="0053">k_sis the sample index for the start of the first pilot sequence, and</li><li id="ul0002-0002" num="0054">C_lis the correlation result for the l-th pilot sequence.</li></ul></li></ul>

The correlation results for multiple pilot sequences may be accumulated, as follows:

<maths id="MATH-US-00002" num="00002"><math overflow="scroll"><mtable><mtr><mtd><mrow><mrow><msub><mi>C</mi><mi>init</mi></msub><mo>=</mo><mrow><munderover><mo>∑</mo><mrow><mi>l</mi><mo>=</mo><mn>1</mn></mrow><msup><mi>S</mi><mi>′</mi></msup></munderover><mo>⁢</mo><msub><mi>C</mi><mi>l</mi></msub></mrow></mrow><mo>,</mo></mrow></mtd><mtd><mrow><mi>Eq</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mn>4</mn><mo>)</mo></mrow></mrow></mtd></mtr></mtable></math></maths><br /> where S′ is the number of delayed correlations performed, which is S′&lt;S, and <ul><li id="ul0003-0001" num="0000"><ul><li id="ul0004-0001" num="0057">C_init=W₁+jW_Qis the accumulated correlation result for all pilot sequences. <br /> Equation (4) performs coherent accumulation on the S′ correlation results and provides C_init, which is a complex value having an inphase component W_Iand a quadrature component W_Q. </li></ul></li></ul>

An initial frequency error estimate may be derived based on the accumulated correlation result, as follows:

<maths id="MATH-US-00003" num="00003"><math overflow="scroll"><mtable><mtr><mtd><mrow><mrow><mrow><mi>Δ</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><msub><mover><mi>f</mi><mo>^</mo></mover><mi>init</mi></msub></mrow><mo>=</mo><mrow><mfrac><mn>1</mn><msub><mi>G</mi><mi>L</mi></msub></mfrac><mo>⁢</mo><mi>arc</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><mrow><mi>tan</mi><mo>⁡</mo><mrow><mo>[</mo><mfrac><msub><mi>W</mi><mi>Q</mi></msub><msub><mi>W</mi><mi>I</mi></msub></mfrac><mo>]</mo></mrow></mrow></mrow></mrow><mo>,</mo></mrow></mtd><mtd><mrow><mi>Eq</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mn>5</mn><mo>)</mo></mrow></mrow></mtd></mtr></mtable></math></maths><br /> where G_Lis a detector gain, which is G_L=2π·L·T_s.

The start of the first pilot sequence may be ascertained by performing a sliding correlation on the input samples and detecting for a peak in the sliding correlation. The input samples may be buffered in sample buffer <b>408</b>, and the delayed correlation in equation (3) may be performed for all pilot sequences after the TDM pilot has been detected. Alternatively, the TDM pilot may be detected using some of the pilot sequences, and the initial frequency error estimate may be derived using the remaining pilot sequences.

<figref idrefs="DRAWINGS">FIG. 5</figref> shows a block diagram of an embodiment of initial frequency error estimator <b>420</b> in <figref idrefs="DRAWINGS">FIG. 4</figref>. In this embodiment, a delayed correlator <b>510</b> receives the input samples x(k) and performs the delayed correlation shown in equation (3). Within delayed correlator <b>510</b>, the input samples are provided to an L-sample delay line <b>512</b> and a multiplier <b>516</b>. Delay line <b>512</b> delays each input sample by L sample periods, which is the length of the pilot sequence. A unit <b>514</b> provides the complex conjugate of each delayed sample from delay line <b>512</b>. Multiplier <b>516</b> multiplies each input sample with the corresponding output from unit <b>514</b> and provides the product x_l*(i)·x_l(i+L) in each sample period. A peak detector <b>520</b>, which may be part of time tracking unit <b>344</b> in <figref idrefs="DRAWINGS">FIG. 3</figref>, detects for the TDM pilot and provides the sample index k_sfor the start of the first pilot sequence. An accumulator <b>518</b> accumulates the output of multiplier <b>516</b> over L sample periods for one pilot sequence and provides the correlation result C_lfor each pilot sequence.

An accumulator <b>530</b>, which is formed with a summer <b>532</b> and a register <b>534</b>, accumulates the correlation results from delayed correlator <b>510</b> for all pilot sequences and provides the accumulated result C_init. An arctan unit <b>540</b> computes the arctangent of C_init. A scaling unit <b>542</b> scales the output of arctan unit <b>540</b> and provides the initial frequency error estimate Δ{circumflex over (f)}_init.

In an embodiment, the arctangent in equation (5) is computed using two look-up tables. Once look-up table is used to efficiently compute the ratio W_Q/W_Iin equation (5), and another look-up table is used to compute the arctangent.

<figref idrefs="DRAWINGS">FIG. 6</figref> shows a block diagram of an embodiment of arctan unit <b>540</b> in <figref idrefs="DRAWINGS">FIG. 5</figref>. Within arctan unit <b>540</b>, a unit <b>612</b><i>a </i>receives the accumulated correlation result C_initand provides the real part, which is W_I=Re {C_init}. A unit <b>612</b><i>b </i>also receives C_initand provides the imaginary part, which is W_Q=Im {C_init}. A sign detector <b>614</b> detects for the sign of the ratio W_Q/W_Iand generates a Sign bit, as follows:

<maths id="MATH-US-00004" num="00004"><math overflow="scroll"><mtable><mtr><mtd><mrow><mi>Sign</mi><mo>=</mo><mrow><mo>{</mo><mtable><mtr><mtd><mn>1</mn></mtd><mtd><mrow><mrow><mi>if</mi><mo>⁢</mo><mstyle><mspace width="0.6em" height="0.6ex" /></mstyle><mo>⁢</mo><mrow><mo>{</mo><mrow><mrow><mo>(</mo><mrow><msub><mi>W</mi><mi>I</mi></msub><mo>&lt;</mo><mn>0</mn></mrow><mo>)</mo></mrow><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mi>AND</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mrow><msub><mi>W</mi><mi>Q</mi></msub><mo>&gt;</mo><mn>0</mn></mrow><mo>)</mo></mrow></mrow><mo>}</mo></mrow><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mi>OR</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>{</mo><mrow><mrow><mo>(</mo><mrow><msub><mi>W</mi><mi>I</mi></msub><mo>&gt;</mo><mn>0</mn></mrow><mo>)</mo></mrow><mo>⁢</mo><mstyle><mspace width="1.1em" height="1.1ex" /></mstyle><mo>⁢</mo><mi>AND</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mrow><msub><mi>W</mi><mi>Q</mi></msub><mo>&lt;</mo><mn>0</mn></mrow><mo>)</mo></mrow></mrow><mo>}</mo></mrow></mrow><mo>,</mo></mrow></mtd></mtr><mtr><mtd><mn>0</mn></mtd><mtd><mrow><mi>otherwise</mi><mo>.</mo></mrow></mtd></mtr></mtable></mrow></mrow></mtd><mtd><mrow><mi>Eq</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mn>6</mn><mo>)</mo></mrow></mrow></mtd></mtr></mtable></math></maths><br /> The Sign bit indicates whether or not to invert the output depending on the quadrant within which C_initfalls.

A unit <b>616</b><i>a </i>receives the real part W_Iand provides the magnitude of W_I, which is V₁=Abs {W_I}, where Abs { } denotes the absolute of the quantity within { }. A unit <b>616</b><i>b </i>receives the imaginary part W_Qand provides the magnitude of W_Q, which is V_Q=Abs {W_Q}. A mapper <b>618</b> maps V_Iand V_Qto a numerator N and a denominator D, as follows: <br />If (<i>V</i>_I<i>≧V</i>_Q) then set <i>N=V</i>_Q<i>, D=V</i>_I, and Flip=0; else set <i>N=V</i>_I<i>, D=V</i>_Q, and Flip=1.  Eq (7)<br /> The mapping in equation (7) moves the larger of V_Iand V_Qto the denominator, which results in the ratio N/D being less than or equal to 1.0, or (N/D)≦1.0. The arctangent of N/D is then limited to a range of 0 to 45°, which allows for use of a smaller arctan look-up table.

A normalize unit <b>620</b> shifts the denominator D to the right so that the most significant bit (MSB) is ‘1’ and provides a normalized denominator D′. Unit <b>620</b> also shifts the numerator N by the same number of bits as the denominator and provides a normalized numerator N′. An inverse look-up table (LUT) <b>622</b> receives D′ and provides 1/D′. A multiplier <b>624</b> multiplies N′ with 1/D′ and provides the ratio N′/D′.

An arctan look-up table <b>626</b> receives the ratio N′/D′ and provides the arctangent of N′/ID′, orθ=arctan(N′/D′), where 0°≦θ≦45° due to the conditioning described above. A multiplexer (MUX) <b>630</b> provides θ if the Flip bit indicates that V_Iand V_Qhave not been flipped by mapper <b>618</b> and provides 90°-θ, which is generated by a unit <b>628</b>, if V_Iand V_Qhave been flipped. An inverter <b>632</b> inverts the output of multiplexer <b>630</b>. A multiplexer <b>634</b> provides the output of multiplexer <b>630</b> as the detected phase θ_initif the Sign bit indicates no inversion and provides the output of inverter <b>632</b> otherwise.

<figref idrefs="DRAWINGS">FIG. 6</figref> shows an embodiment for efficiently computing arctangent for the initial frequency error estimate. The arctangent may also be computed in other manners.

The terminal may receive the RF modulated signal from the base station via one or more signal paths. For each OFDM symbol sent by the base station, the terminal obtains a copy of the OFDM symbol via each signal path. Each OFDM symbol copy is scaled by the complex gain for the associated signal path and is further delayed by the propagation delay for that signal path.

<figref idrefs="DRAWINGS">FIG. 7</figref> shows reception of an OFDM symbol via a wireless channel with multiple signal paths. Each signal path may have any complex gain and any delay, both of which are determined by the channel environment. For simplicity, the gains for the signal paths are assumed to be equal. A first OFDM symbol copy is received via the first arriving path (FAP), and a last OFDM symbol copy is received via the last arriving path (LAP). The start of the last OFDM symbol copy is delayed by Ah samples from the start of the first OFDM symbol copy, where Ah is the delay spread of the wireless channel. The received OFDM symbol is a superposition of all of the OFDM symbol copies.

<figref idrefs="DRAWINGS">FIG. 7</figref> also shows proper placement of an FFT window for the received OFDM symbol. The FFT window has a width of K samples and determines which samples are used for subsequent processing. The start of the FFT window is indicated by an FFT Start pointer. In the example shown in <figref idrefs="DRAWINGS">FIG. 7</figref>, the start of the FFT window is Δ_ssamples earlier than the start of the data portion for the first OFDM symbol copy and is Δ_h+Δ_ssamples earlier than the start of the data portion for the last OFDM symbol copy.

<figref idrefs="DRAWINGS">FIG. 7</figref> also shows an ISI/ICI-free region, which is an overlapping region for the cyclic prefixes for all OFDM symbol copies. The width of the ISI/ICI-free region is dependent on the delay spread Ah and the cyclic prefix length C. If the FFT Start pointer is placed within the ISI/ICI-free region, as shown in <figref idrefs="DRAWINGS">FIG. 7</figref>, then the proper K samples are used for processing, and inter-symbol interference (ISI) and inter-carrier interference (ICI) are not encountered, which is desirable.

Time tracking unit <b>344</b> in <figref idrefs="DRAWINGS">FIG. 3</figref> attempts to maintain the FFT Start pointer within the ISI/ICI-free region for each received OFDM symbol. Unit <b>344</b> may obtain a channel impulse response estimate for the wireless channel, detect for the FAP and LAP, and adjust the FFT Start pointer to be within the ISI/ICI-free region.

As shown in <figref idrefs="DRAWINGS">FIG. 7</figref>, each OFDM symbol copy has a cyclic prefix that is identical to the last C samples of the data portion, which is labeled as “cyclic copy” in <figref idrefs="DRAWINGS">FIG. 7</figref>. A correlation window may be defined from the FFT Start pointer to the end of the cyclic prefix for the first OFDM symbol copy. The samples within the correlation window are periodic for all OFDM symbol copies, so that y(k)=y(k+K) for each sample y(k) within the correlation window.

A frequency error estimate may be computed for each received OFDM symbol based on the cyclic prefix, as follows:

<maths id="MATH-US-00005" num="00005"><math overflow="scroll"><mtable><mtr><mtd><mrow><mrow><mrow><mi>Δ</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><msub><mover><mi>f</mi><mo>^</mo></mover><mi>m</mi></msub></mrow><mo>=</mo><mrow><mi>Im</mi><mo>⁡</mo><mrow><mo>[</mo><mrow><munderover><mo>∑</mo><mrow><mi>i</mi><mo>=</mo><mn>1</mn></mrow><msup><mi>C</mi><mi>′</mi></msup></munderover><mo>⁢</mo><mrow><mrow><msubsup><mi>y</mi><mi>m</mi><mo>*</mo></msubsup><mo>⁡</mo><mrow><mo>(</mo><mi>i</mi><mo>)</mo></mrow></mrow><mo>·</mo><mrow><msub><mi>y</mi><mi>m</mi></msub><mo>⁡</mo><mrow><mo>(</mo><mrow><mi>i</mi><mo>+</mo><mi>K</mi></mrow><mo>)</mo></mrow></mrow></mrow></mrow><mo>]</mo></mrow></mrow></mrow><mo>,</mo></mrow></mtd><mtd><mrow><mi>Eq</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mn>8</mn><mo>)</mo></mrow></mrow></mtd></mtr></mtable></math></maths><br /> where y_m(i) is the i-th output sample for the m-th OFDM symbol, <ul><li id="ul0005-0001" num="0000"><ul><li id="ul0006-0001" num="0078">Δ{circumflex over (f)}_mis a frequency error estimate for the m-th OFDM symbol, and</li><li id="ul0006-0002" num="0079">C′ is the number of samples over which the delayed correlation is performed. <br /> The first output sample y_m(1) in equation (8) may be the first sample within the FFT window. The Im [ ] function provides the imaginary part of the complex value within the square bracket. The Im [ ] function in equation (8) is an approximation of the arctangent function in equation (5). This approximation is reasonably accurate when the quantity within the square bracket in equation (8) is small, which is typically the case once frequency lock is achieved. </li></ul></li></ul>

The delayed correlation in equation (8) is performed over C′ samples, where C′≦C. In general, the delayed correlation may be performed over all or a subset of the C samples for the cyclic prefix. In one embodiment, the delayed correlation is performed over all samples within the correlation window. In the embodiment shown in <figref idrefs="DRAWINGS">FIG. 7</figref>, the correlation window contains Δ_ssamples and is determined by the FFT Start pointer. In another embodiment, the delayed correlation is performed over all samples within the ISI/ICI-free region.

In yet another embodiment, the samples used for frequency error estimation are selected as follows: <br />If 1≦FFT_Start≦<i>C/</i>2, use samples <i>C/</i>2+1 to <i>C; </i><br />If <i>C/</i>2≦FFT_Start≦3<i>C/</i>4, use samples 3<i>C/</i>4+1 to <i>C</i>; and  Eq (9)<br />If 3<i>C/</i>4&lt;FFT_Start≦<i>C</i>, use no samples.<br /> In the embodiment shown in equation (9), a frequency error estimate is derived based on (1) the second half of the cyclic prefix if the FFT Start pointer falls within the first half of the cyclic prefix or (2) the last quarter of the cyclic prefix if the FFT Start pointer falls within the third quarter of the cyclic prefix. A frequency error estimate is not derived if the FFT Start pointer falls within the last quarter of the cyclic prefix.

The samples may also be selected for use for frequency error estimation based on the timing of the received OFDM symbol in other manners.

<figref idrefs="DRAWINGS">FIG. 8</figref> shows a block diagram of an embodiment of frequency error estimator <b>430</b> in <figref idrefs="DRAWINGS">FIG. 4</figref>. In this embodiment, a delayed correlator <b>810</b> receives the output samples y(k) and performs the delayed correlation shown within the square bracket in equation (8). Delayed correlator <b>810</b> includes a delay line <b>812</b>, a complex-conjugate unit <b>814</b>, a multiplier <b>816</b>, and an accumulator <b>818</b> that operate in similar manner as units <b>512</b>, <b>514</b>, <b>516</b> and <b>518</b>, respectively, within delayed correlator <b>510</b> in <figref idrefs="DRAWINGS">FIG. 5</figref>. However, delay line <b>812</b> delays each output sample by K sample periods, which constitute the length of the data portion. Accumulator <b>818</b> accumulates the output of multiplier <b>816</b> over C′ sample periods for the cyclic prefix and provides a correlation result Cm for each received OFDM symbol. C′ may be dependent on the timing of the received OFDM symbol and may be different for different OFDM symbols. A unit <b>820</b> provides the imaginary part of the correlation result C_mas the frequency error estimate Δ{circumflex over (f)}_m.

<figref idrefs="DRAWINGS">FIGS. 5 and 8</figref> show exemplary embodiments of frequency error estimators <b>420</b> and <b>430</b>, respectively. The embodiment in <figref idrefs="DRAWINGS">FIG. 5</figref> relies on the periodic nature of the TDM pilot to derive the initial frequency error estimate. The embodiment in <figref idrefs="DRAWINGS">FIG. 8</figref> relies on the periodic nature of the cyclic prefix in each OFDM symbol to derive a frequency error estimate. In general, frequency error estimation may be performed in various manners depending on the structure of the transmitted signal, the radio technology used for the transmitted signal, and/or other factors.

<figref idrefs="DRAWINGS">FIG. 9</figref> shows a block diagram of an embodiment of phase rotator <b>410</b> and phase accumulator <b>412</b> in <figref idrefs="DRAWINGS">FIG. 4</figref>. Within phase accumulator <b>412</b>, a summer <b>912</b> receives the current frequency error estimate Δ{circumflex over (f)}_errfrom frequency register <b>448</b> in <figref idrefs="DRAWINGS">FIG. 4</figref>, a phase adjustment θ_agcfrom AGC controller <b>332</b> in <figref idrefs="DRAWINGS">FIG. 3</figref>, and the output of a phase register <b>914</b>. Summer <b>912</b> sums all three inputs and provides the result to phase register <b>914</b>. Phase register <b>914</b> is updated by the output of summer <b>912</b> in each sample period. The phase value provided by phase register <b>914</b> in each sample period may be given as θ_k=−2π·k·Δ{circumflex over (f)}_err, which assumes that θ_agc=0.

Referring back to <figref idrefs="DRAWINGS">FIG. 3</figref>, the circuit blocks within receiver <b>160</b> may operate in one of multiple gain states. Each gain state may be associated with a specific set of gain settings for the circuit blocks in order to achieve the desired signal level at the input of ADC <b>318</b>. Different gain settings may be associated with different phases. AGC controller <b>332</b> may store a table of phases for different gain states. Whenever a switch in gain state is made, AGC controller <b>332</b> may determine the difference between the phase of the new gain state and the phase of the prior gain state. AGC controller <b>332</b> may then provide to AFC unit <b>340</b> the negative of this phase difference as the phase adjustment θ_agc. Referring again to <figref idrefs="DRAWINGS">FIG. 9</figref>, the phase adjustment θ_agcis added once to phase register <b>914</b>. Hence, θ_agcis zero except whenever a change in gain state occurs. This phase adjustment results in the output samples y(k) having approximately continuous phase when switching between gain states associated with different phases.

Within phase rotator <b>410</b>, a cos/sin look-up table <b>922</b> receives the phase value θ_kfrom phase accumulator <b>412</b> and provides the cosine and sine of θ_k. A complex multiplier <b>924</b> multiplies each input sample x(k) with the sine and cosine and provides a phase-rotated output sample y(k), which may be given as: <br /><i>y</i>_I(<i>k</i>)+<i>j y</i>_Q(<i>k</i>)=[<i>x</i>_I(<i>k</i>)+<i>j x</i>_Q(<i>k</i>)]·[cos θ_k<i>+j </i>sin θ_k],  Eq (10)<br /> where x(k)=x, (k)+j x_Q(k) is a complex-valued input sample for sample period k, and <ul><li id="ul0007-0001" num="0000"><ul><li id="ul0008-0001" num="0088">y(k)=y_I(k)+j y_Q(k) is a complex-valued output sample for sample period k. <br /> Complex multiplier <b>924</b> may be implemented with four real multiplications and two real additions. </li></ul></li></ul>

Referring back to <figref idrefs="DRAWINGS">FIG. 4</figref>, frequency lock detector <b>432</b> may detect for frequency lock in various manners. In an embodiment, detector <b>432</b> initially resets a counter to zero. Thereafter, detector <b>432</b> compares each frequency error estimate Δ{circumflex over (f)}_mfrom estimator <b>430</b> against a threshold Δ{circumflex over (f)}_th, increments the counter if the frequency error estimate is less than the threshold, and decrements the counter otherwise. Detector <b>432</b> may declare frequency lock if the counter reaches a maximum value and may declare loss of lock if the counter reaches zero. The number of bits for the counter and the threshold Δ{circumflex over (f)}_thmay be selected to achieve good lock detection performance. Frequency lock may also be detected in other manners.

In an embodiment, the AFC loop may be operated in an acquisition mode or a tracking mode. Different parameter values may be used for the AFC loop in the two modes. A larger loop gain α may be used for the acquisition mode, and a smaller loop gain may be used for the tracking mode. The frequency error estimate Δ{circumflex over (f)}_mmay also be limited to within a larger range for the acquisition mode and to within a smaller range for the tracking mode. The acquisition and tracking modes may also be implemented in other manners. The terminal may support different and/or additional modes. For example, the terminal may also support a hold mode in which the AFC loop is maintained fixed, e.g., if the received signal quality is poor or if some other conditions are detected.

The terminal may start in the acquisition mode when powered on, after waking up from an extended sleep, when frequency lock is lost, and/or for other conditions. The terminal may transition from the acquisition mode to the tracking mode upon detecting frequency lock, or if the adjustment applied to frequency register <b>448</b> is below a particular value for some number of updates, or if some other conditions are satisfied.

The terminal may periodically receive data from the broadcast system. For example, each frame may carry many OFDM symbols (e.g., approximately 300 OFDM symbols), and the terminal may receive only few OFDM symbols (if any) in each frame. In this case, the terminal may sleep for most of the frame, wake up several OFDM symbols prior to the first OFDM symbol of interest, and process each OFDM symbol of interest. The terminal may update the AFC loop in each OFDM symbol period while awake and may hold the AFC loop while asleep.

In the embodiment shown in <figref idrefs="DRAWINGS">FIG. 4</figref>, the AFC loop is a first-order loop having a transfer function of _Hafc(s) in the s-domain, which may be expressed as:

<maths id="MATH-US-00006" num="00006"><math overflow="scroll"><mtable><mtr><mtd><mrow><mrow><msub><mi>H</mi><mi>afc</mi></msub><mo>⁡</mo><mrow><mo>(</mo><mi>s</mi><mo>)</mo></mrow></mrow><mo>=</mo><mrow><mfrac><mi>α</mi><mrow><mi>s</mi><mo>+</mo><mi>α</mi></mrow></mfrac><mo>.</mo></mrow></mrow></mtd><mtd><mrow><mi>Eq</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mn>11</mn><mo>)</mo></mrow></mrow></mtd></mtr></mtable></math></maths>

The bandwidth of the AFC loop may be expressed as:

<maths id="MATH-US-00007" num="00007"><math overflow="scroll"><mtable><mtr><mtd><mrow><msub><mi>BW</mi><mi>afc</mi></msub><mo>=</mo><mrow><mfrac><mi>α</mi><mn>4</mn></mfrac><mo>.</mo></mrow></mrow></mtd><mtd><mrow><mi>Eq</mi><mo>⁢</mo><mstyle><mspace width="0.8em" height="0.8ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mn>12</mn><mo>)</mo></mrow></mrow></mtd></mtr></mtable></math></maths><br /> The AFC loop bandwidth may be selected to achieve the desired frequency acquisition and tracking performance. The desired AFC loop bandwidth may be obtained by selecting the proper value for the loop gain α.

For clarity, the AFC loop has been described for a specific broadcast system. Other designs may also be used for the AFC loop. In general, the AFC loop may be designed in accordance with the structure of the signal transmitted by the system and the radio technology used by the system.

Referring back to <figref idrefs="DRAWINGS">FIG. 3</figref>, the terminal may perform frequency control in various manners. In one embodiment, AFC unit <b>340</b> corrects for both short-term frequency variations (e.g., due to Doppler effect) and long-term frequency variations (e.g., due to component tolerances and temperature variations). In another embodiment, the AFC loop within AFC unit <b>340</b> corrects for short-term frequency variations, and an outer loop within AFC controller <b>342</b> corrects for long-term frequency variations. The outer loop may receive the frequency error Δ{circumflex over (f)}_errfrom AFC unit <b>340</b>, average this frequency error over a longer period (e.g., a frame or a burst of data), and generate the frequency control signal for reference oscillator <b>322</b> based on the average frequency error. In yet another embodiment, AFC controller <b>342</b> may receive frequency error estimates from other AFC loops for other systems that are also being received by the terminal. AFC controller <b>342</b> may then generate the frequency control signal such that good performance may be achieved for all systems being received. The frequency control signal may be an analog signal or a digital signal. AFC controller <b>342</b> may perform signal conditioning such as digital-to-analog conversion, level shifting, scaling, and so on. AFC controller <b>342</b> may also generate a pulse width modulated (PWM) control signal.

<figref idrefs="DRAWINGS">FIG. 10</figref> shows an embodiment of a process <b>1000</b> for performing frequency control in a system that transmits a pilot along with OFDM symbols. Frequency acquisition is performed based on a received pilot, e.g., a TDM pilot multiplexed with OFDM symbols (block <b>1012</b>). Frequency tracking is performed based on received OFDM symbols (block <b>1014</b>). For frequency acquisition, an initial frequency error estimate may be derived based on the received pilot, and an AFC loop may be initialized with the initial frequency error estimate. For frequency tracking, a frequency error estimate may be derived for each received OFDM symbol, and the AFC loop may be updated with the frequency error estimate.

The pilot may comprise multiple pilot sequences. The initial frequency error estimate may then be derived by performing delayed correlation on the pilot sequences. The frequency error estimate for each received OFDM symbol may be derived by performing delayed correlation between samples for the cyclic prefix and samples for the data portion of the received OFDM symbol. Frequency error in input samples is corrected by the AFC loop with the initial frequency error estimate as well as the frequency error estimate for each received OFDM symbol.

<figref idrefs="DRAWINGS">FIG. 11</figref> shows an embodiment of an apparatus <b>1100</b> for performing frequency control. Apparatus <b>1100</b> includes means for performing frequency acquisition based on a received pilot (block <b>1112</b>), and means for performing frequency tracking based on received OFDM symbols (block <b>1114</b>).

<figref idrefs="DRAWINGS">FIG. 12</figref> shows an embodiment of a process <b>1200</b> for deriving a frequency error estimate. A variable number of samples of a received OFDM symbol are selected for use for frequency error estimation (block <b>1212</b>). A frequency error estimate is derived based on the selected samples (block <b>1214</b>). For block <b>1212</b>, the timing of the received OFDM symbol may be determined, and samples may be selected based on the timing of the received OFDM symbol. In an embodiment, the start of an FFT window is determined based on the timing of the received OFDM symbol. The samples to use for frequency error estimation are then selected from among the samples within the FFT window and for the cyclic prefix of the received OFDM symbol. In an embodiment, if the start of the FFT window is within a first range of the cyclic prefix, then samples in a second range of the cyclic prefix are selected. The first range may cover the first half of the cyclic prefix, and the second range may cover the second half of the cyclic prefix, as shown in equation (9). The first range may cover the third quarter of the cyclic prefix, and the second range may cover the fourth quarter of the cyclic prefix, as also shown in equation (9). No samples may be selected for frequency error estimation if the start of the FFT windows falls within a particular range (e.g., the fourth quarter) of the cyclic prefix.

<figref idrefs="DRAWINGS">FIG. 13</figref> shows an embodiment of an apparatus <b>1300</b> for deriving a frequency error estimate. Apparatus <b>1300</b> includes means for selecting a variable number of samples of a received OFDM symbol for use for frequency error estimation (block <b>1312</b>) and means for deriving a frequency error estimate based on the selected samples (block <b>1314</b>).

<figref idrefs="DRAWINGS">FIG. 14</figref> shows an embodiment of a process <b>1400</b> for determining an initial frequency error estimate. Correlation is performed on input samples to obtain a correlation result, which is a complex value having an inphase component and a quadrature component (block <b>1412</b>). The inphase and quadrature components are mapped to a numerator and a denominator, e.g., such that the numerator is less than or equal to the denominator (block <b>1414</b>). The ratio of the numerator to the denominator is determined using a first look-up table, which may store a table of inverse values (block <b>1416</b>). The arctangent of the ratio is determined using a second look-up table, which may store a table of arctangent values for a range of 45 degrees (block <b>1418</b>). The arctangent of the complex value is then determined based on the arctangent of the ratio, e.g., by flipping the phase and/or inverting the sign of the arctangent of the ratio, if needed (block <b>1420</b>). A frequency error estimate is then derived based on the arctangent of the complex value (block <b>1422</b>).

<figref idrefs="DRAWINGS">FIG. 15</figref> shows an embodiment of an apparatus <b>1500</b> for determining an initial frequency error estimate. Apparatus <b>1500</b> includes means for performing correlation on input samples to obtain a correlation result, which is a complex value having an inphase component and a quadrature component (block <b>1512</b>), means for mapping the inphase and quadrature components to a numerator and a denominator (block <b>1514</b>), means for determining the ratio of the numerator to the denominator using a first look-up table (block <b>1516</b>), means for determining the arctangent of the ratio using a second look-up table (block <b>1518</b>), means for determining the arctangent of the complex value based on the arctangent of the ratio (block <b>1520</b>), and means for deriving a frequency error estimate based on the arctangent of the complex value (block <b>1522</b>).

<figref idrefs="DRAWINGS">FIG. 16</figref> shows an embodiment of a process <b>1600</b> for performing frequency control in a manner to account for phase shift due to changes in circuit blocks. A frequency error estimate for input samples is obtained (block <b>1612</b>). This frequency error estimate may be derived based on a received pilot, a received OFDM symbol, or some other part of a received signal. A phase adjustment due to change in at least one circuit block used to generate the input samples is also obtained (block <b>1614</b>). This phase adjustment may be due to a change in the gain setting for the circuit block(s). The frequency error estimate may be accumulated in each sample period. The phase adjustment may be accumulated once when the change in the circuit block(s) occurs. A phase value is determined for each input sample based on the frequency error estimate and the phase adjustment (block <b>1616</b>). Each input sample is rotated by the phase value for that input sample (block <b>1618</b>).

<figref idrefs="DRAWINGS">FIG. 17</figref> shows an embodiment of an apparatus <b>1700</b> for performing frequency control with phase compensation. Apparatus <b>1700</b> includes means for obtaining a frequency error estimate for input samples (block <b>1712</b>), means for obtaining a phase adjustment due to change in at least one circuit block used to generate the input samples (block <b>1714</b>), means for determining a phase value for each input sample based on the frequency error estimate and the phase adjustment (block <b>1716</b>), and means for rotating each input sample by the phase value for that input sample (block <b>1718</b>).

The frequency control techniques described herein may be implemented by various means. For example, these techniques may be implemented in hardware, firmware, software, or a combination thereof. For a hardware implementation, the processing units used for frequency control may be implemented within one or more ASICs, DSPs, digital signal processing devices (DSPDs), programmable logic devices (PLDs), field programmable gate arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, electronic devices, other electronic units designed to perform the functions described herein, or a combination thereof.

For a firmware and/or software implementation, the techniques may be implemented with modules (e.g., procedures, functions, and so on) that perform the functions described herein. The firmware and/or software codes may be stored in a memory (e.g., memory <b>182</b> in <figref idrefs="DRAWINGS">FIG. 1</figref>) and executed by a processor (e.g., processor <b>180</b>). The memory may be implemented within the processor or external to the processor.

The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.