High performance adsl line conditioner system and method

TECHNICAL FIELD [0001] The invention relates generally to the Digital Subscriber Line (DSL), and more specifically to systems and methods of line conditioning, providing high-performance digital broadband service in long local subscriber loops using ADSL (asymmetric DSL) technology.

BACKGROUND [0002] Most telecommunications companies provide analogue telephony services, often referred to as Plain Old Telephone Service (POTS), and other services for customers using existing infrastructure, typically including Twisted Pair (TP) cables for each telephone number. Each TP line is referred to as a subscriber loop or a subscriber line. POTS lines have been constructed to carry a single voice signal using a 3.4 kHz channel. DSL technology allows high speed and high bandwidth digital transmission over lossy TP lines, but requires signal processing to overcome transmission damage resulting, for example, from signal attenuation, crosstalk noise from signals present on other lines, signal reflection, radio frequency noise and impulse noise. Due to the very low frequency performance of conventional TP lines, high-speed DSL (10+ Mbps) operation is usually limited to short lengths of local loops between the Central Office (CO) and the Customer Premises - CPE) of 8,000-10000 feet (about 2.5-3 km), due to DSL signal degradation at higher frequencies. Signal attenuation, which is the largest part of transmission damage, increases with the frequency and length of the line. Accordingly, for a given transmission method, the achieved transmission bandwidth decreases as the line length increases. The achievable data transfer speed is also limited by other factors such as group delay, which is also a function of frequency, as well as crosstalk and noise.

[0003] Asymmetric DSL (ADSL) is a DSL technology that has a wider frequency spectrum in the descending direction from CO towards CPE than the frequency spectrum in the upward direction to CO, and uses much higher downward transmission than in the direction of ascending. This reduces near crosstalk and the frequency spectrum supports simultaneous transmission of POTS duplex and data to TP. ADSL1 has a maximum signal bandwidth of 1.1 MHz. ADSL2 + has a maximum signal bandwidth of 2.2 MHz. Usually a frequency spectrum between 0-30 kHz is reserved for the POTS service. The spectrum between 34-125 kHz is upstream data and the spectrum above 164 kHz is intended for downstream data.

[0004] To be able to compete with cable service providers, some telephone service providers use ADSL2 + technology for digital broadband data such as Internet access and Internet Protocol Television (IPTV) services on the same TP. Each NTSC TV channel (National Television System Committee) requires approximately 4.4 Mbps when using MPEG-2 compression, and an HDTV channel requires approximately 8 Mbps. Some telecommunications companies provide two TV channels "video on"

PZ / 4413 / AG EP 1 964 377 B1 as well as a 3.0 Mbps broadband Internet service that requires a total bandwidth of 11.8 Mbps. ADSL2 + technology achieves this total bandwidth, however, service providers can usually provide such a combination of video and broadband services to local loops up to 8,000 feet (about 2.5 km).

[0005] In an earlier application, cited above, a serial signal amplifier for TP cable signals is disclosed, which allows a significant improvement in the speed and range of conventional ADSL technology. However, it is also desirable to achieve greater improvement and better performance (speed and range) to enable newer DSL technologies, i.e. ADSL2 + and VDSL2, to operate reliably, along with conventional POTS services, over greater distances than are currently available. And at the same time it is desirable to implement technology that is easy to install, inexpensive and with low energy consumption, which can be powered directly from TP lines, on which the usual POTS service works without interference. These are the aims of the present invention.

[0006] US 2002106013 discloses a loop extension for improving DSL signal transmission in a local loop. The loop extension includes a Selective Line Termination and Equalization (SLTE) amplification circuit. The circuit is capacitively connected to the local loop through a Diagnostic / Control Unit (DCU) connected to the local loop to receive and process control signals from the control panel.

[0007] US 2005127993 discloses a system and method that provides automatic gain control of a multi-stage system. The method includes determining at least one parameter that is adapted to at least one of: maximizing the hardware capacity of each gain stage and mitigating the changes between parts of the multi-stage system.

SUMMARY OF THE INVENTION [0008] The invention provides systems and methods for conditioning DSL lines that provide high performance by optimizing the Signal-to-Noise Ratio (SNR) and signal quality over DSL frequency spectrum, using components and circuits with low noise and low power, which ensure good line correction and compensation, high rejection in normal mode and small crosstalk. More specifically, the invention provides faster data transfer over larger loop lengths than is possible with current technology and can provide a 12 Mbps data transfer rate with high signal quality, enabling simultaneous transmission of two TV channels and 3 Mbps broadband data over distances of 12,000 feet (approx. 3.7 km).

[0009] In one aspect, the invention provides a method of conditioning a twisted pair telephone line having the features of claim 1. 1.

[0010] In another aspect, the invention provides a line conditioner for use on a twisted pair telephone line having the features of claim 1. 10.

[0011] In specific embodiments, the invention uses peak correction to provide the desired frequency gain characteristics for line correction for DSL operation. The invention uses low noise and low power analog circuits to condition the lines that optimize the signal-to-noise ratio and signal quality, and operate at low power so that they can be powered via a twisted pair telephone line without interfering with normal telephone service.

VP / 4413 / AG

BRIEF DESCRIPTION OF THE FIGURES

EP 1 964 377 B1 [0012]

Figure 1 is a block diagram of a line conditioner according to the invention used in the local loop between a telephone exchange and the customer's subscriber's office;

Figure 2 is a diagram of an embodiment of the POTS filter / distributor that can be used with the invention;

Figure 3 is a diagram of an embodiment of the peak detector that can be used in the conditioner of the line of Figure 1;

Figure 4 is a schematic of a mixer that can be used in the conditioner of the line of figure 1;

Figure 5 is a diagram of an adjustable preamplifier in a descending direction that can be used in the conditioner of the line of Figure 1;

Figure 6 is a diagram of an embodiment of an adjustable preamplifier in an upward direction that can be used in the conditioner of the line of Figure 1;

Figure 7 is a diagram of an embodiment of a descending filter that can be used in the line conditioner of Figure 1;

Figure 8 is a diagram of embodiments of the peaks correcting amplifier and the amplifier that can be used in the line conditioner of Figure 1;

Figure 9 is a diagram of embodiments of the filter in the upward direction and the amplifier in the upward direction that can be used in the conditioner of the line of Figure 1;

Figure 10 is a representative characteristic of the frequency gain response of the peak-correcting amplifier of Figure 8.

DESCRIPTION OF PREFERRED EMBODIMENTS [0013] The invention is particularly well adapted to high performance line conditioner for use with ADSL2 + technology and will be described here in this context. It is worth noting that this is an illustration of only one use of the invention, and the invention can be applied to other high-bandwidth digital works, including VDSL2 and interlaced DSL systems, as well as other types of high-bandwidth digital works on twisted pair (TP) telephone lines .

[0014] Figure 1 is a block diagram of a preferred embodiment of a bidirectional broadband conditioner line 100 according to the invention. The line conditioner can be built into a local subscriber loop containing a twisted pair POTS in an intermediate position between the telephone exchange (CO) and customer premises (CPE) to provide high speed data, broadband and video services. In one embodiment, the line conditioner 100 can provide approximately 12 Mbps data transfer speed, which can support two separate 4.4 Mbps television video signals and one 3.0 Mbps broadband signal with the same or better performance than conventional solutions as also provide conventional POTS telephone service in a local loop of 12,000 feet (about 3.7 km) or more.

[0015] As indicated in Figure 1, a conventional local loop includes a twisted pair (TP) line with different signals, including the Tip (T) 102, 103, 106 and the Ring (R) 104, 105, 108. Tip lines and Ring carry analog signals that provide conventional POTS service in the frequency range up to about 30 kHz. The DSL modem (not shown) in the CPE processes digital signals

PZ / 4413 377 B1 to analog form from uplink transmission from CPE to CO on TP Tip and Ring lines on, for example, a frequency band from about 34 kHz to 125 kHz. Digital DSL subscriber line concentrator

DSL Access Multiplexer - DSLAM) (also not shown) in CO converts digital signals into analogue form for downstream transmission to CPE on Tip and Ring lines. The descending frequency spectrum may range from about 164 kHz to 2.2 MHz (for ADSL2 +).

[0016] The ascending signal from the CPE travels along the length of lines TP 102 and 104 between the CPE and the intermediate position of the line conditioner 100. The ascending path through the line conditioner 100 passes through mixer 110, adjustable preamplifier in the upstream direction 112, filter 114, amplifier 116 and subsequent mixer 120 (downward direction). Mixer 120 provides an ascending signal to CO on a different length of lines TP 106 and 108, respectively, between the line conditioner and the CO. The descending signal path from CO to CPE passes through Tip and Ring lines 106 and 108, and through mixer 120, adjustable broadband preamplifier in descending direction 122, filter 124, peak correction amplifier 126, amplifier 128 and mixer in ascending direction 110. Mixer 110 provides signals descending from amplifier 128 to CPE via lines Tip and Ring 102 and 104, respectively. The conventional telephone service does not pass through the line conditioner, but is transferred on the Tip and Ring 103 and 105 lines through the POTS 130 filter / splitter. The POTS filter / splitter blocks high-frequency ascending and descending signals so that they pass through the 100 conditioner, and provides low impedance path for low frequency telephone signals.

[0017] An embodiment of a conventional POTS filter / distributor is illustrated in figure 2. As shown, it may comprise a pair of transformers 132, 134 connected to the Tip and Ring lines 103 and 105 and a pair of capacitors 136 and 138 connected between the Tip and Ring lines, as shown. Positive and negative DC + V and -V voltages can be supplied from the Tip and Ring lines, respectively, to draw useful power from the circuits in the line conditioner. As will be described, the line conditioner 100 is preferably designed to draw low power, so that the current drawn from the Tip and Ring lines to power the line conditioner is small enough so as not to interfere with normal telephone operation.

[0018] The mixer in the upward direction 110 and the mixer in the downward direction 120, which can be substantially identical, perform two basic functions. They block the flow of conventional low-frequency telephone signals through the line conditioner from CO to CPE, causing them to pass through the POTS 130 filter / splitter, and separate the upstream and downstream signals for processing by the line conditioner. Descending mixer 120 couples signals descending from CO on lines 106 and 108 to the preamplifier in descending direction 122, receives signals coming on lines 140, 142 from the amplifier 116 and couples them to CO. The mixer additionally suppresses unwanted upstream signals that are coupled by the mixer to the preamplifier in the downstream direction 122. Similarly, the upstream mixer 110 couples the upstream signals from the CPE to the preamplifier in the upstream direction 112 and couples the downstream signals on lines 144, 146 from the amplifier 128 to CPE, via lines Tip and Ring 102 and 104, respectively. The upstream mixer 110 similarly suppresses unwanted downstream signals that are coupled by the mixer to the ascending path to the preamplifier in the ascending direction 112.

[0019] The line conditioner 100 is not an ordinary signal strength amplifier to overcome the signal attenuation introduced by the TP line to increase range, which is the approach used by conventional ADSL line regenerators. Instead, 100 conditioner

PZ / 4413 / AG optimizes the signal-to-noise ratio (SNR) and signal quality, resulting in better performance and further coverage, while minimizing power consumption to stay within the desired small energy budget imposed by telephone lines. The line conditioner optimizes SNR by reducing the background noise of the system to a low level by using operational amplifiers (WO) with low power, low noise and a very wide band and other active elements and by using a circuit design and structure that minimizes the contribution of thermal and ohmic noise in components circuit and provides good out-of-band signal attenuation, good common-mode attenuation and low crosstalk. In addition, the line conditioner optimizes signal parameters, especially the amplitude and group delay due to pre- and post-correction as well as pre- and post-compensation of descending and ascending signals entering and leaving the line conditioner to compensate for signal degradation introduced by TP lines. As used herein, the terms "correction", "correcting" and "equalizer" refer to adjusting the level of signal amplitude, for example, to correct TP attenuation. The terms "offset" and "offset" refer to adjusting the group delay parameter, for example, to correct for signal degradation resulting from the group delay introduced by TP lines.

[0020] The line conditioner automatically adjusts the gain correction applied to the signals based on the actual attenuation to which the signals are subject during the transmission on TP between CO and CPE. The line conditioner additionally applies predefined and post-compensation to signals to compensate (i.e. correct) the effects of TP line group delay from CO to the line conditioner and from the line conditioner to the CPE, respectively. Accordingly, by dealing with the total signal degradation introduced by TP lines and focusing on optimizing high signal performance and quality, the line conditioner achieves faster data transmission over the length of the line and significantly better range than conventional line regenerators that only deal with signal attenuation. In fact, the invention can achieve the same distance as conventional solutions at a quarter power. Line conditioner 100 corrects and compensates for both upstream and downstream signals, however, it processes downstream signals more than upstream signals because they are more degraded by TP due to their higher frequency spectrum.

[0021] Descending signals from the mixer 120 are fed to the broadband preamplifier 122, which provides a steady gain (or suppression) of the descending signal, determined automatically for the current session during the pre-operational calibration process performed by the microcontroller or microprocessor 160 (as described later). The descending signal from the preamplifier is then fed to the filter in the descending direction 124, which in the case of ADSL is a high-pass filter that passes downward frequencies above 164 kHz and blocks downstream frequencies below 125 kHz. The filter 124 is designed to have a sharp cut-off to strongly suppress unwanted upstream frequencies that are passed through mixer 120 and preamplifier 122 to the filter and to guarantee a low background noise of the system. In a preferred embodiment, the filter 124 is an order 11 high-frequency elliptic filter that provides attenuation at from about 80 dB to 90 dB of frequencies falling below 125 kHz.

[0022] The descending signals from the filter 124 are then routed to the peak-correcting amplifier 126. The peak corrector has maximum gain at a fixed frequency, which is preferably in the upper part of the ADSL descending signal frequency spectrum. "Q" peaks corrector response

PZ / 4413 / AG determines the sharpness of the peak and gain imposed on the frequency range around the set frequency. Correction affects the ability to carry signal data because it changes the relationship between the amplitude of the fundamental and harmonics of the ADSL signal. The peak correction amplifier post-corrects for attenuation of the frequency dependent signal of the descending signal caused by TP 106, 108 lines between CO and the line conditioner, and pre-corrects for the predicted frequency-dependent attenuation caused by lines TP 102, 104 from the signal conditioner to CPE, so that the descending signal coming to the CPE modem has the desired corrected gain-frequency characteristic. Amplifier 128 amplifies the descending signal from the peak-correcting amplifier 126 to provide the desired signal level in CPE, and feeds the signal to mixer 110, which couples the signal to Tip and Ring lines 102 and 104 for transmission to CPE.

[0023] The ascending path through the conditioner 100 is somewhat similar to the descending path, except that it may not include a peak correction amplifier. A correction amplifier is not required for an ascending ADSL signal because it operates on a lower (and narrower) frequency spectrum than the descending signal and is not subject to as much frequency-dependent degradation as the descending signal. Ascending signals leaving the mixer 110 are coupled to an adjustable amplifier / attenuator preamplifier in the ascending direction 112, which also applies a fixed gain or attenuation to the ascending signal, determined automatically by the microcontroller 160 during the pre-operational calibration process (described later). The incoming signals from the preamplifier 112 are fed to the low-pass filter 114, which preferably has a sharp cut-off slightly above 125 kHz. The upstream signals from the filter are passed through an amplifier 116, which amplifies the upstream signals and couples them through mixer 120 and lines Tip and Ring 106 and 108 to CO. Filter 114 suppresses all unwanted frequencies of the descending signal coupled by mixer 110 to preamplifier 112, and also achieves a lower background noise of the circuit. In a preferred embodiment, the low-pass filter 114 may also be an 11 elliptical filter that provides attenuation of 80 dB to 90 dB downward frequencies above 164 kHz.

[0024] As also illustrated in Figure 1, the line conditioner 100 may further include a peak detector of the downstream signal 162 connected to positive line 164 from the peak correction amplifier 126 and may include a peak detector in the upstream direction 166 connected to the positive line 168 from the filter in the upward direction 114 to the amplifier in the ascending direction 116. Peak detector outputs are fed to the microcontroller 160, which uses the obtained peak values ​​during the calibration / pre-correction correction procedure to automatically adjust the gain and attenuation settings of preamplifiers 112 and 122 to correct for attenuation of the ascending and descending signals introduced by the actual TP lines in which it is used is the line conditioner.

[0025] Briefly summarizing, during the pre-operative correction setting procedure, the line conditioner automatically calibrates to the current TP line on which it is used. The line conditioner does this first by determining the effective impedance of the TP line between the CPE and the line conditioner, which is a function mainly of the length and thickness of the cable between the CPE and the line conditioner, while the preamplifier in the ascending direction 112 is set to the default settings. Then, it sets the gain or attenuation of the preamplifier in the upstream direction 112 to a fixed nominal setting, preferably by selecting a value from a series of stored values

PZ / 4413 / AG EP 1 964 377 B1 to strengthen and suppress empirically determined. Similarly, it determines the downhill line impedance between the line conditioner and the CO, while the downstream preamplifier 122 is set to the default settings and uses the specific downhill impedance to select a fixed gain or nominal attenuation setting, also preferably from a range of empirically determined gain and attenuation values ​​that optimize performance.

[0026] More specifically, the pre-operative procedure for setting the line conditioner correction uses the standard DSL synchronization protocol that sets DSLAM in the CO and CPE modem. During the first setup step, the upstream amplifier 116 is turned off by the microcontroller 160, while the CPE modem sends standard handshaking tones in the CO direction with spaced frequencies in the uplink frequency spectrum. The tone settings sent by the CPE modem have a fixed power level and are spaced at fixed frequencies in accordance with the DSL synchronization protocol. Because amplifier 116 is off, the tones do not reach CO. Thus, DSLAM in CO remains unchanged and does not respond to setting tones. The ascending peak detector 166 detects the peak value of the complex tones on line 168 from the filter 114 and passes the detected peak value to the microcontroller 160, which may include an A / C converter that converts the detected peak value into a numerical value. Thus, the digitized detected peak value of composite tones at the upstream filter exit is a measure of the effective impedance of the TP line (depending on the length and thickness of the cable) between the CPE and the line conditioner. The microcontroller 160 uses a numerical, detected peak value to determine the gain and nominal attenuation setting in the preamplifiers in the ascending direction 112. When setting the upstream signal path, the amplifier 128 is turned on so that the CPE modem sees the active termination and tone synchronization signals are terminated as required for normal operation.

[0027] After setting the upstream signal path, the downstream signal path is set by turning on the amplifier in the upstream direction 116 and turning off the amplifier in the downstream direction 128. This allows the setting tones from the CPE modem to reach DSLAM in CO. CO responds by sending standard handshaking confirmation tones that are spaced at fixed frequencies in the descending frequency spectrum in accordance with the DSL synchronization protocol. As the descending amplifier 128 is off, the CPE modem does not respond to these descending tones and the CO continues sending them. Peak detector 162 detects the peak value of complex tones on line 164 from the peak corrector 126 and provides the detected peak value to the microcontroller 160, where the detected peak value can be converted to a numerical value. As with the ascending path setting, the detected peak value of the descending complex tones is a measure of the effective impedance of the TP line (depending on the length and thickness of the cable) between the line conditioner and DSLAM CO. The microcontroller 160 uses the detected peak to determine the next gain and nominal attenuation setting in the broadband preamplifier in the downward direction 122.

[0028] Thus, during the correction setting procedure, the microcontroller 160 analyzes two detected peaks from the ascending and descending signal path and can generate a pair of X and Y indices that are used as indexes of the two-dimensional (2-D) vector in the microcontroller memory, which stores the nominal values ​​of gain and attenuation. The 2-D vector responds to the indices X and

By generating the corresponding pair of gain (or damping) values, and the microcontroller 160 programs the gain (or damping) of the two preamplifiers 112 and 122. The gain values ​​stored in the 2-D vector can be determined empirically, using a reference system containing telephone line simulators that simulate a 26 AWG copper TP telephone line of varying length. The reference system enables empirical determination of preamplifier settings for optimal performance for various combinations of impedance and telephone line data transfer speed. Finally, when amplifiers 116 and 128 are turned on, with the selected nominal settings of gain and attenuation from the 2-D vector, which were previously loaded into preamplifiers 112 and 122, the CPE modem and DSLAM CO can negotiate and synchronize with each other. Due to the better signal quality provided by the line conditioner, DSLAM CO and CPE modem see each other as if they are closer than they really are, and can synchronize with faster data transfer and higher performance than would be possible conventionally for real distance.

[0029] A preferred embodiment of the peak detector that can be used in the present invention is illustrated in figure 3. The peak detector in the descending direction 162 and the peak detector in the ascending direction 166 may be substantially the same and be as illustrated in the figure. As shown, the peak detector may include a low noise and low power comparator 180 that compares the voltage input 182 with the reference voltage set by the voltage distribution network containing resistors 184, 185 connected to the output of the comparator, and charges capacitor 188 to a voltage determined by the level on entrance. The voltage on the capacitor determines the level of complexity of tones that make up the voltage input 182, whose amplitude in this preferred embodiment is proportional to the impedance of the TP line, which is a function mainly of the length and thickness of the cable. The analog-to-digital (A / C) 190 converter (which can be either separate or contained in the microcontroller 160 as described above) converts the voltage to a digital value. The default gain and attenuation values ​​that are set in preamplifiers 112 and 122 may be convenient values ​​chosen so that the peak detectors 162 and 166 work in the range compatible with the A / D converter.

[0030] Figure 4-9 illustrates in more detail preferred embodiments of the various components of the line conditioner 100 shown in Figure 1. The figures show the parameters of various circuit components such as resistors, capacitors and coils. They are merely representative parameters of the circuit elements that can be used in the specific embodiments illustrated in the figures to achieve the objectives of the invention in an ADSL2 + environment. Neither the specific embodiments illustrated in the figures for the various components of the line conditioner, nor the parameters of the circuit components selected for these embodiments are necessary for the invention. On the contrary, as will be appreciated by one skilled in the art, various embodiments, as well as circuits having other parameters, may be used to practice the invention.

[0031] Figure 4 shows a preferred embodiment of the mixer in the downward direction 120. As mentioned above, the mixer in the upward direction 110 may be preferably identical to the mixer in the downward direction 120. As shown in the figure, positive and negative signals from the Tip and Ring lines 106 and 108, respectively, enter the mixer with CO. The signals on these lines contain the entire ADSL2 + frequency spectrum from DC to 2.2 MHz. The differential signal entering the mixer through these lines is routed to the DSL 200 converter, which separates the CO from the line conditioner and blocks telephone signals from DC to

PZ / 4413 / AG EP 1 964 377 B1 kHz. A second DSL 200 converter is connected to a toroid 202 that provides noise suppression of the signal common to the signals entering the mixer and provides downward differential signals on the positive and negative lines 206 and 208 to the wideband preamplifier 122.

[0032] Positive and negative upstream signals from the amplifier in the upstream direction 116 on lines 140 and 142, respectively, enter the mixer from the other side of the DSL converter at the output of the torus 202. The "up" signals from the amplifier 116 enter the mixer through a pair of resistors 50 Ohms 210, 212, which match the impedance of the DSL 200 converter to the standard impedance of 100 Ohm CO. The "up" signals entering the mixer pass through the toroid 202 suppressing the common signal and the DSL converter 200 to lines 106 and 108 to CO. Capacitor 204 on the first DSL converter and capacitor 205 on the other side block DC current. Downstream signals from the second DSL converter on lines 206 and 208 are passed through an R / 2R 220 circuit / circuit containing two pairs of resistors 222, 224, 226 and 228. The parameters of the two resistors from each pair are in a ratio of 2: 1 and the nominal values ​​are indicated in the figure. The R / 2R 220 circuit / circuit provides approximately 4 dB of unwanted downstream signal entering the lines 140, 142 that are connected to the broadband preamplifier on lines 206 and 208.

[0033] Figure 5 shows a preferred embodiment of a wideband preamplifier with programmable gain 122. As shown in the figure, the preamplifier 122 may include two identical, one-sided, programmable amplifiers 240p and 240n, respectively, receiving a differential signal from the mixer on the positive and negative lines 206 and 208. Each programmable amplifier may contain a pair of 242, 244 low-noise and low-power wide-band operational amplifiers whose inverting inputs 250, 252, respectively, are connected to the outputs of the 246 single-bit multiplexer and the 248 three-bit multiplexer. Multiplexers can be low-power CMOS devices. Each multiplexer has multiple inputs that are connected to the connectors between the corresponding chain of multiple resistors connected in series. The multiplexers respectively receive single or three-bit, digital control signals from the microcontroller through the 254 collective line and act as switches that connect their outputs with the inverting inputs 250, 252 operational amplifiers 242, 244 with one of the inputs to the multiplexers from connectors between resistors of the corresponding resistor chain connected series. As shown, the outputs 256, 258 of operational amplifiers are also connected to the corresponding resistor chains connected in series. Accordingly, each multiplexer, by switching the multiplexer's output to a specific input from the resistor chain, changes the return ratio of the WO resistor to the input value of the WO resistor and, accordingly, changes the WO gain (or attenuation). Accordingly, by properly selecting the parameters of resistors in the resistor chain, the desired range of reinforcements and attenuations can be provided. For the resistor parameters illustrated in figure 5, the gain of WO 242 can be set by the 1 bit 246 multiplexer to 0 dB or 1.5 dB. Similarly, the WO 244 gain can be set by a 3-bit 248 multiplexer in the range from -10.5 db (attenuation) to +10.5 dB (amplification) in 3 dB steps. Because WO 242 and 244 are connected in series, the gain of the preamplifier 240 can be controlled and programmed by the microcontroller in the range from 10.5 dB to +12,0 dB.

[0034] Figure 6 shows an embodiment of a preamplifier in the upstream direction 112 with programmable gain. The preamplifier in the ascending direction 112 may indeed have one

The design itself, like the top-down preamplifier 122, in addition to being able to use different resistor parameters to achieve a different range of gain and attenuation. As shown in Figure 6, the upstream preamplifier may include a pair of identical, one-sided preamplifiers 270p and 270n for amplifying or attenuating the upstream differential signal on positive and negative lines 266 and 268, respectively, from the upstream mixer 110. As with the downstream preamplifier 122, each upstream preamplifier 270 may contain a pair of broadband low-noise and low-power WO 272, 274 whose corresponding inverting inputs 276, 278 are connected to the outputs of single-bit multiplexer 280 and three-bit multiplexer 282, respectively. Each multiplexer can have multiple inputs connected to resistor connectors in corresponding chains of multiple resistors connected in series, and each multiplexer allows you to change the ratio of the resistance response to the input resistance corresponding to WO by connecting WO inputs to different resistor connectors in the resistor chain. Figure 6 illustrates representative parameters of resistors that can be used in resistor chains that allow WO 272 to be set to either 0 dB or -3 dB (attenuation); and allow you to change the gain of WO 274 from -21 dB to +21 dB and in steps of 6 dB. This allows you to change the total preamplifier gain in the ascending direction 112 between -24 dB and +21 dB under the control of the microcontroller 160. As with the downstream preamplifier, the preamplifier segments in the upward direction 270p and 270n can be set and programmed by the microcontroller using control line 284 to have the same gain setting, so that input signals from mixer 110 on lines 266 and 268 are reinforced or suppressed for the same amount. The greater control range of the preamplifier in the upstream direction 112 than that provided by the preamplifier in the downstream direction 122 is due to the fact that the CPE modem may be close (e.g. significantly next to) the line conditioner.

[0035] As will be appreciated by those skilled in the art, because the reinforcements (or suppression) of WO in downward direction 242, 244 and WO in upward direction 272 and 274 are controlled by the ratio of the value of the return and input resistance, various parameters of the resistors can be used the same ratio and, respectively, the same gain or attenuation. However, since it is desirable to obtain the smallest background noise of the system to maximize SNR, it is desirable to minimize ohmic (thermal) noise. Accordingly, it is desirable to use small resistor parameters to reduce the proportion of ohmic noise caused by current flow through the resistors. However, using low resistances increases the current flowing through the resistors, which increases the operating power and can affect the rate of increase of the output voltage WO. Since the line conditioner is preferably powered from the TP line on which it is used, it is necessary to have a small power budget to avoid disrupting the normal telephone service. Accordingly, the resistor parameters in series of preamplifier resistor chains can be selected to achieve the desired balance between low noise, low power and high speed. Resistor parameters shown in Figures 5 and 6 were selected based on these considerations and representative parameters can be used to achieve the objectives of the invention. Similar considerations to the above were also used to select the parameters of representative resistors of other line conditioner circuits.

[0036] Figure 7 illustrates a preferred embodiment of the descending filter 124. For ADSL, the descending filter is a high-pass filter as mentioned above. As shown in the figure, the descending filter 124 preferably includes two identical, one-sided high pass filters 290p and 290n that respectively receive the differential signal from the positive and negative lines 292 and 294 from preamplifier 122. The two one-sided filters are preferably one differential filter for low noise performance. It is important that the high-pass filters provide good attenuation of the unwanted out-of-band frequencies that are coupled to the mixer 120 by the preamplifier 122. This minimizes the noise contribution of such external signals and contributes to a low background noise of the system. High pass filters 290p and 290n, in a preferred embodiment, include elliptic filters of the order of 11 with a low frequency cutoff at approximately 164 kHz, the lower end of the descending frequency spectrum. The parameters of the capacitive and inductive element illustrated in Figure 7 are suitable for a high-order 11 order high-pass filter at the desired frequency and attenuation from about 80 dB to 90 dB of unwanted upstream frequencies coupled to the filter by the preamplifier 122. This allows the noise background of the line conditioner to be maintained on or below 70 dB, which provides low out of band noise and high SNR. The 50 Ohm resistors 292 and 294 of each filter are designed for impedance matching. Two single-sided filter outputs on positive and negative lines 296 and 298, respectively, are provided as differential input to the peak corrector amplifier 126 as indicated in figure 8.

[0037] Referring to Figure 8, preferred embodiments of the peak-correcting amplifier 126 and the amplifier 128 are shown. As illustrated in the figure, the peak-correcting amplifier and the amplifier preferably include differential amplifier circuits that may have substantially the same structure. The main function of the peak correction amplifier is to provide post-correction of the descending signal to overcome signal degradation due to frequency-dependent attenuation introduced by the TP line between the CO and the line conditioner, and to provide pre-correction to correct in advance the expected degradation caused by the attenuation that the downstream signal experiences between the line conditioner and the CPE.

[0038] The peak correction amplifier 126 may include a pair of identical broadband WO 300 low noise and low power connected together as shown in figure 8 to form a differential amplifier. WOs receive descending one-sided signals from lines 296 and 298 at their non-inverting inputs, and have feedback resistors 306 connected between their outputs 308 and their inputs 310. The inverting inputs of the two WOs can also be connected to each other via a RC circuit comprising resistor 312 and capacitor 314, as shown. To the output 308 WO 300 of the amplifier correcting the peaks on the positive line 296 can be pinned as shown at 316 and it can be provided as input to the peak detector in the descending direction 162 (see figure 1).

[0039] The gain-frequency characteristic of the peak-correcting amplifier is determined by the parameters of the resistive and capacitive elements 306, 312 and 314. Figure 10 shows a representative characteristic of the gain-frequency response of the peak corrector 126. The gain of the peak-correcting amplifier is determined by the ratio of the back impedance to impedance entrance WO. As shown, the frequency gain characteristic increases to a peak at a specific frequency of 360 and then decreases

PZ / 4413 / AG above this frequency. The capacitor 314 parameter specifies the frequency at which the gain is at its peak, and the resistor parameter 312 determines the slope of the gain characteristic from the peak and the "Q" peak corrector. The actual frequency at which the gain reaches a peak is also a function of the internal parameters of WO 300. Using SPICE models for WO and simulation, the actual parameters of the resistance and capacitive elements can be tailored to specific WOs to provide the desired response characteristics of the gain-frequency response. Figure 10 shows a representative characteristic of the frequency gain amplifier of the peak correction amplifier for the values ​​indicated in Figure 8. As shown, the gain reaches a peak preferably at a frequency of 1.6-1.7 MHz. Increasing the capacitor 314 parameter shifts the peak gain down the frequency and, conversely, decreasing the parameter increases the peak gain frequency. Decreasing the resistor parameter 312 increases "Q" and provides a sharper peak and increases the gain drop factor, and, accordingly, increasing the resistor parameter flattens the gain characteristic. When the peak-frequency gain-curve curve is selected for the peak-correcting amplifier, for optimal correction performance for the predicted combinations of TP lines and distance (TP impedance as a function of cable thickness and cable length), the peak-correction amplifier correction characteristics can be determined. Then, the further correction provided by the line conditioner for the actual line on which it is used is achieved by setting the gain-attenuation characteristics of preamplifiers 112 and 122 as described above.

[0040] Amplifier 128 may similarly contain a pair of identical WO 320 which receive outputs 308 of WO 300 on their inverting inputs. WO 320 may similarly have feedback resistors 326 connected between their outputs on lines 144 and 146 and their inverting inputs 324, the inverting inputs may also be connected to each other via a RC circuit comprising resistors 330 and capacitor 332. Although the amplifier 128 has a similar structure to the peak-correcting amplifier, the resistance and capacitance parameters of the amplifier can be selected so that the amplifier does not make a significant contribution to the down-path gain-frequency response. In particular, the capacitor 332 in the amplifier can be selected to have a parameter such that the amplifier gain reaches a peak for the low frequency range of telephone signals and to achieve a substantially flat gain-frequency characteristic. Rather than being used to set the gain response, the capacitor 332 in the amplifier is mainly used to decouple the DC from the amplifier's WO.

[0041] To achieve high signal quality, the invention not only relates to the effects of TP line suppression on the downstream signal, but also relates to the effects of group delay. Accordingly, the line conditioner 100 of the invention is also built to compensate for ascending and descending signals to correct for unwanted effects of frequency-dependent group delay and, preferably, to provide a relatively constant group delay. Compensation is mainly used for the descending signal, for which, due to the higher frequency spectrum, the deleterious effect of group delay is the strongest. The post conditioner compensates for signals entering the line conditioner to correct the group delay introduced by TP lines through which the incoming signals are transmitted, and compensates for the signals coming out of the line conditioner for

PZ / 4413 377 EP 1 964 377 B1 of the predicted group delay that will be introduced by the TP lines through which the outgoing signals will be transmitted.

[0042] The group delay compensation provided by the line conditioner 100 for the descending signal is mainly determined by the peak-correcting amplifier 126 and by other circuits in the downward direction, such as wideband preamplifier 122 and filter 124. Accordingly, the characteristics of the peak correction amplifier, and the characteristics of the preamplifier and filter can be selected so that the pre- and post-compensation of the line conditioner provides the complete, desired, optimal group delay response by. This can be achieved using the reference system (described above) and models simulating the expected nominal characteristics of the TP line, on which the line conditioner and circuit characteristics in the descending signal chain will be used, to select the parameters of the elements providing the desired response.

[0043] Now considering the uplink path, as described above, the upstream mixer 110 may be substantially the same as the downstream mixer 120; and the preamplifier in the ascending direction 112 may be substantially the same as the preamplifier in the downward direction 122, except that the upstream preamplifier may be provided with different gain and attenuation settings. Because the ascending signal has a lower frequency spectrum, it does not undergo the same degradation as the descending signal and can have a significantly different level.

[0044] From the preamplifier in the upstream direction 112, one-sided upstream signals on the positive and negative output lines 400, 402 of the preamplifier in the upstream direction are fed to the filter in the upstream direction 114, which, as in the case of the downstream filter, preferably includes one-way filters instead of a differential filter. Preferred embodiments of the filter in the upstream direction 114 and the amplifier in the upstream direction 116 are illustrated in Figure 9. As shown, the filter in the upstream direction preferably includes two identical, elliptic low-pass filters in the order of 11 410p and 410n. The filters can be designed to have a sharp cut-off frequency slightly above the upper end of the ascending frequency spectrum at 125 kHz and to provide good attenuation of out-of-band frequencies outside the range of 80 dB to 90 dB that can be coupled by mixer 110 and preamplifier 112. The resistance, capacitive and induction parameters indicated in Figure 9 are exemplary values ​​that provide low pass filters with the desired characteristics. 50 Ohm 412 input resistors and 50 Ohm 414 output resistors are designed for impedance matching. One-sided outputs from the 410p and 410n filters on lines 420 and 422 are given as a differential input signal to the differential preamplifier in the upstream direction 116. The positive output signal line 420 from the 410p filter can be pinned as shown at 424 and can be directed to the peak detector in the upstream direction 166 (see Figure 1).

[0045] The upstream amplifier 116, in a preferred embodiment, comprises a differential amplifier that may have the same structure as the downward amplifier 128. As shown, the upward amplifier comprises a pair of WO 440 and 442 differential connected whose inputs non-inverting are connected to the positive and negative outputs 420 and 422, respectively, from the filter. Feedback resistors 446 and 448 can be connected between outputs 140 and 142, respectively, and inverting inputs WO, as shown. Inverting inputs can be

PZ / 4413 / AG also connected to each other via an RC network comprising a resistor 450 and a capacitor 452 as shown.

The preamp differential output in the ascending direction on lines 140 and 142 is fed to the mixer in the downward direction 120 as described above.

[0046] As described above, to achieve the desired high performance goal, it is desirable that the additional noise coming from the components and parts of the circuit is as low as possible and to use designs, patterns and structures for different line conditioner circuits that provide good common-mode attenuation and small crosstalk. In addition, since the operating power for the line conditioner is preferably drawn from the TP lines on which it is used, it is desirable that the operating power is low to avoid interference with the normal telephone service. To meet the desired objectives of low noise and low power, it is necessary to carefully select broadband WO with low noise and low power. In addition, it is desirable to minimize ohmic noise by using low-performance resistors in circuits. However, because resistance values ​​decrease as power consumption increases, you can make the right trade-offs to achieve your goals.

[0047] Filters in the upward and downward directions are also important for achieving high efficiency. This is particularly important for the descending filter because the higher frequencies of the descending signal are subject to degradation and loss more than the ascending signal. The type of filter selected and the slope falling characteristics not only affect the amount of suppression of unwanted out-of-band frequencies, which can contribute to an increase in the background noise, the filter also affects the group delay through the signal path. Elliptic filters of the order of 11 used in the invention have desirable characteristics with a steep downward slope and high attenuation, and their perimeter components can be easily adjusted to provide the desired group delay characteristics. It is worth noting, however, that other types of filters and other designs and elements may be used to achieve the objectives of the invention.

[0048] The standard ADSL protocol uses Reed-Solomon (RS) correction coding, which has the ability to recover lost data. The amount of lost data that can be recovered depends on the number of RS bits used for correction coding. If the data loss exceeds the set amount of RS data that can be corrected, encoding violation occurs and the data remain uncorrected. An alternative measure of data loss is the bit error rate Bit Error Rate - "BER"). The television picture uses the lossy UDP / IP protocol. As the number of RS encoding violations increases, pixelation occurs in which the pixel groups of the image are lost. This may cause the video image to "flash". Data integrity is a function of both signal quality and SNR. The high data transfer speed that can be achievable at high signal levels does not have to ensure high data integrity because the signal quality can be sufficiently reduced by group delay, crosstalk and inter-symbol interference to create an undesirable level of coding violations. The invention uses the number of coding and / or BER violations as a measure of high performance and optimizes performance by reducing the number of code or BER violations to an acceptable level.

[0049] The gain-attenuation levels of the preamplifiers 112 and 122 that are stored in the 2-D table can be derived empirically to these values ​​to obtain optimal performance for a given set of line conditions. Accordingly, detected peak voltage values ​​from peak detectors 162 and 166 for descending and ascending signals are an effective measure of line characteristics

TP between CPE, CO and the line conditioner. PZ / 4413 / AG EP 1 964 377 B1 Based on the detected values ​​that are compared to the effective impedance or distance, the values ​​derived empirically in the 2-D table can be selected to set the gains and attenuations to ensure the optimal level of performance for the specific TP line on which the line conditioner is used. It has been found that the invention, by focusing on performance, by maximizing SNR and optimizing signal quality, achieves the desired level of performance that is achievable at much greater distances between the line conditioner and CO and CPE than is achievable without the line conditioner.

[0050] Although the above description refers to specific embodiments of the invention, it will be recognized by those skilled in the art that changes can be made to these embodiments without departing from the spirit of the invention, the scope of which is defined in the appended claims.

VP / 4413 / AG

EP 1 964 377 B1