High performance adsl line conditioner system and method

TECHNICAL FIELD The present invention relates generally (DSL) to Digital Subscriber Line technology, and more particularly to a line of systems and methods for providing a broadband high quality digital service in a long local loop of subscriber conditioning using ADSL (asymmetric) DSL technology.

Background [0002] Most analog telephone companies provide telephone service, often referred to as Plain Old Telephone Service (POTS), and other services for clients using legacy infrastructure typically containing single twisted pair (TP) wires for each telephone number. Each TP line continues on the subscriber's loop or subscriber line. POTS lines have been designed to carry a single voice signal from the 3.4 kHz bandwidth. DSL technology enables high speed, high bit rate digital transmission over TP loss line, but requires signal processing to overcome transmission problems caused by, for example, signal suppression of crosstalk from signals occurring on other lines, reflected signal, radio-frequency noise and impulsive noise. Due to the very low high frequency performance of conventional TP lines, DSL broadband operations (over 10 Mbps) are usually limited to short local loop lengths between the telephone exchange (CO) and subscriber's customer premises (CPE) for 8,000 - 10,000 feet due to signal degradation DSL, on higher frequencies. signal attenuation, which is the largest component of transmission disturbances, increases the frequency and length of the line. In connection with the above, for a given transmission method, the maximum achieved transmission rate decreases with the increase of the line length. The speed of data transmission is also limited by other factors, such as group delay, which is also a function of frequency as well as crosstalk and noise.

[0003] Asymmetrical DSL (ADSL) is a DSL technology that has a larger downstream frequency range from CO towards CPE than upstream frequency spectrum to CO and employs a much higher downstream transmission rate than upstream bitrate. This reduces almost the end of crosstalk and the frequency spectrum enables simultaneous POTS duplex transport and data on tp. ADSL1 has a maximum bandwidth of the order of 1.1 MHz. ADSL2 + has a maximum signal bandwidth of the order of 2.2 MHz. Typically, the frequency spectrum between 0-30 kHz is reserved for the POTS service. Spectrum in the 34-125 kHz range upstream data and spectrum above 164 kHz for lower order data.

[0004] In order to compete with cable service providers, some telephone service providers use ADSL2 + technology of broadband digital data, such as Internet access and television services (IPTV) in the same TP. Each NTSC television channel requires approximately 4.4 Mbps using MPEG-2 compression, and the HDTV channel requires approximately 8 Mbps. Some companies

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The telecommunications services provide two channels of video-on-demand service as well as 3.0 Mbps of broadband Internet service, which requires a total data rate of 11.8 Mbps. ADSL2 + technology can achieve total transmission speed, however, service providers can provide such usually combined video and broadband services to local loops with a length of about^8,000 s^top.

[0005] The earlier application mentioned above discloses an in-line amplifier for TP wire signals that allows a significant improvement in speed and distance from conventional ADSL technology. However, it is also advisable to achieve greater improvement and higher performance (speed and range) to allow reliable operation of newer DSL technologies, i.e. ADSL2 + and VDSL2, along with conventional POTS services, over greater distances than are currently available. And in this way, it is also desirable to implement a simple installation, inexpensive, low power consumption technology that can^{to be} zasHane ^bthis^pabout^śre^dn^andAbout Nnia TP on ^ktrack^sc^h about^pforaging ^bI agree with the norm of us^Theat^{g POTS}. It is these ends that the present invention is directed.

[0006] US 2002106013 discloses a loop extension for improving the transmission of DSL signals over a local loop. The extension cord contains a selective line termination and leveling (SLTE) amplification DSL circuits. The system is capacitively coupled to the local loop via a diagnostic / control device (DCU), coupled to a local loop to receive and process control signals from the centralized control .

[0007] US 2005127993 discloses a system and method for providing automatic gain control of a multi-step system. The method includes determining at least one parameter that is adapted to at least one maximizing the hardware capabilities of each of the plural steps and reducing the increment proportion to a portion of the multistage system variations.

BRIEF DESCRIPTION OF THE INVENTION The invention provides DSL line air conditioning systems and methods that provide high performance by optimizing the signal-to-noise ratio (SNR) and signal quality across the entire DSL frequency spectrum using low noise, low power components, and circuits that provide good line alignment and compensation, high common rejection mode and low crosstalk. More specifically, the invention provides more data over longer loop lengths than is possible from the current state of the art and can provide data transmission of 12 Mbps with high signal quality, allowing simultaneous transmission of two television channels and 3 Mbps of broadband data transmission at distances of 12,000 feet.

[0009] In one aspect, the invention provides a conditioning method for a twisted pair of telephone lines having the features of claim 1.

[0010] In another aspect, the invention provides a line conditioner for use in a twisted pair of telephone lines having the features of claim 10.

[0011] In specific embodiments, the invention uses peak alignment to provide the desired frequency response characteristics to compensate for the DSL line. The invention uses low-noise, low-power analog circuits for line conditioning, which

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EP 1 964 377 B1 provides optimum signal-to-noise quality and signal quality while operating at low power, so that they can be powered by twisted telephone line pairs without interfering with a normal telephone service.

BRIEF DESCRIPTION OF THE DRAWING FIGURES [0012]

Figure 1 is a block diagram of a line conditioning apparatus according to the invention used in the local loop between a telephone exchange and premises of a subscriber's customer;

Figure 2 is a schematic diagram of an embodiment of a POTS / splitter filter that can be used in accordance with the invention;

Figure 3 is a schematic diagram of an embodiment of a peak detector that can be used in the conditioner of the line of Figure 1;

Figure 4 is a schematic diagram of a mixer that can be used in the nutrient line of Figure 1;

Figure 5 is a schematic diagram for further amplification of the pre-amplifier that can be used in the conditioner of the line of Figure 1;

Figure 6 is a schematic diagram of an embodiment of inlet gain adjustment of a pre-amplifier that can be used in the conditioner of the line of Figure 1;

Figure 7 is a schematic diagram of an embodiment of an outlet filter that can be used in the conditioner of the line of Figure 1;

Figure 8 is a schematic diagram of embodiments of a balance peak amplifier and controller amplifier that can be used in the conditioner line of Figure 1;

Figure 9 is a schematic diagram of filter inlet and driver amplifier embodiments that can be used in the conditioner of the line of Figure 1; and

Figure 10 shows a representative characteristic frequency gain of the peak corrector amplifier of Figure 8.

DESCRIPTION OF THE PREFERRED EMBODIMENTS [0013] The invention is particularly well adapted to the high performance line of conditioner for use in ADSL2 + technology and will be described in this context. It will be understood, however, that this illustrates only one tool according to the invention, and the invention applies to other high bitrate digital operations, including VDSL2 and interleaved DSL chips, as well as other types of bit rate digital operations on twisted pair (TP) telephone lines.

[0014] Figure 1 is a block diagram of a preferred embodiment of a bi-directional conditioning medium of broadband lines 100 in accordance with the invention. The air conditioner line can be inserted into a local subscriber loop including twisted pair pots in an intermediate position between the central office (CO) and the residential customer (CPE) to enable high data rates, broadband data transmission and video services. In one embodiment, the line

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EP 1 964 377 B1

Conditioner 100 can provide data rates of approximately 12 Mbs, which can support two separate 4.4Mbps TV video signals and one 3.0Mbps broadband signal with the same or better performance than conventional methods, as well as providing a conventional POTS telephone service , through a local loop of 12,000 feet or more.

[0015] As indicated in figure 1, a conventional local loop includes twisted pair (TP) diversified signal lines including the terminal (T) of lines 102, 103, 106 and Ring (R) of lines 104, 105, 108. TIP and RING analogue transmission lines that provide conventional POTS services in the frequency range up to about 30 kHz. The DSL modem (not shown) The CPE converts the digital signals into analog form on the upper transmission from CPE to CO through the terminal and the TP ring line, e.g., a frequency band from approximately 34 kHz to 125 kHz. DSL (DSLAM) (also not shown) CO converts digital signals to analog form for further forwarding to CPE on the apex and ring line. The further frequency spectrum may extend from approximately 164 kHz to 2.2 MHz (for ADSL2 +).

[0016] The upstream signal from CPE travels along the line of TP 102 and 104 between the CPE and the intermediate line conditioner 100. above the path along the nutrient line 100 is through the mixer 110, and the regulation of the upper preamplifier gain 112, filter 114, the controller amplifier 116, and another (downstream) mixer 120. The mixer 120 provides a signal for the upstream CO on a different length of the lines TP 106 and 108, respectively, between the conditioner lines and CO. the further signal path from the CO CPE is via the Tip ring and lines 106 and 108, and through the mixer 120, and gain control below the broadband preamplifier 122, filter 124, amplifier 126, amplifier equalizer peak 128, and the upper mixer 110. The mixer 110 provides further signals from the driver amplifier 128 to the CPE via the tip and the ring of lines 102 and 104, respectively. Conventional telephone service does not pass the line conditioner, but takes place at the terminal and ring 103 and 105 lines by POTS / POTS 130 splitter. Filter / high frequency splitter blocks upstream and downstream signals so that they go through conditioner line 100 and provide low impedance path for low frequency telephone signals.

[0017] An embodiment of a conventional POTS / divider filter is shown in Figure 2. As shown in the drawing, it may include a pair of transformers 132, 134 connected to the terminal and ring with lines 103 and 105 and a pair of capacitors 136 and 138 connected along the tip and annular wires, as shown. Positive and negative DC + V and -V voltages can be provided by the Tip and the ring lines, respectively, for the output of operating power for circuits in the line conditioner. As will be described, the Line 100 air conditioner is preferably intended for low power consumption so that the current drawn from the tip and the ring by the line feeding the nutrient line is low enough not to interfere with the normal functioning of the telephone.

The mixer 110 before and behind the mixer 120, which may be substantially the same, has two main functions. They block conventional low-frequency phone signals from the flow through the conditioner lines from CO and CPE, causing them to flow through the POTS filter / manifold 130, and they separate the upstream and downstream signals from each other for processing through the conditioner lines, Further signal mixer 120 pairs down from CO on lines 106 and 108 for the further preamplifier 122, receives upstream signals on lines 140, 142 with amplifier 116 and pairs of them

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EP 1 964 377 B1 to CO. The mixer additionally suppresses unwanted feedback signals that are connected through the mixer to the lower preamplifier 122. Likewise, above the mixer 110, it couples feedback signals from the CPE to the upper preamplifier 112 and further pairs signals on lines 144, 146 with the controller 128 to the CPE amplifier via terminals and ring lines 102 and 104, respectively. The inlet mixer 110 likewise mitigates the unwanted signals that are a few further through the mixer to the inlet path to the initial preamplifier 112.

The Line 100 nutrient is not just an amplifier that has signal strength to overcome the signal attenuation imposed by the TP line to expand the range that is the approach adopted by conventional ADSL line converters. Conversely, the Line Conditioner 100 optimizes signal-to-noise ratio (SNR) and signal quality, resulting in higher performance and greater coverage while minimizing power consumption to stay in the desired low energy consumption imposed by telephone lines. The nutrient line optimizes SNR by reducing to a low level of base noise of the system through the use of low noise, high bandwidth, low power operational amplifiers ("opamps") and other active components as well as structure and structure designs, which minimizes thermal and resistive noise components of the circuit components and ensures high rejection of out-of-band signal, high common signal attenuation and low crosstalk. In addition, the Line Conditioner optimizes the signal characteristics, mainly amplitude and group delay, through before and after alignment and pre- and post-compensation of the downstream and upstream input and output signals from line conditioner to compensate for signal degradation imposed by TP issue. As used herein, the terms "correction", "equalizes" and "equalizer" refers to the regulation of the amplitude level of signals, e.g. to correct the attenuation TP line. Compensation terms "i" compensates "refers to the regulation of the group delay characteristics, for example,

[0020] The air conditioner line automatically adjusts the gain balancing applied to the signals based on the actual attenuation of the experience signals during TP transmission between CO and CPE. The conditioner line additionally uses pre-programmed remuneration and post-compensation for signals to compensate (ie correct) for the effects of group TP line delay from the CO to the line conditioner and from the line conditioner to the CPE, respectively. Accordingly, by addressing the overall signal degradation imposed by TP lines, and by focusing on the optimization of high performance and signal quality, the air conditioner line of the invention achieves a higher data rate per line length and a much better range than conventional linear amplifiers that only apply to signal attenuation. Indeed, the present invention can achieve the same distance as traditional methods on one-fourth of a power. Conditioner line 100 equalizes and compensates for both upper and further signals, although it processes signals more than the feedback signals since the occurrence of the greatest degradation due to TP due to their higher frequency range.

[0021] Further signals from the mixer 120 are provided to the broadband preamplifier 122, which applies a predetermined gain (or attenuation) to the outlet signal, which is automatically determined for the current session during the pre-operative pre-calibration process.

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EP 1 964 377 B1 formed by the microcontroller or microprocessor 160 (in a manner that will be described). The downstream signal from the preamplifier is then supplied to the lower filter 124, which in the case of ADSL is a high pass filter that flows below frequencies above 164 kHz and upstream frequency blocks below 125 kHz. The filter 124 is designed to have a sharp cut-off to strongly reject the unwanted stream of frequencies that are connected through the mixer 120 and the preamplifier 122 to the filter, and also help to ensure low noise levels of the system. In a preferred embodiment, the filter 124 is an eleventh-order elliptical high-pass filter that provides about 80 dB to 90 dB downstream frequency rejection below 125 kHz.

[0022] For further signals from the filter 124 are then applied to the equalizer amplifier peak 126. The peak corrector has a maximum gain at a preset frequency, which is preferably at the top of the frequency spectrum of signals after ADSL. "Q" on the peak corrector effect o^kre^śland the sharpness of happiness^{yt and} reinforcement applied ^dabout the frequency wold for^dfrequencies. ^Your^aboutwnanto in^stovetopswa ^dane with^dOtoo^SC s^whoon^Theat ^pontowa^with the torso of the basic and harmonic frequencies of the ADSL signal is protected. The equalizer amplifier peak after equalizes to signal attenuation, the distal signal caused by the line TP 106, 108 between the CO and the Frequency-dependent nutrient and pre-equalizes for the expected attenuated frequency due to the line TP 102, 104 from the CPE conditioning, so that the signal down arrival of the CPE modem requested a balanced frequency response characteristic. The controller 128 amplifier amplifies the signal from the downstream repeater amplifier 126 to provide a suitable signal level on the CPE and provides a signal to the mixer 110 that couples the signal to the Tip and the line ring 102 and 104 for transmission to the CPE.

[0023] The upstream path through the conditioner line 100 is somewhat similar to the bottom path, except that it can not include the equalizer peak amplifier. The alignment amplifier is not necessary for the upper ADSL signal, because it operates in the lower frequency range (narrower) than the cell signal, it does not experience as much frequency dependent distribution as the derivative signal. The upstream outgoing signals of the mixer 110 are coupled to an upper regulated gain, preamplifier damping 112, which is also applied to a predetermined gain or suppression of the upper signal as automatically determined by the microcontroller 160 during the pre-calibration process (which will be described). The upstream signals from the preamplifier 112 are fed to the filter^dabout^lWell^psculpture^pus^tthose^goI^{4, k}track^s the root has people^s about^dc^outrays of light ^pthat^YZabove ^{125 kH}with. ^ups^tream s^whothe filter mats are passed to the driver amplifier 116, which amplifies the upstream signals of the dry pair via a mixer 120 on the tip and the annular lines 106 and 108 to the CO. The filter 114 will reject any of the unwanted further frequencies of the signals coupled through the mixer 110 to the 112 preamplifier, and also helps to obtain low system noise. In a preferred embodiment, the low pass filter 114 may also be an 11th order elliptical filter that provides 80 dB to 90 dB to reject further frequencies above 164 kHz. [0024] As also shown in figure 1,

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A microprocessor 160 that uses peaks detected during the initial operational calibration setup / equalizer to automatically adjust the gain and suppression of the preamplifiers 112 and 122 alignment for the suppression and down suppression of signals caused by real TP lines in which the conditioner line.

[0025] The pre-operative alignment summarized during the adjustment procedure, the air conditioner automatically calibrates the line to the actual TP line in which it is used. Conditioner line whether first determining the real TP line impedance between CPE and line conditioner, which is a function primarily the wire length and CPE spacing and line conditioner, in which before Preamplifier 112 is set to the default setting, Next sets the gain or suppression of the upper preamplifier 112, to a predetermined nominal setting, preferably by selecting values ​​from the array of experimentally stored benefit values ​​and attenuation. Similarly, the impedance determines the bottom line between the conditioner line and CO,

[0026] In more detail, the pre-operative operation of the air line line alignment configuration uses a standard DSL synchronization protocol that sets up DSLAM on the CO and the CPE modem. During the first configuration step, the upper driver 116 is turned off by the microcontroller 160, while the CPE modem sends towards the standard "reconciliation" of the CO configuration tones at certain frequencies in the upper frequency range. The configuration tones emitted by the CPE modem are at a fixed power level and arranged at fixed frequencies according to the DSL synchronization protocol. Because the controller 116 amplifier is turned off, the sounds do not reach CO. Thus, the DSLAM on the CO remains idle and does not respond to the tones of the configuration. Upstream peak detector 166 detects the peak of composite noise on line 168 with filter 114 and provides a peak value detected to microcontroller 160, which may include an A / D converter that converts the peak detected to a digital value, thus, the digital form of the peak detected composite of tones at the inlet filter output is a measure of the effective impedance of the TP line (constituting the wire with the length and width of the track) between the CPE and line conditioner. The microcontroller 160 uses the sensed digital peak value to set the nominal gain or the attenuation setting in the upstream direction of the preamplifiers 112. During the upstream signal path configuration, the downstream controller 128 is turned on so that the CPE modem sees the active ending and synchronization of the tone signals are completed required for normal operation.

After the configuration of the upper signal path, the distal signal path is set by switching on the upstream and downstream amplifier 116. The installation sounds with the CPE modem reach DSLAM on the CO. The CO responds by sending the "reconciliation" standard tones that are arranged at preset frequencies in the lower frequency spectrum, according to the DSL synchronization protocol. Since the downstream driver 128 is turned off, the CPE modem does not respond to these further sounds, and the CO continues to transmit them. The peak detector 162 detects the peak of the composite noise on line 164 from the top of the equalizer 126 and delivers

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The peak value detected for the microcontroller 160, where the peak value detected can be converted into a digital value. As with the upstream path configuration, the peak value detected by the composite further sounds is a measure of the effective impedance of the TP line (being the wire length and gauge) between the line conditioner and the CO DSLAM. The microcontroller 160 uses a peak value detected to program another nominal gain or attenuation setting to a further broadband preamplifier 122.

[0028] Thus, during the equalization installation procedure, the microcontroller 160 analyzes the two peak detected voltages from the upstream and downstream signal paths and can produce a pair of X and Y indices that are used as indicators in the two-dimensional (2-D) array in the microcontroller memory. that holds a specific nominal gain or value damping. The 2-D table is the response to X and Y of the indices by producing a corresponding pair of gain (or attenuation) values, and the microcontroller 160 programs the gain (or attenuation) to the two preamplifiers 112 and 122. The gain values ​​stored in the 2-D array can be experimentally determined using a reference system that includes telephone wires that simulate copper 26 AWG line simulators, TP telephone lines of various lengths. The reference system enables empirical determination of the preamplifier settings to obtain optimal performance for various combinations of telephone line impedances and data transmission rates. Finally, when the 116 and 128 amplifier drivers are turned on, with selected nominal gain or attenuation settings from the 2-D array has previously been loaded into the preamplifiers 112 and 122, the CPE and CO DSLAM modems can negotiate and synchronize with each other. Due to the better signal quality provided by the line conditioner, the CO DSLAM and the CPE modem seem closer to each other than in reality, and are able to synchronize with higher data rates and higher performance than other typically available real distance.

[0029] A preferred embodiment of the peak detector that can be used in the present invention is shown in Figure 3. The further peak detector 162, and the peak current detector 166 may be substantially the same and are as shown in the drawing. As shown, the peak detector may include low noise, low power consumption of the comparator 180, which compares the voltage input 182 with the reference voltage determined by the network. A voltage divider consisting of resistors 184, 185 connected to the comparator output and charging^khe^densatora ¹⁸⁸ That's on^Occ^andthat one^smarked^ypslaughter ^poz^andom at the entrance on^Octie on ^khe^dthe sender represents the level of the composite tones that make up the input voltage 182, whose amplitude in this preferred embodiment is proportional to the impedance of the TP line, which is a function primarily of the length of the wire and the indicator. The analog-to-digital (A / D) converter, converter 190 (which may be separate or within the microprocessor 160, as described above) converts the voltage to a digital value. The default gain or attenuation values ​​that are set under preamplifiers 112 and 122 may be convenient. Values ​​selected so that the peak detectors 162 and 166 operate in a compatible range of the A / D converter.

[0030] Figures 4-9 show more particularly preferred embodiments of these different ingredients in the nutrient line 100 shown in Figure 1. The figures show values ​​for many components of the system, such as resistors, capacitors and coils. These values ​​are only

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Representative elements of the printed elements that can be used for specific embodiments shown in the drawing to achieve the objectives of the present invention in the ADSL2 + environment. Neither the particular embodiments illustrated in the figures in relation to the various components of the line conditioner, nor the values ​​of the circuit elements selected from these embodiments are necessary for the invention. Rather, as will be understood by those skilled in the art, other embodiments as well as circuits that meet other values ​​may be used in the practice of this invention.

[0031] Figure 4 shows a further preferred embodiment of a mixer 120. As already mentioned, the above mixer 110 can be substantially identical to a further mixer 120. As shown in the figure, positive and negative signals on the top and annular lines 106 and 108, respectively, inputs mixer with CO. signals on these lines cover the entire ADSL2 + frequency spectrum from DC to 2.2 MHz. The differential signal of the mixer input of these lines is to the DSL 200 transformer, which isolates the CO from the line air conditioner and blocks the 30 kHz telephone signals. The secondary transformer DSL 200 is connected to the toroid 202, which is in the common mode of rejection of noise for signals input into the mixer and provides differential signals of further positive and negative lines 206 and 208 from the preamplifier 122 of the broadband. [0032] Positive and negative signals upstream of the controller amplifier 116 on lines 140 and 142, respectively, inputs to the mixer on the secondary side of the DSL transformer at the toroid output 202. Signals in the up link from the amplifier 116 enter into the mixer by means of a pair 50 ohm resistors 210, 212 that match the DSL 200 transformer's impedance to a standard 100 ohm impedance with CO. uplink signals coming into the mixer are connected via common rejection mode toroid 202 and DSL transformer 200 on lines 106 and 108 to CO. a capacitor 204 after the primary DSL transformer and a capacitor 205 on the other side of the DC block. The secondary signals from the secondary DSL transformer on lines 206 and 208 are connected via an R / 2R network 220 comprising two pairs of resistors 222, 224, 226 and 228. The values ​​of the two resistors in each pair have a 2: 1 ratio, and nominal values ​​are given in the figure. R / 2R network 220 contains about 4 dB decrease in signal strength of the undesired input signal input on lines 140, 142 that are coupled to the preamplifier on Niw^{206 and 208}.

[0033] Figure 5 shows a preferred embodiment of the programmable broadband further amplification of preamplifier 122. As shown in the figure, Amplifier 122 can include two identical one-node programmable 240p and 240n amplifiers respectively receiving a differential signal from a positive and negative mixer of lines 206 and 208. Each programmable amplifier may contain low low power low-power noise with high-capacity operational amplifiers 242, 244 having their input Reversal 250,

252, which are connected to the one-bit outputs of the multiplexer 246 and the three-bit 244 multiplexer. The multiplexers can be low power to the CMOS devices. Each multiplexer has a plurality of inputs that are connected to the nodes between the corresponding chain of multiple resistors connected in series. Multiplexers respectively receive single-bit signals or three-bit digital control systems from a microprocessor through the line bus 254, and serve as switches that connect them

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The inverter outputs 250, 252 from the operational amplifiers 242, 244 to one of the inputs to the multiplexer from the intersection between the resistors of the corresponding series of resistors of the series. As shown at outputs 256, 258 operational amplifiers are connected to the appropriate resistor networks. Accordingly, each multiplexer by switching the output of the multiplexer to a specific signal from the network of resistors changes the value of the opamp feedback resistor to the value of the opamp input resistor, and therefore depends on the OpAmp gain (or attenuation). Accordingly, by the appropriate selection of resistor values ​​in resistor chains, the desired range of gain or damping can be provided. For the resistor values ​​shown in Figure 5, the opamp 242 gain may be controlled by the 1-bit multiplexer 246 will be 0 dB and 1.5 dB. Similarly, the gain of opamp 244 can be controlled by a 3-bit multiplexer 248 ranging from -10.5 dB (attenuation) to +10.5 dB (amplification) 3 dB. Since opamps 242 and 244 are in series, the preamplifier 240 can be controlled and programmed by the microcontroller in the range of -10.5 dB to +12.0 dB.

[0034] Figure 6 shows an embodiment of programmable preamplifier gain up 112. the preamplifier 112 may have substantially the same structure as the further preamplifier 122, however, it may use different values ​​of resistors and provide a different range of benefits and damping. As shown in Figure 6, the upper preamplifier may include a pair of identical twin ends of 270p and 270N preamplifiers for amplifying or attenuating the differential signal upstream of positive and negative lines 266 and 268, respectively, from above^Iwill be ΊΊ⁰. Talk me^k in ^pgovernment^ypand^minat ^das well^s amplifier on^pulsen^{and 122,} with ^ktrack^sc^hpand^psculpture^dThe amplifier 270 may include low noise pair, low power, high band opamps 272, 274 with their inverting input 276, 278 connected to the outputs of the one bit 280 multiplexer and to the three bit multiplexer 282, respectively. Each multiplexer can have multiple inputs connected to the resistor nodes in the respective multiple resistor chains and each multiplexer allows the ratio of the reverse resistance to the input resistance corresponding to its opamp to be changed by connecting the opamps inputs to the different resistor nodes in the resistor chains. Figure 6 shows representative values ​​of resistors that can be used in a network of resistors that allow the opamp 272 to be amplified for control to be 0 dB -3 dB (attenuation); and allows the opamp 274 to amplify to -21 dB to +21 dB in^kro^kac^h What ^{6 d}B. ^pit was on ca.^Lukeowrte reinformeme ^himrnej ^psculpture^dwzmacn^andACZA ¹¹² s^out varies from -24 dB to +21 dB under the control of the microprocessor 160. As with the further preamplifier, the upper preamplifier segments 270p and 270N can be controlled and programmed by the microcontroller via the control line 284 have the same gain setting so that the input signals from the mixer 110 on lines 266 and 268 are amplified or suppressed by the same amount. The greater the range of control of the preamplifier 112 with the preamplifier 122 offered in the lower preamplifier is because the CPE modem can be near (e.g. much further) the conditioner to the line.

[0035] As will be appreciated by those skilled in the art, as reinforcements (or damping) with further opamps 242, 244, and also before opamps 272 and 274 are controlled by feedback coefficients to the value of the input resistance, different values ​​of resistors can be

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EP 1 964 377 B1 used to achieve the same index, and thus to the gain or weakening itself. However, since it is desirable to achieve the lowest possible system noise as possible to maximize SNR, it is desirable to minimize ohmic (thermal) disturbance. Therefore, it is desirable to use low values ​​of resistors to minimize the ohmic contribution of noise caused by the current flowing through the resistors. However, using low resistance increases the amount of current flowing through the resistors, which increases the operating power and can affect the prices killed opamps. Since the air conditioner line is preferably fed from the TP line in which it is used, it is necessary to have a low power budget to avoid disrupting the normal telephone service. Accordingly, they can be chosen, to achieve the desired balance between low noise, low power and high speed resistor values ​​in the resistor network. Preamplifier series. The resistor values ​​shown in Figures 5 and 6 are selected based on these considerations, and are values ​​that can be used to achieve the objectives of the invention. Similar considerations in mind above have also been used in the selection of representative values ​​of resistors of other circuits of the line conditioner.

[0036] Figure 7 shows a preferred embodiment of an exhaust filter 124. The ADSL bottom filter is a high pass filter as previously indicated. As shown in the drawing, the filter below 124 preferably comprises two identical single-ended highpass filters and 290n 290p which respectively receive the differential signal positive and negative lines 292 and 294 from the preamplifier 122. The two single end filters are preferred to one differential filter for low noise emission. It is important that the high mid filters provide high rejection of the unwanted out-of-band upstream frequencies that are connected to the 120 mixer by the preamplifier 122. This minimizes the contribution of such external signals and contributes to the low noise of the system. 290p high-pass filters and 290n, in a preferred embodiment, comprise an order of eleventh elliptical Filters having a low cutoff of approximately 164 kHz, the lower end of the lower frequency spectrum. The capacitive and inductive elements The values ​​shown in Figure 7 are suitable for the eleventh high pass filter with sharp cut at the required frequency and about 80 dB to 90 dB rejection of the unwanted frequency stream in combination with the filter through the preamplifier 122. this allows the noise line conditioner to be maintained at or below 70 dB, which gives a low out-of-band noise and high SNR. At 50 ohm, the resistors 292 and 294 of each filter are for impedance matching. Two single-ended filter outputs on positive and negative lines 296 and 298, respectively,

[0037] Referring to Figure 8, preferred embodiments of the equalizer enhancer 126 and controller amplifier 128 are shown. As shown in the figure, the equalizer amplifier and amplifier peak The controller preferably includes differential amplifier circuits that can have substantially the same structure. The primary task of the equalizer amplifier is to provide post-correction of the signal further overcome the attenuation of the distribution signal dependent on the frequency of the imposed TP line between the CO and line conditioner and provide initial equalization to correct the predicted degradation of the prediction earlier that the signal will be further

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EP 1 964 377 B1 experiments between a line conditioner and CPE.

[0038] Equalizer peak emitter 126 may include a pair of identical low noise low power high opamps 300 connected to each other as shown in Figure 8 to form a differential amplifier. Opamps receive further unbalanced signals on lines 296 and 298 on their non-inverting inputs and have feedback resistors 306 connected between their 308 outputs and their inverting inputs 310. The inverting two opamps inputs can also be connected to each other via an RC circuit including a 312 resistor and capacitor 314 as shown in the picture. The output 308 opamp amplifier peaks 300 on positive line 296 can be used as shown in 316 and provided as a contribution to a further peak detector 162 (see Figure 1).

[0039] The characteristic frequency gain of the equalizer peak amplifier is determined by the resistance and capacitive elements 306, 312 and 314. Figure 10 shows a representative characteristic frequency gain for the equalizer peak 126. Amplifier gain The equalizer is the peak determined by the impedance ratio response of the opamps input impedance. As shown, the characteristic frequency gain increases to the peak at a specific frequency of 360 and then decreases above this frequency. The value of capacitor 314 determines the frequency at which peak gain occurs, and the value of resistor 312 determines the flow of the peak-gain and Q-characteristics of the equalizer peak. Actual frequency, at which peak gain is also a function of the inherent features of opamps 300. Using SPICE models for opamps and simulations, the actual values ​​of resistive and capacitive elements can be tailored to specific opamps to provide the desired frequency response response feature. Figure 10 shows a representative characteristic of the amplifier peak frequency gain to the values ​​given in Figure 8. As shown, the peak gain is preferably at a frequency of 1.6-1.7 MHz. Increasing the value of the capacitor 314 moves the gain of the lower frequency peak, and vice versa, reducing the value increases the peak amplification frequency. Lowering the resistor value of 312 increases the "Q" and provides a sharper peak and faster roll off of profit, and by increasing the value of the resistor, the flattening of the gain characteristic. When the peak of the frequency enhancement curve for the peak amplifier is selected to obtain optimal leveling efficiency for the predicted applications of the distance tp line (TP impedance as a function of wire diameter and wire length), the characteristic alignment of the equalizer amplifier can be repaired. Then, further alignment through the nutrient line for the actual line in which it is applied is done by determining the gain-suppression characteristics of the preamplifier 112 and 122, as previously described. When the peak of the frequency enhancement curve for the peak amplifier is selected to obtain optimal leveling efficiency for the predicted applications of the distance tp line (TP impedance as a function of wire diameter and wire length), the characteristic alignment of the equalizer amplifier can be repaired. Then, further alignment through the nutrient line for the actual line in which it is applied is done by determining the gain-suppression characteristics of the preamplifier 112 and 122, as previously described. When the peak of the frequency enhancement curve for the peak amplifier is selected to obtain optimal leveling efficiency for the predicted applications of the distance tp line (TP impedance as a function of wire diameter and wire length), the characteristic alignment of the equalizer amplifier can be repaired. Then, further alignment through the nutrient line for the actual line in which it is applied is done by determining the gain-suppression characteristics of the preamplifier 112 and 122, as previously described.

[0040] The driver amplifier 128 may similarly include a pair of identical opamps 320 that they receive at outputs 308 opamps 300 at their inverting inputs. The opamps 320 may also have combined feedback resistors 326 between their outputs on lines 144 and 146 and their inverting inputs 324, and the inverting inputs may also be connected to each other via an RC comprising resistor 330 and capacitor 332. Although the actuator 128 has a similar structure for the amplifier, peak, resistive and capacitive equalizer

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The values ​​of the driver amplifier can be chosen such that the driver amplifier does not significantly contribute to enhancing the response at all frequencies to the lower path. In particular, the capacitor 332 in the driver amplifier can be selected to have a value such that the driver peak gain is in the low frequency range of the telephone signals and provides a substantially flat frequency response characteristic. Instead of being used to set the response amplification, the capacitor 332 in the controller serves primarily as a DC separating the opamps of the controllers.

[0041] In order to ensure high quality of the signal, the invention concerns not only the effects of attenuation of the TP line on the lower signal, it also concerns the influence of the group delay. Accordingly, the Conditioner line 100 of the invention is also formed to compensate for the directions of the signals at the same time to correct the negative effects of the dependent group frequency delay and preferably to obtain a relatively constant group delay. Compensation is used primarily for signals for which because of the higher frequency range, the detrimental impact of group delay most severe. The conditioner line balances the signals entering the line conditioner to the group delay correction imposed by the TP lines to which the input signals were transmitted, and pre-aligns the signals leaving the band delay line conditioner predicted

[0042] Compensation group delay provided by conditioner line 100 on the downstream signal is primarily determined by the equalizer peak 126 and other end circuits such as the broadband preamplifier 122 and the filter 124. Therefore, peak amplifier characteristics and equalizer and characteristics of the preamplifier and filter can be chosen so that the pre-swath line and after compensation give the general desired group the optimal delay response. This can be done by using the reference system (described above) and simulation modeling of the nominal expected characteristics of the TP line in which the Line conditioner and the conditioner circuit characteristics will be used in the downstream signal chain to select the component values ​​that provide the desired response.

[0043] When considering the upstream signal path, as previously described the upstream mixer 110 can be substantially the same as the lower mixer 120; and the external amplifier 112 may be substantially the same as in the lower preamplifier 122, except that before the amplifier may be equipped with different gain and damping settings. Since the upward signal is in the lower frequency range, it is not the same as the degradation of the cell signal, it can have a significantly different level.

[0044] From the upper preamplifier 112, the single-ended upstream signal, the positive and negative preamble output lines 400, 402 are supplied to the inlet filter 114, which, hereinafter the filter, preferably includes two mono-jute filters rather than a differential filter. Preferred embodiments of the inlet filter 114 and the amplifier upstream of the controller 116 are illustrated in Figure 9. As shown, the above filter preferably includes two identical 11th order elliptic low-pass filters 410P and 410n. The filters can be designed to have a sharp cut-off frequency just above the upper limit of the upper frequency range at 125 kHz, and also provide a high rejection of 80 dB to 90 dB to out-of-band further

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The frequency, which can be connected through the mixer 110 and the preamplifier 112, the capacitive and inductive values ​​indicated in Figure 9 are exemplary values ​​that provide low-pass filters with the desired properties. At 50 ohm input resistors 412 and 50 ohm filter output resistors 414 are for impedance matching. Individual-ended outputs Filters 410P and 410n on lines 420 and 422 are provided in the form of a differential difference input signal before the preamplifier 116. A positive output line 420 with a 410P filter can be used as shown at 424 and brought to peak current detector 166 ( see Figure 1).

[0045] Prior to the amplifier, the controller 116 in a preferred embodiment consists of a differential amplifier that can have the same structure as the further controller amplifier 128. As shown, the upper amplifier includes a pair of connected opamps 440 and 442 with their non-inverting inputs connected positive and negative outputs 420 and 422, respectively, with filter. Couplings resistors 446 and 448 can be connected between outputs 140 and 142, respectively, and the inverting opamps input as shown. The reversing inputs can be further connected to each other via a network, RC comprising a resistor 450 and a capacitor 452, as shown. The differential output of the upper preamplifier on lines 140 and 142 is supplied to a further mixer 120 as previously described.

[0046] As described above, in order to achieve the desired target, with high efficiency, it is desirable to keep the added noise due to the components of the systems and components as low as possible, to use the designs, systems and structures for the various nutrient circuits a line that has become a high-frequency attenuation signal and low crosstalk. In addition, since the operating power on the line conditioner preferably comes from the TP line on which it is used, it is desirable to keep the operating power low in order to avoid interference in normal telephone service. In order to achieve the desired goals low noise and low power consumption, careful selection of low noise, high throughput and low power plant opamps is necessary. Furthermore, it is desirable to minimize ohmic disturbances by using low values ​​of resistors in the circuit.

[0047] Filters above and below are also important for achieving high efficiency. This is especially true for the downstream filter from higher signal frequencies for further degradation and loss experiments than to make upstream signals. The type of filter selected and its trailing characteristics not only affects the amount of unwanted out-of-band frequencies that can contribute to raising the noise level, the filter also affects the group delay through the signal path. The 11th order elliptical filters used in the invention have the desired sharp damping and high blocking characteristics, and their circuit components can be easily adapted to provide the required group delay characteristics. As will be appreciated, however, it may be used to achieve the objectives of the invention,

[0048] Standard ADSL protocol uses Reed Solomon (RS) error correction codes that have properties to reconstruct lost data. The amount of lost data that

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EP 1 964 377 B1 may be reproduced depending on the number of bits used RS for error correction. If the data loss exceeds a predetermined amount of correctible RS data, the violation code does not occur and the data remains unrecoverable. An alternative measure of data loss is the bit error rate ("BER"). video The TV uses the lost / IP UDP protocol. Because the number of RS codes increases pixilation increases, the groups of video pixels are lost. This can cause a "glow" in the video image. data integrity is a function of both the quality of the SNR signal. High data rates that can be reached with high signal levels do not necessarily provide high data integrity, because the signal quality may deteriorate due to sufficient group delays, crosstalk and interference between the symbol to produce an undesirable level of code breaking. The invention uses a series of code and / or BER violations as a measure of high performance and optimizes performance by reducing the number of Code or BER violations to an acceptable level.

[0049] The gain-dampening level of the preamplifier 112 and 122, which are stored in Table 2-D, can be experimentally obtained from these values ​​to produce optimal efficiency for a given set of linear conditions. Accordingly, the detected voltage peak from peak detectors 162 and 166 for downstream and upstream signals are an effective measure of TP line characteristics between CPE, CO and line conditioner. Based on the detected values, which is equivalent to the effective impedance or distance, the values ​​empirically in Table 2-D can be selected to set positive and damping to provide the desired optimal performance level for the given TP line in which the line nutrient is employed. It has been found that, according to the invention, focusing on optimal performance, maximizing SNR and optimizing signal quality,

While the invention has been above with reference to specific embodiments of the invention, it will be understood by those skilled in the art that variations to these embodiments may be made without departing from the subjects of the invention, the scope of which is determined by the attached claims.

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EP 1 964 377 B1