Apparatus and method for transmitting and receiving a signal

ES 2 608 553 T3

DESCRIPTION

Apparatus and method for transmitting and receiving a signal

Background of the invention

The present invention relates to a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal, and more particularly to a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal that are capable of improve the efficiency of data transmission.

Description of Related Art

As a broadcast technology has been developed, users have received a high definition (HD) moving image. With the continuous development of a compression algorithm and the high performance of the hardware, a better environment will be provided to the users in the future. A digital television (DTV) system can receive a broadcast digital signal and provide a variety of supplementary services to users, as well as a video signal and an audio signal.

Digital Video Broadcasting (DVB) -C2 is the third specification to join the DVB family of second generation transmission systems. Developed in 1994, today DVB-C is deployed on more than 50 million cable tuners around the world. In line with the other second generation DVB systems, DVB-C2 uses a combination of Low Density Parity Check (LDPC) and BCH codes. This powerful No Return Channel Error Correction (FEC) provides an improvement of around 5dB in carrier-to-noise ratio over DVB-C. Suitable bit interleaving schemes optimize the overall robustness of the FEC system. Spread across a header, these frames are called Physical Layer Pipes (PLP). One or more of these PLPs are multiplexed into a data segment. Two-dimensional interleaving (in the time and frequency domains) is applied to each segment, allowing the receiver to eliminate the impact of burst decays and frequency selective interference, such as single frequency input.

With the development of these digital broadcast technologies, the demand for a service such as a video signal and an audio signal increased and the size of the data desired by users or the number of broadcast channels gradually increased. The DVB project “Frame structure channel and modulation for a second generation terrestrial television broadcasting system (DVB-T2)”, Digital Video Broadcasting [online] June 2008 (06-01-2008) describes structure channel coding techniques raster for broadcast systems.

Compendium of the invention

Any occurrence of the term "embodiment" in the description is to be considered as an "aspect of the invention", the invention being defined in the accompanying independent claims. Accordingly, the present invention is directed to a method for transmitting and receiving a signal and an apparatus for transmitting and receiving a signal, which substantially obviates one or more problems due to limitations and disadvantages of the related art.

An object of the present invention is to provide a method of transmitting a broadcast signal to a receiver according to claim 1.

Another aspect of the present invention provides a method of receiving a broadcast signal according to claim 9.

Still another aspect of the present invention provides a transmitter for transmitting a broadcast signal to a receiver according to claim 5.

Still another aspect of the present invention provides a receiver for receiving a broadcast signal according to claim 10.

Brief description of the drawings

The accompanying drawings, which are included to provide a further understanding of the invention and which are incorporated in and constitute a part of this application, illustrate one embodiment (s) of the invention and, together with the description, serve to explain the principle of invention. In the drawings:

Fig. 1 is an example of Quadrature 64 Amplitude Modulation (QAM) used in European DVB-T.

Fig. 2 is a Binary Reflected Gray Code (BRGC) method.

Fig. 3 is a close to Gaussian modifying 64-QAM output used in DVB-T.

ES 2 608 553 T3

Fig. 4 is the Hamming distance between pairs reflected in the BRGC.

Fig. 5 are the characteristics in QAM where the reflected pair exists for each I axis and Q axis.

Fig. 6 is a QAM modification method using a reflected pair of BRGC.

Fig. 7 is an example of modified 64/256/1024/4096-QAM.

Figs 8 to 9 are an example of modified 64-QAM using a BRGC Mirrored Pair.

Figs. 10-11 are an example of modified 256-QAM using a BRGC Mirrored Pair.

Figs. 12-13 are an example of modified 1024-QAM using a BRGC Mirrored Pair (0-511).

Figs. 14-15 are an example of modified 1024-QAM using a BRGC Mirrored Pair (512-1023).

Figs. 16-17 are an example of modified 4096-QAM using a BRGC Reflected Pair (0-511).

Figs. 18 to 19 are an example of modified 4096-QAM using a BRGC Mirrored Pair (512-1023).

Figs. 20-21 are an example of modified 4096-QAM using a BRGC Mirrored Pair (1024-1535).

Figs. 22-23 are an example of modified 4096-QAM using a BRGC Reflected Pair (1536-2047).

Figs. 24-25 are an example of modified 4096-QAM using a BRGC Reflected Pair (2048-2559).

Figs. 26-27 are an example of modified 4096-QAM using a BRGC Reflected Pair (2560-3071).

Figs. 28 to 29 are an example of modified 4096-QAM using a BRGC Mirrored Pair (3072-3583).

Figs. 30-31 are an example of modified 4096-QAM using a BRGC Reflected Pair (3584-4095).

Fig. 32 is an example of a modified QAM bitmap where 256-QAM is modified using BRGC.

Fig. 33 is an example of MQAM transformation into a non-uniform constellation.

Fig. 34 is an example of a digital transmission system.

Fig. 35 is an example of an input processor.

Fig. 36 is information that can be included in Baseband (BB).

Fig. 37 is an example of BICM.

Fig. 38 is an example of a shortened / punched encoder.

Fig. 39 is an application example of various constellations.

Fig. 40 is another example of cases where compatibility between conventional systems is considered.

Fig. 41 is a frame structure comprising a preamble for L1 signaling and a data symbol for PLP data.

Fig. 42 is an example of a frame builder.

Fig. 43 is an example of the pilot insert (404) shown in Fig. 4.

Fig. 44 is a structure of SP.

Fig. 45 is a new structure SP or Pilot Pattern (PP) 5 '.

Fig. 46 is a suggested PP5 'structure.

Fig. 47 is a relationship between data symbol and preamble.

Fig. 48 is another relationship between data symbol and preamble.

Fig. 49 is an example of a cable channel delay profile.

Fig. 50 is a scattered pilot structure using z = 56 and z = 112.

ES 2 608 553 T3

Fig. 51 is an example of an OFDM-based modulator.

Fig. 52 is an example of a preamble structure.

Fig. 53 is an example of Preamble decoding.

Fig. 54 is a process for designing a more optimized preamble.

Fig. 55 is another example of a preamble structure.

Fig. 56 is another example of preamble decoding.

Fig. 57 is an example of a preamble structure.

Fig. 58 is an example of L1 decoding.

Fig. 59 is an example of an analog processor.

Fig. 60 is an example of a digital receiver system.

Fig. 61 is an example of an analog processor used in the receiver.

Fig. 62 is an example of a demodulator.

Fig. 63 is an example of a frame parser.

Fig. 64 is an example of a BICM demodulator.

Fig. 65 is an example of LDPC decoding using shortening / punching.

Fig. 66 is an example of an output processor.

Fig. 67 is an example of 8 MHz L1 block repetition rate.

Fig. 68 is an example of 8 MHz L1 block repetition rate.

Fig. 69 is a new L1 block repetition rate of 7.61 MHz.

Fig. 70 is an example of L1 signaling that is transmitted in a frame header.

Fig. 71 is a preamble and L1 structure simulation result.

Fig. 72 is an example of a symbol interleaver.

Fig. 73 is an example of a L1 block transmission.

Fig. 74 is another example of L1 signaling transmitted within a frame header.

Fig. 75 is an example of frequency or time interleaving / deinterleaving.

Description of preferred embodiments

Reference will now be made in detail to preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings. Wherever possible, the same reference numerals will be used throughout the drawings to refer to the same or like parts.

In the following description, the term "service" is indicative of any of the broadcast content that can be transmitted / received by the signal transmitting / receiving apparatus.

Quadrature Amplitude Modulation (QAM), using Binary Reflected Gray Code (BRGC) is used as modulation in a broadcast transmission environment, where conventional Bit Interleaved Coded Modulation (BICM) is used. Fig. 1 shows an example of 64-QAM used in European DVB-T.

The BRGC can be made using the method shown in Fig. 2. An n-bit BRGC can be made by adding a BRGC reverse code of (n-1) bits (ie, reflected code) to a backing of (n- 1) bits, adding zeros to a leading part of the original (n-1) bit BRGC and adding ones to a leading part of the reflected code. The BRGC code made by this method has a Hamming distance between adjacent codes of one (1). Also, when a BRGC is applied to QAM, the Hamming distance between a point and the four points that are most closely adjacent to the point is one (1) and the Hamming distance between the point and four other points which are the seconds more

ES 2 608 553 T3 closely adjacent to the point is two (2). Such characteristics of Hamming distances between a specific constellation point and other adjacent points can be referred to as a Gray correlation rule in QAM.

To make a system robust against Additive Gaussian White Noise (AWGN), the distribution of signals transmitted from a transmitter can be made close to a Gaussian distribution. To be able to do that, the locations of points in the constellation can be modified. Fig. 3 shows a close Gaussian output modifying the 64-QAM used in DVB-T. Such a constellation can be called Non-uniform QAM (NU-QAM).

To form a constellation of Non-uniform QAM, a Gaussian Cumulative Distribution Function (CDF) can be used. In the case of 64, 256 or 1024 QAM, that is, 2^ΛΝ AM, the QAM can be divided into two independent N-PAMs. By dividing the Gaussian CDF into N sections of identical probability and allowing a signal point in each section to represent the section, a constellation can be made that has a Gaussian distribution. In other words, the xj coordinate of a newly defined non-uniform N-MAP can be defined as follows:

<img file="ES2608553T3_D0001.tif" />

(Eq. 1)

Fig. 3 is an example of transformation of DVB-T 64QAM into NU-64QAM using the above methods. Fig. 3 represents a result of modifying the coordinates of each I axis and Q axis using the above methods and correlating the previous constellation points to newly defined coordinates. In the case of 32, 128 or 512 QAM, that is, cross QAM, which is not 2^ΛΝ QAM, by modifying Pj appropriately, a new coordinate can be found.

An embodiment of the present invention can modify QAM using BRGC using BRGC features. As shown in Fig. 4, the Hamming distance between the Reflected Pair in BRGC is one because it differs by only one bit that is added to the front of each code. Fig. 5 shows the characteristics in QAM where there is a Reflected Pair for each I axis and Q axis. In this figure, there is a Reflected Pair on each side of the black dotted line.

Using existing Reflected Pairs in QAM, an average power of a QAM constellation can be reduced while maintaining the Gray correlation rule in QAM. In other words, in a constellation where an average power is normalized to 1, the minimum Euclidean distance in the constellation can be increased. When this modified QAM is applied to broadcast or communication systems, it is possible to implement either a more robust system against noise that uses the same power as a conventional system or a system with the same performance as a conventional system, but uses less energy.

Fig. 6 shows a QAM modification method using a BRGC Mirrored Pair. Fig. 6a shows a constellation and Fig. 6b shows a flow chart for modifying QAM using a BRGC Reflected Pair. First of all, you need to find a destination point that has the highest power among the constellation points. Candidate points are points where the target point can move and are the closest neighboring points to the reflected pair of the target point. Then, it is necessary to find an empty point (that is, a point not yet taken by other points) that has the lowest power among the candidate points and the power of the target point and the power of a candidate point are compared. If the power of the candidate point is less, the target point moves to the candidate point. These processes are repeated until an average power of the points in the constellation reaches a minimum, while maintaining the Gray correlation rule.

Fig. 7 shows an example of a modified 64/256/1024/4096 QAM. The correlated Gray values ​​correspond to Figs. 8 to 31, respectively. In addition to these examples, other types of modified QAM can be performed that allow identical power optimization. This is because a destination point can move to multiple candidate points. The suggested modified QAM can be applied to not only 64/256/1024/4096-QAM, but also to cross QAM, to a larger QAM or to modulations that use another BRGC other than QAM.

Fig. 32 shows an example of Modified QAM bitmap where 256-QAM is modified using a BRGC. Fig. 32a and Fig. 32b show the Most Significant Bit (MSB) correlation. Points indicated as solid circles represent correlations of ones and points indicated as blank circles represent correlations of zeros. In the same way, each bit is mapped as shown in figures from (a) to (h) in Fig. 32, until the Least Significant Bits (LSB) are correlated. As shown in Fig. 32, a Modified QAM can allow bit decision using only the I or Q axes like a conventional QAM, except for one bit that is close to the MSB (Fig. 32c and Fig. 32d). Using these features, a simple receiver can be made by partially modifying a receiver for QAM. An efficient receiver can be implemented by checking both values ​​of I and Q

ES 2 608 553 T3 only when determining the bit next to the MSB and calculating only I or Q for the rest of the bits. This method can be applied to the Approximate LLR, the Exact LLR or the Hard decision.

Using Modified QAM or MQAM, which uses the characteristics of the previous BRGC, a non-uniform constellation or NU-MQAM can be made. In the above equation, where a Gaussian CDF is used, Pj can be modified to fit MQAM. Like QAM, in MQAM, two PAMs can be considered that have an I axis and a Q axis. However, unlike QAM where a number of points corresponding to a value of each PAM axis are identical, the number of points changes in MQAM. If nj is defined as a number of points that corresponds to the value of order j of a PAM in an MQAM where there are a total of M constellation points, then Pj can be defined as follows:

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<img file="ES2608553T3_D0003.tif" />

fio = ° (Eq. 2)

Using the newly defined Pj, MQAM can be transformed into a Non-uniform constellation. Pj can be defined as follows for the 256-MQAM example.

2.5 10 22 36 51 67 84 102 119.5 136.5 154 172 189 205 220 234 246 253.5]

256 * 256'256'256'256'256'256'256 '256' 256 '256'256'256'256'256'256'256' 256 J

Fig. 33 is an example of MQAM transformation into a Non-uniform constellation. NU-MQAM made using these methods can preserve characteristics of MQAM receivers with modified coordinates of each PAM. In this way, an efficient receiver can be implemented. In addition, a more robust system against noise than the previous NU-QAM can be implemented. For a more efficient diffusion transmission system, hybridization of MQAM and NU-MQAM is possible. In other words, a more robust system against noise can be implemented using MQAM for an environment where a high code rate error correcting code is used and using NU-MQAM otherwise. For such a case, a transmitter can let a receiver have code rate information of a currently used error correction code and a currently used modulation type so that the receiver can demodulate according to the currently used modulation.

Fig. 34 shows an example of a digital transmission system. The inputs may comprise a number of MPEG-TS or GSE (General Stream Encapsulation) streams. An input processor module 101 can add transmission parameters to the input stream and perform scheduling for a BICM module 102. The BICM module 102 can add redundancy and interleave data for transmission channel error correction. A frame builder 103 can form frames by adding pilot and physical layer signaling information. A modulator 104 can perform modulation on input symbols in efficient methods. An analog processor 105 can perform various processes to convert digital input signals to analog output signals.

Fig. 35 shows an example of an input processor. The input MPEG-TS or GSE stream can be transformed by the input preprocessor into a total of n streams to be processed independently. Each of these streams can be either a full TS frame, which includes multiple service components, or a minimal TS frame, which includes a service component (ie, video or audio). Furthermore, each of these flows can be a GSE flow that transmits either multiple services or a single service.

The input interface module 202-1 can allocate a certain number of input bits equal to the maximum data field capacity of a Baseband (BB) frame. A padding can be inserted to complete the LDPC / BCH code block capacity. The input stream synchronization module 203-1 may provide a mechanism to regenerate the Transport Stream (or Generic Stream in packets) clock at the receiver to ensure constant bit rates and delay of end to end.

In order to allow the Transport Stream to recombine without requiring additional memory in the receiver, the incoming Transport Streams are delayed by the delay compensators 204-1 ~ n considering the interleaving parameters of the data PLPs in a group. and the corresponding common PLP. Null packet eraser modules 205-1 ~ n can increase transmission efficiency by extracting the inserted null packet for a VBR (variable bit rate) service case. Cyclic Redundancy Check (CRC) encoder modules 206-1 ~ n can add CRC parity to increase the transmission reliability of the BB frame. BB inserter modules 207-1 ~ n can add a BB frame header at the beginning of the BB frame. The information that can be included in the header in BB is shown in Fig. 36.

A merger / segmenter module 208 can perform BB frame segmentation from each PLP, merging BB frames from multiple PLPs and scheduling each BB frame into a transmission frame.

ES 2 608 553 T3

Therefore, the merger / segmenter module 208 can output L1 signaling information that relates to PLP allocation in a frame. Finally, a BB encoder module 209 can randomize input bit streams to minimize inter-bit correlation within bit streams. The shaded modules in Fig. 35 are modules used when the transmission system uses a single PLP and the other modules in Fig. 35 they are modules used when the transmitting device uses multiple PLPs.

Fig. 37 shows an example of a BICM module. Fig. 37a shows a data path and Fig. 37b shows an L1 path of a BICM module. An external encoder module 301 and an internal encoder module 303 can add redundancy to input bit streams for error correction. An external interleaver module 302 and an internal interleaver module 304 can interleave bits to prevent burst errors. The external interleaver module 302 can be omitted if the BICM is specifically for DVB-C2. A bit demultiplexer module 305 can monitor the reliability of each bit output from the internal interleaver module 304. A symbol mapper module 306 can map input bit streams to symbol streams. At this time, it is possible to use any between a conventional QAM, an MQAM that uses the BRGC mentioned above for performance improvement, a NU-QAM that uses Non-uniform modulation or a NU_MQAM that uses the BRGC applied to Non-uniform modulation for improvement. performance. To build a system that is more robust against noise, combinations of modulations using MQAM and / or NU-MQAM can be considered depending on the code rate of the error correction code and the capacity of the constellation. At this time, the Symbol correlator module 306 can use a suitable constellation according to the code rate and capacity of the constellation. Fig. 39 shows an example of such combinations.

Case 1 shows an example of using only NU-MQAM at low code rate for a simplified system implementation. Case 2 shows an example of using constellation optimized at each code rate. The transmitter can send information about the code rate of the error correction code and the constellation capability to the receiver so that the receiver can use a suitable constellation. Fig. 40 shows another example of cases where compatibility between conventional systems is considered. In addition to the examples, additional combinations are possible to optimize the system.

The ModCod 307 Header Inserter module shown in Fig. 37 can take adaptive coding and modulation (ACM) / variable coding and modulation (VCM) feedback information and add parameter information used in coding and modulation to a block of FEC as the header. The modulation / code rate header (ModCod) can include the following information:

* FEC type (1 bit) - LDPC long or short * Code rate (3 bits) * Modulation (3 bits) - up to 64K QAM * PLP identifier (8 bits)

Symbol interleaver module 308 can perform symbol domain interleaving for additional interleaving effects. Similar processes performed on the data path can be performed on the L1 signaling path but with possibly different parameters (301-1-308-1). At this point, a shortened / punctured code module (303-1) can be used for the internal code.

Fig. 38 shows an example of LDPC encoding using shortening / puncturing. The shortening process can be performed on input blocks that have fewer bits than a required number of bits for LDPC encoding, since many zero bits required for LDPC encoding can be padded (301c). Zero-padded input bit streams can have parity bits through LDPC (302c) encoding. At this time, for bit streams corresponding to original bit streams, zeros (303c) can be extracted and for parity bit streams, code rates can be punched (304c). These information bit streams and processed parity bit streams can be multiplexed to original streams and output (305c).

Fig. 41 shows a frame structure comprising a preamble for L1 signaling and a data symbol for PLP data. It can be seen that the preamble and data symbols are generated cyclically, using a frame as a unit. Data symbols comprise PLP type 0, which is transmitted using fixed modulation / encoding, and PLP type 1, which is transmitted using variable modulation / encoding. For PLP type 0, information such as modulation, FEC type and FEC code rate, are transmitted in the preamble (see Fig. 42 - frame header insert 401). For PLP type 1, corresponding information can be transmitted in the FEC block header of a data symbol (see Fig. 37 - ModCod 307 header insert). By separating PLP types, ModCod oversizing can be reduced by 3-4% of a total transmission rate, for PLP type 0, which is transmitted at a fixed bit rate. In a receiver, for a PLP type 0 fixed coding / modulation PLP, the r401 frame header extractor shown in Fig. 63 can extract

ES 2 608 553 T3 information on FEC modulation and code rate and provide the extracted information to a BICM decoding module. For PLP type 1 variable encoding / modulation PLP, the ModCod extraction modules r307 and r307-1 shown in Fig. 64 can extract and provide the necessary parameters for BICM decoding.

Fig. 42 shows an example of a frame former. A frame header inserter module 401 may form a frame from input symbol streams and may add a frame header at the front of each transmitted frame. The frame header can include the following information:

* Number of joined channels (4 bits) * Guard interval (2 bits) * PAPR (2 bits) * Pilot pattern (2 bits) * Digital system identification (16 bits) * Frame identification (16 bits) * Length of frame (16 bits) - number of Orthogonal Frequency Division Multiplexing (OFDM) symbols per frame * Super frame length (16 bits) - number of frames per super frame * number of PLPs (8 bits) * for each PLP

PLP identification (8 bits)

Channel link identifier (4 bits) PLP start (9 bits) PLP type (2 bits) - common PLP or other PLP payload type (5 bits) MC type (1 bit) - modulation and fixed encoding /variable

If MC type == modulation and fixed encoding FEC type (1 bit) - LDPC long or short

Code rate (3 bits)

Modulation (3 bits) - up to 64K QAM end yes;

Number of notch channels (2 bits) for each notch

Notch start (9 bit)

Notch width (9 bits) end for;

PLP width (9 bits) - maximum number of PLP FEC blocks PLP time interleaved type (2 bits) end stop;

* CRC-32 (32 bits)

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A channel bonding environment is assumed for the L1 information transmitted in the frame header and the data corresponding to each data segment is defined as PLP. Therefore, information such as the PLP identifier, the channel bonding identifier and the PLP start address are required for each channel used in the bonding. One embodiment of this invention suggests transmitting the ModCod field in the FEC frame header if the PLP type supports variable modulation / encoding and transmitting the ModCod field in the frame header if the PLP type supports fixed modulation / encoding to reduce overhead. Signaling. Furthermore, if there is a notch band for each PLP, transmitting the starting direction of the notch and its width, decoding of the corresponding carriers in the receiver may become unnecessary.

Fig. 43 shows an example of Pilot Pattern 5 (PP5) applied in a channel bonding environment. As shown, if the SP positions are coincident with preamble pilot positions, an irregular pilot structure can occur.

Fig. 43a shows an example of the pilot insertion module 404 as shown in Fig. 42. As shown in Fig. 43, if a single frequency band (eg 8 MHz) is used, the width available band is 7.61 MHz, but if multiple frequency bands are joined together, guard bands can be removed, thus, you can extremely increase the frequency efficiency. Fig. 43b is an example of the preamble inserter module 504 as shown in Fig. 51, which is transmitted in the front of the frame and, even with channel bonding, the preamble has a repetition rate of 7.61 MHz, which is the bandwidth of the L1 block. This is a structure that considers the bandwidth of a tuner that performs the initial channel scan.

There are Pilot Patterns for both the Preamble and Data Symbols. For data symbol, scattered pilot (SP) patterns can be used. T2 Pilot Pattern 5 (PP5) and Pilot Pattern 7 (PP7) may be good candidates for frequency-only interpolation. The PP5 has x = 12, y = 4, z = 48 for GI = 1/64 and the PP7 has x = 24, y = 4, z = 96 for GI = 1/128. An additional temporal interpolation is also possible for a better channel estimation. The pilot patterns for the preamble can cover all possible pilot positions for the initial channel acquisition. Furthermore, the preamble pilot positions should be coincident with the SP positions and a single pilot pattern is desired for both the preamble and the SP. The preamble pilots could also be used for time interpolation and each preamble could have an identical pilot pattern. These requirements are important for C2 detection in scanning and necessary for frequency offset estimation with randomization sequence correlation. In a channel bonding environment, coincidence in the channel bonding pilot positions should also be maintained because the uneven pilot structure can degrade interpolation performance.

In detail, if a distance z between scattered pilots (SP) in an OFDM symbol is 48 and if a distance y between the SPs corresponding to a specific SP carrier along the time axis is 4, an effective distance x after the time interpolation becomes 12. This is when a fraction of the guard interval (GI) is 1/64. If the fraction of the IG is 1/128, we can use x = 24, y = 4, and z = 96. If channel bonding is used, the SP positions can be made coincident with the preamble pilot positions by generating non-continuous points in the spread pilot structure.

At this time, the preamble pilot positions may be coincident with each data symbol SP position. When channel bonding is used, the data segment on which a service is transmitted can be determined regardless of the 8 MHz bandwidth granularity. However, to reduce the overhead for addressing data segments , you can choose a transmission that starts from the SP position and ends at the SP position.

When a receiver receives such SPs, if necessary, the channel estimation module r501 shown in Fig. 62 can perform time interpolation to obtain the pilots shown in dashed lines in Fig. 43 and perform frequency interpolation. At this time, for non-continuous points of which intervals are indicated as 32 in Fig. 43, you can implement either perform left and right interpolations separately or perform interpolations only on one side, then perform interpolation on the other side using the already interpolated pilot positions of which the interval is 12, as a reference point . At this time, the width of the data segment can vary within 7.61 MHz, in this way, a receiver can minimize power consumption by performing a channel estimation and decoding only the necessary subcarriers.

Fig. 44 shows another example of PP5 applied in channel bonding environment or SP structure to maintain effective distance x as 12 to avoid irregular SP structure shown in Fig. 43 when channel bonding is used. Fig. 44a is an SP structure for data symbol and Fig. 44b is an SP structure for preamble symbol.

As shown, if the SP distance is kept consistent in case of channel bonding, there will be no problem in frequency interpolation, but the pilot positions between a data symbol and a preamble may not be coincident. In other words, this structure does not require additional channel estimation for an irregular SP structure, however, the SP positions used in channel bonding and the preamble pilot positions reach

ES 2 608 553 T3 be different for each channel.

Fig. 45 shows a new structure of SP or PP5 ', to provide a solution to the two aforementioned problems in the channel bonding environment. Specifically, a pilot distance of x = 16 can solve those problems. To preserve the pilot density or to maintain the same overhead, a PP5 'can have x = 16, y = 3, z = 48 for GI = 1/64 and a PP7' can have x = 16, y = 6, z = 96 for GI = 1/128. The frequency-only interpolation capability can still be maintained. The pilot positions are depicted in Fig. 45 for comparison with the PP5 structure.

Fig. 46 shows an example of a new SP Pattern or PP5 'structure, in the channel bonding environment. As shown in Figure 46, if either a single channel or channel bonding is used, an effective pilot distance x = 16 can be provided. Also, because the SP positions can be made coincident with the SP positions. Preamble pilot, channel estimation deterioration caused by SP irregularity or mismatched SP positions can be avoided. In other words, there is no irregular SP position for the frequency interpolator and a match is provided between the preamble and the SP positions.

Accordingly, the proposed new SP patterns may be advantageous in that a single SP pattern can be used for both single channel and bonded channel; no irregular pilot structure can be made and thus a good channel estimation is possible; both the preamble and SP pilot positions can be kept coincident; the pilot density can be kept the same as for PP5 and PP7, respectively; and the frequency-only interpolation capability can also be preserved.

Furthermore, the preamble structure can satisfy requirements such as that the preamble pilot positions should cover all possible SP positions for initial channel acquisition; the maximum number of carriers should be 3409 (7.61 MHz) for the initial scan; Exactly the same pilot patterns and scrambling sequence should be used for C2 detection; and no specific detection preamble is required like P1 in T2.

In terms of relation to the frame structure, the granularity of the position of the data segment can be modified to 16 carriers instead of 12, in this way, there can be less overhead for position addressing and no other problem can be expected. regarding the data segment condition, null slot condition, etc.

Therefore, in the channel estimation module r501 of Fig. 62, pilots can be used in each preamble when performing data symbol SP time interpolation. Therefore, channel acquisition and channel estimation at frame boundaries can be improved.

Now, regarding the requirements related to the preamble and the pilot structure, there is consensus that the positions of the preamble pilots and the SPs should coincide regardless of the union of channels; the number of total carriers in the L1 block should be divisible by the pilot distance to avoid an irregular structure at the edge of the band; the L1 blocks should be repeated in the frequency domain; and the L1 blocks should always be decodable at an arbitrary tuner window position. Additional requirements would be that the pilot positions and patterns should be repeated over an 8 MHz period; correct carrier frequency offset should be estimated without knowledge of channel bonding; and decoding (reordering) of L1 is impossible before the frequency offset is compensated.

Fig. 47 shows a relationship between data symbol and preamble when using preamble structures as shown in Fig. 52 and Fig. 53. The L1 block can be repeated over a 6 MHz period. For decoding L1, both the frequency offset and the preamble shift pattern should be found. Decoding of L1 is not possible at an arbitrary tuner position without channel bonding information and a receiver cannot differentiate between the preamble shift value and the frequency offset.

In this way, a receiver, specifically for the r401 frame header extractor shown in Fig. 63 to perform L1 signal decoding, needs to obtain the channel bonding structure. Because the amount of preamble change expected in the two vertically shaded regions in Fig. 47 is known, the time / frequency synchronization module r505 in Fig. 62 can estimate the carrier frequency offset. Based on the estimate, the L1 signaling path (r308-1 ~ r301-1) in Fig. 64 can decode L1.

Fig. 48 shows a relationship between a data symbol and a preamble when using a preamble structure as shown in Fig. 55. The L1 block can be repeated in a period of 8 MHz. For L1 decoding, only the frequency offset needs to be found and knowledge of channel bonding may not be required. The frequency offset can be easily estimated using the known Pseudo-Random Binary Sequence (PRBS). As shown in Fig. 48, the preamble and data symbols are aligned, thus, an additional sync search may become unnecessary. Therefore, for a receiver, specifically

ES 2 608 553 T3 for the r401 frame header extractor module shown in Fig. 63, it may only be necessary to obtain the peak correlation with the pilot scrambling sequence to perform L1 signal decoding. The time / frequency synchronization module r505 in Fig. 62 can estimate the carrier frequency offset from the peak position.

Fig. 49 shows an example cable channel delay profile.

From a pilot design standpoint, the current GI already overprotects cable channel delay spread. In the worst case, redesign of the channel model may be an option. To repeat the pattern exactly every 8 MHz, the pilot distance should be a divisor of 3584 carriers (z = 32 or 56). A pilot density of z = 32 can increase the pilot overload, thus z = 56 can be chosen. Slightly less delay coverage may not be important on a cable channel. For example, it can be 8 ps for PP5 'and 4 ps for PP7', compared to 9.3 ps (PP5) and 4.7 ps (PP7). Significant delays can be covered by both pilot patterns even in the worst case. For a preamble pilot position, no more than all SP positions are required in a data symbol.

If the -40 dB delay path can be ignored, the actual delay spread can be 2.5 ps, 1/64 GI = 7 ps, or 1/128 GI = 3.5 ps. This shows that the pilot distance parameter, z = 56, can be a pretty good value. Furthermore, z = 56 can be a convenient value for structuring the pilot pattern that enables the preamble structure shown in Fig. 48.

Fig. 50 shows a sparse pilot structure using z = 56 and z = 112 that is built into pilot insert module 404 in Fig. 42. A PP5 '(x = 14, y = 4, z = 56) and a PP7 '(x = 28, y = 4, z = 112). Edge carriers could be inserted for the closing edge.

As shown in Fig. 50, the pilots are aligned 8 MHz from each edge of the band, each pilot position and pilot structure can be repeated every 8 MHz. In this way, this structure can support the preamble structure. shown in Fig. 48. In addition, a common pilot structure can be used between the preamble and the data symbols. Therefore, the channel estimation module r501 in Fig. 62 It can perform channel estimation using interpolation on the preamble and data symbols, because no irregular pilot pattern may occur, regardless of the window position that is decided by the data segment locations. At this time, the use of frequency interpolation alone may be sufficient to compensate for channel distortion from delay propagation. If time interpolation is additionally performed, a more accurate channel estimation can be performed.

Accordingly, in the proposed new pilot pattern, the position and the pilot pattern can be repeated based on a period of 8 MHz. A single pilot pattern can be used for both the preamble and the data symbols. Decoding of L1 can always be possible without knowledge of channel bonding. Furthermore, the proposed pilot pattern may not affect the parts in common with T2 because the same pilot strategy of the sparse pilot pattern can be used; T2 already uses 8 different pilot patterns; and no significant receiver complexity may be increased by the modified pilot patterns. For a pilot randomization sequence, the PRBS period can be 2047 (m sequence); PRBS generation can be restarted every 8 MHz, of which the period is 3584; the pilot repetition rate of 56 can also be co-premium with 2047; and no PAPR problem can be expected.

Fig. 51 shows an example of an OFDM-based modulator. The input symbol streams can be transformed into the time domain by IFFT module 501. If necessary, the peak-to-average power ratio (PAPR) can be reduced in PAPR reduction module 502. For PAPR methods , an active constellation extension (ACE) or tone reservation can be used. The GI inserter 503 can copy a last part of the effective OFDM symbol to fill the guard interval as a cyclic prefix.

The preamble inserter module 504 can insert the preamble at the front of each transmitted frame, so that a receiver can detect a digital signal, frame, and acquire a time / frequency offset acquisition. At this time, the preamble signal can perform physical layer signaling, such as FFT size (3 bits) and guard interval size (3 bits). The preamble insert module 504 can be omitted if the modulator is specifically for DVB-C2.

Fig. 52 shows an example of a preamble structure for channel bonding, generated in preamble insert module 504 in Fig. 51. A complete L1 block should be "always decodable" at any arbitrary tuner window position. 7.61 MHz and no loss of L1 signaling should occur regardless of the tuner window position. As shown, the L1 blocks can be repeated in the frequency domain in a period of 6 MHz. The data symbols can be channel-linked for every 8 MHz. If, for L1 decoding, a receiver uses a tuner such as the tuner r603 shown in Fig. 61, which uses a bandwidth of 7.61 MHz, frame header extractor r401 in Fig. 63 needs to rearrange block L1

ES 2 608 553 T3 cyclic changed received (Fig. 53) to its original form. This rearrangement is possible because the L1 block repeats for each 6 MHz block. Fig. 53a can be rearranged to Fig. 53b.

Fig. 54 shows a process for designing a more optimized preamble. The preamble structure of Fig. 52 only uses 6 MHz of the total tuner bandwidth of 7.61 MHz for L1 decoding. In terms of spectral efficiency, the 7.61 MHz tuner bandwidth is not fully utilized. Therefore, there may be a further optimization in spectral efficiency.

Fig. 55 shows another example of preamble structure or preamble symbol structure for full spectral efficiency, generated in the frame header insertion module 401 in Fig. 42. Like the data symbol, the blocks L1s can be repeated in the frequency domain over a period of 8 MHz. An entire L1 block is still "always decodable" at any arbitrary 7.61 MHz tuner window position. After tuning, the 7.61 MHz data can be considered as a virtually punctured code. Having exactly the same bandwidth for both the preamble and data symbols and exactly the same pilot structure for both the preamble and data symbols can maximize spectral efficiency. Other features, such as the cyclical change property and not sending the L1 block in the event that no data segment can be kept unchanged. In other words, the bandwidth of the preamble symbols can be identical to the bandwidth of the data symbols or, as shown in Fig. 57, the bandwidth of the preamble symbols can be the width tuner band (here it is 7.61 MHz). Tuner bandwidth can be defined as a bandwidth that corresponds to a number of total active carriers when using a single channel. That is, the bandwidth of the preamble symbol can correspond to the number of total active carriers (here it is 7.61 MHz).

Fig. 56 shows a virtually punctured code. The 7.61 MHz data between the 8 MHz L1 block can be considered as punctured encoded. When a tuner r603 shown in Fig. 61 uses 7.61 MHz bandwidth for L1 decoding, the r401 frame header extractor in Fig. 63 needs to rearrange the received, changed cyclic L1 block to its original shape as shown in Fig. 56. At this time, L1 decoding is performed using the entire bandwidth of the tuner. Once the L1 block is rearranged, a spectrum from the rearranged L1 block may have a blank region within the spectrum as shown in the upper right side of Fig. 56 because an original size of the L1 block is a width 8 MHz band.

Once the blank region is padded with zeros either after deinterleaving in the symbol domain by the frequency deinterleaver r403 in Fig. 63 or by the symbol deinterleaver r308-1 in Fig. 64 or after the bit domain deinterleaving by symbol decorrelator r306-1, bit multiplexer r305-1 and internal deinterleaver r304-1 in Fig. 64, the block may have a shape that appears to be perforated as shown in the lower right side of Fig. 56.

This L1 block can be decoded in the punched / shortened decoder module r303-1 in Fig. 64. Using this preamble structure, the full bandwidth of the tuner can be used, thus spectral efficiency and gain can be increased. encoding. Furthermore, identical bandwidth and pilot structure can be used for the preamble and data symbols.

Also, if the preamble bandwidth or the preamble symbols bandwidth is set as a tuner bandwidth as shown in Fig. 58 (it is 7.61 MHz in the example), you can get a full L1 block after reorganization even without drilling. In other words, for a frame that has preamble symbols, where the preamble symbols have at least one layer 1 (L1) block, it can be said that the L1 block has 3408 active subcarriers and the 3408 active subcarriers correspond to 7 , 61 MHz of the 8 MHz Radio Frequency (RF) band.

In this way, the spectral efficiency and decoding performance of L1 can be maximized. In other words, in a receiver, decoding can be performed in the punched / shortened decoder module r303-1 in Fig. 64, after only performing deinterleaving in the symbol domain.

Accordingly, the new proposed preamble structure may be advantageous in that it is fully compatible with the previously used preamble, except that the bandwidth is different; the L1 blocks are repeated in a period of 8 MHz; the L1 block can always be decodable regardless of the position of the tuner window; the full bandwidth of the tuner can be used for L1 decoding; maximum spectral efficiency can guarantee more coding gain; the incomplete L1 block can be considered as punctured encoded; a simple and equal pilot structure can be used for both the preamble and the data; and identical bandwidth can be used for both the preamble and the data.

Fig. 59 shows an example of an analog processor. A 601 DAC module can convert digital signal input to analog signal. After the transmit frequency bandwidth is upconverted (602) and analog filtered (603) the signal can be transmitted.

ES 2 608 553 T3

Fig. 60 shows an example of a digital receiver system. The received signal is converted into a digital signal in an r105 analog process module. An r104 demodulator can convert the signal to data in the frequency domain. A frame parser r103 can extract pilots and headers and allow selection of service information that needs to be decoded. A BICM r102 demodulator can correct errors in the transmission channel. An output processor r101 can restore the originally transmitted service flow and timing information.

Fig. 61 shows an example of an analog processor used in which you can select the desired frequency bandwidth from downstream r602 you can restore the baseband. A digital r601 ADC module.

receiver. A tuner / AGC module r603 the received signal. A conversion module can convert the analog signal into signal

Fig. 62 shows an example of a demodulator. A frame detector module r506 can detect the preamble, check for a corresponding digital signal and detect a start of a frame. A time / frequency synchronization module r505 can perform synchronization in the time and frequency domains. At this time, for time domain synchronization, a guard interval correlation can be used. For frequency domain synchronization, correlation can be used or offset can be estimated from phase information of a subcarrier that is transmitted in the frequency domain. A preamble extractor module r504 can extract the preamble from the front of the detected frame. A GI r503 extractor module can extract the guard interval. An r501 FFT module can transform the time domain signal into a frequency domain signal. A channel estimation / equalization module r501 can compensate for errors by estimating the distortion in the transmission channel using a pilot symbol. The r504 preamble extraction module can be omitted if the demodulator is specifically for DVB-C2.

Fig. 63 shows an example of a frame parser. An r404 pilot extractor module can extract a pilot symbol. An r403 frequency deinterleaver module can perform frequency domain deinterleaving. An OFDM symbol merger r402 can restore the data frame from symbol streams transmitted in OFDM symbols. A frame header extractor module r401 can extract physical layer signaling from the header of each transmitted frame and extract the header. The extracted information can be used as parameters for the following processes at the receiver.

Fig. 64 shows an example of a BICM demodulator. Fig. 64a shows a data path and Fig. 64b shows a L1 signaling path. A symbol deinterleaver r308 can perform symbol domain deinterleaving. A ModCod r307 extractor can extract ModCod parameters from the front of each frame in BB and leave the parameters available for subsequent adaptive / variable demodulation and decoding processes. A symbol demapper r306 can decore input symbol streams into Logarithmic Likelihood Ratio (LLR) bit streams. The output bitstream LLRs can be calculated using a constellation used in a transmitter symbol mapper 306 as a reference point. At this point, when using the aforementioned MQAM or NU-MQAM, calculating both the I axis and the Q axis when calculating the bit closest to the MSB and calculating either the I axis or the Q axis when calculating the bits Remaining, an efficient symbol decorelator can be implemented. This method can be applied, for example, to Approximate LLR, Exact LLR or Hard Decision.

When a constellation optimized according to the constellation capacity and the code rate of the error correction code is used in the symbol mapper 306 of the transmitter, the symbol de-mapper r306 of the receiver can obtain a constellation using the code rate and the information constellation capacity transmitted from the transmitter. The bit multiplexer r305 of the receiver can perform an inverse function to that of the bit demultiplexer 305 of the transmitter. The receiver's internal deinterleaver r304 and r302 can perform inverse functions of the transmitter's internal interleaver 304 and external interleaver 302, respectively, to obtain the bit stream in its original sequence. The r302 external deinterleaver can be omitted if the BICM demodulator is specifically for DVB-C2.

The internal decoder r303 and the external decoder r301 of the receiver can respectively perform the corresponding decoding processes for the internal encoder 303 and the external encoder 301 of the transmitter, to correct errors in the transmission channel. Similar processes performed on the data path can be performed on the L1 signaling path, but with different parameters (r308-1 ~ r301-1). At this point, as explained in the preamble part, a shortened / punctured code module r303-1 can be used for L1 signal decoding.

Fig. 65 shows an example of LDPC decoding using shortening / punching. A demultiplexer r301 can separately output an information part and a parity part of the systematic code from input bit streams. For the information part, zero padding (r302a) can be performed according to a number of input bitstreams of the LDPC decoder, for the parity part, input bitstreams can be generated for (r303a) of the

ES 2 608 553 T3 LDPC decoder, spoiling the perforated part. LDPC decoding (r304a) can be performed on the generated bit streams and zeros in the information part can be extracted and output (r305a).

Fig. 66 shows an example of an output processor. A descrambler in BB r209 can restore the scrambled bit streams (209) at the transmitter. A Splitter r208 can restore the BB frames that correspond to multiple PLPs that are multiplexed and transmitted from the transmitter according to the PLP path. For each PLP path, a BB r207-1 header extractor can extract the header that is transmitted at the front of the BB frame. A CRC decoder r206-1 ~ n can perform CRC decoding and leave reliable BB frames available for selection. A null packet inserter r205-1 ~ n can restore null packets that were extracted, for better transmission efficiency, to their original location. A delay recovery module r204-1 ~ n can restore a delay that exists between each PLP path.

Output clock recovery modules r203-1 ~ n can restore the original timing of the service flow from the timing information transmitted from input stream synchronization modules 203-1 ~ n. Some r202-1 ~ n output interface modules can restore the data in the TS / GS packet from input bit streams that are frame segmented in BB. A few r201-1 ~ n output post-processing modules can restore multiple TS / GS streams to a full TS / GS stream, if required. The shaded blocks shown in Fig. 66 represent modules that can be used when a single PLP is processed at a time and the rest of the blocks represent modules that can be used when multiple PLPs are processed at the same time.

The preamble pilot patterns were carefully designed to avoid an increase in PAPR, thus, it is necessary to consider whether the L1 repetition rate can increase the PAPR. The number of information bits in L1 varies dynamically depending on the channel junction, the number of PLPs, etc. In detail, you need to consider things such as that L1's fixed block size can introduce unnecessary overhead; L1 signaling should be more strongly protected than data symbols; and the time interleaving of the L1 block can improve robustness to channel deterioration, such as the need for impulse noise.

For an 8 MHz L1 block repetition rate, as shown in Fig. 67, full spectral efficiency (26.8% increase in BW) is exhibited with virtual perforation, but the PAPR can be increased since the bandwidth of L1 is the same as that of data symbols. For the 8 MHz repetition rate, 4K-FFT DVB-T2 frequency interleaving can be used for the common parts and the same pattern can be repeated in a period of 8 MHz after interleaving.

For a 6 MHz L1 block repetition rate, as shown in Fig. 68, a reduced spectral efficiency can be exhibited without virtual perforation. A similar PAPR problem can occur as for the 8 MHz case, since the bandwidths of L1 and data symbols share LCM = 24 MHz. For the 6 MHz repetition rate, 4K-FFT DVB-T2 frequency interleaving can be used for the common parts and the same pattern can be repeated in a 24 MHz period after interleaving.

Fig. 69 shows a new L1 block repetition rate of 7.61 MHz or full tuner bandwidth. Full spectral efficiency (26.8% BW magnification) can be obtained without virtual drilling. There may not be any PAPR problem since the bandwidths of L1 and data symbols share LCM = 1704 MHz. For the 7.61 MHz repetition rate, 4K-FFT DVB-T2 frequency interleaving can be used for the common parts and the same pattern can be repeated in 1704 MHz period after interleaving.

Fig. 70 is an example of L1 signaling that is transmitted in the frame header. Each information in the L1 signaling can be transmitted to the receiver and can be used as a decoding parameter. Especially, the information can be used in the L1 signal path shown in Fig. 64 and PLPs can be transmitted in each data segment. Increased robustness can be obtained for each PLP.

Fig. 72 is an example of a 308-1 symbol interleaver as shown in the L1 signaling path in Fig. 37 and may also be an example of its corresponding r308-1 symbol deinterleaver as shown in the L1 signaling path in Fig. 64. Blocks with slanted lines represent L1 blocks and solid blocks represent data carriers. L1 blocks can be transmitted not only within a single preamble, but can also be transmitted within multiple OFDM blocks. Depending on the size of an L1 block, the size of the interleaving block can vary. In other words, num_L1_sym and L1_span can be different from each other. To minimize unnecessary overhead, data can be transmitted within the rest of the OFDM symbol carriers where the L1 block is transmitted. At this point, full spectral efficiency can be guaranteed because the L1 block repeat cycle is still a full tuner bandwidth. In Fig. 72, the numbers in the slanted blocks represent the bit order within a single LDPC block.

Accordingly, when bits are written into an interleaving memory in the row direction according to a symbol index as shown in Fig. 72 and read in the column direction according to a carrier index, it is

ES 2 608 553 T3 can obtain a block interleaving effect. In other words, an LDPC block can be interleaved in the time domain and in the frequency domain and then it can be transmitted. Num_L1_sym can be a default value, for example, a number between 2 ~ 4 can be set as a number of OFDM symbols. At this point, to increase the granularity of the L1 block size, a punctured / shortened LDPC code having a minimum codeword length can be used for L1 protection.

Fig. 73 is an example of a L1 block transmission. Fig. 73 illustrates Fig. 72 in the frame domain. As shown in Fig. 73a, the L1 blocks can be spanning the full bandwidth of the tuner or, as shown in Fig. 73b, the L1 blocks can be partially spanned and the rest of the carriers can be used for data carrier. In any case, it can be seen that the repetition rate of the L1 block can be identical to a full tuner bandwidth. Furthermore, for OFDM symbols using the L1 signaling including the preamble, only symbol interleaving can be performed while data transmission is not allowed on those OFDM symbols. Accordingly, for an OFDM symbol used for L1 signaling, a receiver can decode L1 by performing deinterleaving without data decoding. At this point, the L1 block can transmit L1 signaling of the current frame or L1 signaling of a later frame. On the receiver side, L1 parameters decoded from the L1 signaling decoding path shown in Fig. 64 for the decoding process for the data path from the frame parser of the subsequent frame.

In summary, in a transmitter, the interleaving of blocks of the L1 region can be performed by writing the blocks to a memory in the direction of the rows and reading the blocks written from the memory in the direction of the columns. In a receiver, the deinterleaving of blocks from the L1 region can be performed by writing blocks to memory in the direction of the columns and reading the blocks written from memory in the direction of the rows. The read and write addresses of the transmitter and receiver can be interchanged.

When doing a simulation with assumptions that are made such as that CR = 1/2 for protection of L1 and for the common parts of T2; 16-QAM symbol mapping; pilot density of 6 in the Preamble; short LDPC number implies a required amount of perforations / shortenings, results or conclusions can be obtained such as that only one preamble for L1 transmission may not be sufficient; the number of OFDM symbols depends on the amount of the L1 block size; the shortest LDPC codeword (eg, 192-bit information) can be used among the shortened / punctured code for greater flexibility and fine granularity; and a refill can be added if required with negligible overhead. The result is summarized in Fig. 71.

Therefore, for a L1 block repetition rate, full tuner bandwidth without virtual punching may be a good solution and a PAPR problem with full spectral efficiency may still not arise. For L1 signaling, the efficient signaling structure can allow a maximum configuration in an environment of 8 channel junctions, 32 notches, 256 data segments, and 256 PLPs. For L1 block structure, L1 flexible signaling can be implemented according to L1 block size. Time interleaving can be done for better robustness for T2 commons. Less overhead can allow data transmission in a preamble.

Block interleaving of L1 block can be performed for better robustness. Interleaving can be performed with a predefined fixed number of L1 symbols (num_L1_sym) and a number of carriers spanned by L1 as a parameter (L1_span). The same technique is used for P2 preamble interleaving in DVB-T2.

A variable size L1 block can be used. The size can be adaptable to the number of L1 signaling bits, causing reduced overhead. Full spectral efficiency can be obtained without any PAPR problem. A repeat of less than 7.61 MHz may mean that more redundancy can be sent but not used. No PAPR problem may arise due to the 7.61 MHz repetition rate for the L1 block.

Fig. 74 is another example of L1 signaling transmitted within a frame header. This Fig. 74 is different from Fig. 70 in that the L1_span field, which has 12 bits, is divided into two fields. In other words, the L1_span field is divided into a L1_column field that has 9 bits and a L1_row field that has 3 bits. The L1_column field represents the carrier index spanned by L1. Because a data segment begins and ends every 12 carriers, which is the pilot density, the 12 bits of overhead can be reduced by 3 bits to reach 9 bits.

The L1_row field represents the number of OFDM symbols, where L1 is spanning when time interleaving is applied. Therefore, time interleaving can be performed within an area of ​​L1_columns multiplied by L1_rows. Alternatively, the total L1 block size can be transmitted so that L1_span shown in Fig. 70 can be used when no time interleaving is performed. For such a case, the L1 block size is 11,776 x 2 bits in the example, thus 15 bits is sufficient. Therefore, the L1_span field can be made up of 15 bits.

Fig. 75 is an example of frequency or time interleaving / deinterleaving. Fig. 75 shows a part of an entire transmission frame. Fig. 75 also shows the joining of multiple 8Mhz bandwidths. A plot

ES 2 608 553 T3 may consist of a preamble that transmits L1 blocks and a data symbol that transmits data. Different types of data symbols represent data segments for different services. As shown in Fig. 75, the preamble transmits L1 blocks every 7.61 MHz.

For the preamble, a frequency or time interleaving is performed within the L1 blocks and is not performed between the L1 blocks. That is, for the preamble, it can be said that interleaving is performed at the L1 block level. This allows L1 blocks to be decoded by transmitting L1 blocks within a tuner window bandwidth even when the tuner window has been moved to a random location within a channel bonding system.

To decode a data symbol in a tuner window random bandwidth, no interleaving should occur between data segments. That is, for data segments, it can be said that interleaving is done at the data segment level. Therefore, frequency interleaving and time interleaving should be performed within one data segment. Therefore, a symbol interleaver 308 in a data path of a transmitter BICM module, as shown in Fig. 37, can perform symbol interleaving for each data segment. A symbol interleaver 308-1 on an L1 signal path can perform symbol interleaving for each L1 block.

A frequency interleaver 403 shown in FIG. 42 needs to perform interleaving on the preamble and data symbols separately. Specifically, for preamble, frequency interleaving can be performed for each L1 block and for data symbol frequency interleaving can be performed for each data segment. At this point, time interleaving in the data path or the L1 signal path may not be performed considering the low latency mode.

Using the suggested methods and devices, among other advantages, it is possible to implement an efficient digital transmitter, receiver and physical layer signaling structure.

By transmitting ModCod information in each BB frame header that is necessary for ACM / VCM and transmitting the rest of the physical layer signaling in one frame header, signaling overhead can be minimized.

Modified QAM can be implemented for more power efficient transmission or a more robust digital broadcast system against noise. The system may include a transmitter and receiver for each example described and combinations thereof.

Enhanced non-uniform QAM can be implemented for more power efficient transmission or a more robust digital broadcast system against noise. A method of using the code rate of the NU-MQAM and MQAM error correcting code is also described. The system may include a transmitter and receiver for each example described and combinations thereof.

The suggested L1 signaling method can reduce overhead by 3 ~ 4%, minimizing signaling overhead during channel bonding.

It will be apparent to those skilled in the art that various modifications and variations can be made in the present invention without departing from the invention.