Apparatus and method for transmitting and receiving a signal

DESCRIPTION

DEVICE AND PROCESS FOR TRANSMITTING AND RECEIVING A

SIGNAL

Background of the invention

Field of the invention

The present invention relates to a method for transmitting and receiving a signal and a device for transmitting and receiving a signal, and more particularly to a process for transmitting and receiving a signal and a device for the transmission and reception of a signal, which may improve the efficiency of data transmission.

Description of Related Art

With the development of digital broadcasting technology, users are now receiving a high definition (HD) moving image. With the continued development of a compression algorithm and high hardware performance, users will be provided with a better environment in the future. A digital television (DTV) system can receive a digital broadcasting signal and provide a variety of services.

322323331 additional to users, as well as a video signal and an audio signal.

Digital Video Broadcasting (DVB) - C2 is the third specification to join the DVB family of second generation transmission systems. Developed in 1994, today's DVB-C is deployed in over 50 million cable tuners worldwide. In line with other second generation DVB systems, the DVB-C2 uses a combination of Lowdensity Parity Check (LDPC) and BCH codes. This powerful Forward Error Correction FEC provides about 5dB of carrier-to-noise ratio improvement over DVB-C. Appropriate bit interleaving schemes optimize the overall robustness of the FEC system. Extended by a header, these frames are called Physical Layer Pipes (PLP). One or more of these PLPs are multiplexed into a data slice. Two dimensional interleaving (in the time and frequency domains) is applied to each plot allowing the receiver to eliminate the impact of burst deficiencies and selective frequency interference such as single frequency input.

With the development of these digital broadcasting technologies, the demand for a service such as a video signal and an audio signal has increased and the size

322323331 of the data desired by users or the number of transmission channels gradually increased. 0 DVB project Frame structure channel coding and modulation for a second generation digital terrestrial televisions broadcasting system (DVB-T2), Digital Video Broadcasting [online] June 2008 (01.06.2008) describes frame structure channel coding techniques for systems broadcasting

Summary of the Invention

Any occurrence of the term embodiment in the description should be considered as an aspect of the invention, the invention being defined in the appended independent claims. Accordingly, the present invention relates to a process for transmitting and receiving a signal and a device for transmitting and receiving a signal that substantially avoids one or more problems due to the limitations and disadvantages of the related art.

An object of the present invention is to provide a method of transmitting a broadcasting signal to a receiver according to claim

1.

Another aspect of the present invention provides a method of receiving a broadcast signal according to claim 9.

322323331

Yet another aspect provides a broadcast transmitter for a receiver, claim 5.

of the present invention transmit signal according to the

Yet another aspect of the present invention provides a receiver for receiving a broadcast signal, the receiver being described in the appended claim 10.

Brief Description of Drawings

The accompanying drawings, which are included to provide a greater understanding of the invention, forming a part of this application, illustrate embodiments of the invention and together with the description serve to explain the principle of the invention. The figures represent:

Figure 1

Figure 2

Figure 3

Figure 4

Figure 5 is an example of quadrature amplitude modulation (QAM) used in European DVB-T.

Gray Reflected Binary Code (BRGC) process.

Gaussian freckle by modifying the 64-QAM used in DVB-T.

Hamming distance between the pair reflected in

BRGC.

QAM where there is reflected for each I axis and Q axis.

pair

322323331

<td colspan="3">Figure 6 Process for Modifying QAM</td><td rowspan="3">using modified.</td><td rowspan="3">O</td><td rowspan="3">pair</td>

<td rowspan="2">Figure</td><td colspan="2">reflected from the BRGC.</td>

<td>7 example of</td><td>64/256/1024/4096-QAM</td>

<td>Figures</td><td> 8-9</td><td></td><td></td><td></td><td></td>

<td></td><td colspan="3">64-QAM examples modified using</td><td>O</td><td>pair</td>

<td></td><td>reflected</td><td>from BRGC.</td><td></td><td></td><td></td>

<td>Figures</td><td> 10-11</td><td></td><td></td><td></td><td></td>

<td></td><td>example of</td><td>256-QAM modified</td><td>using</td><td>O</td><td>pair</td>

<td></td><td>reflected</td><td>from BRGC.</td><td></td><td></td><td></td>

<td>Figures</td><td> 12-13</td><td></td><td></td><td></td><td></td>

<td></td><td>example of</td><td>1024-QAM modified</td><td>using</td><td>O</td><td>pair</td>

<td></td><td>reflected</td><td>of -B-RGC (0 ~ 511).</td><td></td><td></td><td></td>

<td>Figures</td><td> 14-15</td><td></td><td></td><td></td><td></td>

<td></td><td>example of</td><td>1024-QAM modified</td><td>using</td><td> 0</td><td>pair</td>

<td></td><td>reflected</td><td>BRGC (512-1023).</td><td></td><td></td><td></td>

<td>Figures</td><td> 16-17</td><td></td><td></td><td></td><td></td>

<td></td><td>example of</td><td>4096-QAM modified</td><td>using</td><td>O</td><td>pair</td>

<td></td><td>reflected</td><td>RRGC (0 ~ 511).</td><td></td><td></td><td></td>

<td>Figures</td><td> 18-19</td><td></td><td></td><td></td><td></td>

<td></td><td>example of</td><td>4096-QAM modified</td><td>using</td><td>O</td><td>pair</td>

<td></td><td>reflected</td><td>BRGC (512-1023).</td><td></td><td></td><td></td>

<td>Figures</td><td> 20-21</td><td></td><td></td><td></td><td></td>

<td></td><td>example of</td><td>4096-QAM modified</td><td>using</td><td>O</td><td>pair</td>

<td></td><td>reflected</td><td>of the BRGC (1024-1535).</td><td></td><td></td><td></td>

<td>Figures</td><td> 22-23</td><td></td><td></td><td></td><td></td>

<td></td><td>example of</td><td>4096-QAM modified</td><td>using</td><td>O</td><td>pair</td>

<td></td><td>reflected</td><td>of the BRGC (1530-2047).</td><td></td><td></td><td></td>

<td>Figures</td><td> 24-25</td><td></td><td></td><td></td><td></td>

4096-QAM example modified using pair

ΡΕ2323331 reflected from BRGC (2048-2559).

Figures 26-27 example of 4096-QAM modified using BRGC reflected pair (2560-3071).

Figures 28-29 example of 4096-QAM modified using BRGC reflected pair (30Ί2-3583).

Figures 30-31 are an example of 4096-QAM modified using the BFGC reflected pair (3584-4095).

Fig. 32 is an example of modified QAM bit mapping where 256-QAM is modified using BRGC.

Fig. 33 is an example of transforming MQAM into a nonuniform constellation.

Fig. 34 is an example of digital transmission system.

Fig. 35 is an example of an input processor.

36 is information that may be included in the baseband (BB).

Fig. 37 is an example of BICM.

Fig. 38 is an example of shortened / punctured encoder. Fig. 39 is an example of applying various constellations. Fig. 40 is another example of cases where compatibility between conventional systems is considered. Fig. 41 is a frame structure comprising the preamble for the L1 signaling and data symbol for PLP data.

42 is an example of frame builder.

Fig. 43 is an example of pilot insert 404 shown in Fig. 4.

322323331

<td>Figure Figure</td><td> 44 45</td><td colspan="2">SP structure. new SP structure or pilot pattern</td><td>(PP)</td><td colspan="2"> 5' .</td>

<td>Figure</td><td> 46</td><td>PP5 structure '</td><td>suggested.</td><td></td><td></td><td></td>

<td>Figure</td><td> 47</td><td>relationship</td><td>between symbol</td><td>in</td><td>Dice</td><td>and</td>

preamble.

Fig. 48 is another relationship between data symbol and preamble.

<td>Figure</td><td> 49</td><td>example</td><td>in</td><td>delays profile</td><td>of channel by</td><td>cable.</td>

<td>Figure</td><td> 50</td><td colspan="2">structure Z = 112.</td><td>scattered pilot</td><td>that uses</td><td>z = 5 6 and</td>

<td>Figure</td><td> 51</td><td>example</td><td>in</td><td>based modulator</td><td>in OFDM.</td><td></td>

<td>Figure</td><td> 52</td><td>example</td><td>in</td><td colspan="2">preamble structure.</td><td></td>

<td>Figure</td><td> 53</td><td>example</td><td>in</td><td>decoding of</td><td>preamble.</td><td></td>

<td>Figure</td><td> 54</td><td colspan="2">process</td><td>to conceive</td><td>preambles</td><td>more</td>

optimized.

Fig. 55 Another example of preamble structure

Fig. 56 is another example of preamble decoding. Fig. 57 is an example of preamble structure.

Fig. 58 is an example of decoding Ll.

Fig. 59 is an example of analog processor.

Fig. 60 is an example of digital reception system.

Fig. 61 is an example of analog processor used in the receiver.

Fig. 62 is an example of demodulator.

Fig. 63 is an example of frame parser.

Fig. 64 is an example of a BICM demodulator.

Fig. 65 is an example of LDPC decoding using shortening / puncturing.

Fig. 66 is an example of an output processor.

322323331

Figure 67 Example of L1 Block Repeat Rate of 8

MHz.

Figure 68 Example of L1 Block Repeat Rate of 8

MHz.

Figure 69 new 7.61 MHz L1 block repetition rate.

Fig. 70 is an example of L1 signaling that is transmitted in the frame header.

Figure 71 preamble result and structure simulation

Ll.

Fig. 72 is an example of symbol interlacing device.

Fig. 73 is an example of a L1 block transmission.

Fig. 74 is another example of L1 signaling transmitted within a frame header.

Fig. 75 is an example of frequency or time interlacing / deinterlacing.

Description of Preferred Embodiments

Detailed reference will now be made to preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings. Wherever possible, the same reference numbers will be used in all drawings to designate the same or similar parts.

In the following description, the term service is broadcast content that may be indicative of any transmitted / received by the transmission device.

Transmissão2323331 signal transmission / reception.

Quadrature Amplitude Modulation (QAM) using Reflected Gray Binary Code (BRGC) is used as modulation in a broadcast transmission environment where bit interleaved coding (BICM) is used. Figure 1 shows an example of 64-QAM used in European DVB-T.

BRGC can be performed using the process shown in the figure. 2. An n-bit BRGC can be accomplished by adding an inverse BRGC (n-1) bit code (ie reflected code) to a (n-1) bit rear end by adding Os to a BRGC (n-1) front. -1) original bit, and adding ls to a reflected code front. The BRGC code performed by this process has a Hamming distance between the contiguous codes of one (1). In addition, when BRGC is applied to QAM, the Hamming distance between one point and the four nearest points is one (1) and the Hamming distance between the point and four nearest second points. from the point, it's two (2). Such characteristics of Hamming distances between a specific constellation point and other contiguous points can be called the QAM Gray mapping rule.

To make a system robust against Additive White Gaussian Noise AWGN, the distribution of signals transmitted from a

ΡΕ2323331 transmitter can be made near Gaussian distribution. To be able to do this, the locations of the points in the constellation can be modified. Figure 3 shows an output close to Gaussian modifying the 64-QAM used in DVB-T. Such a constellation can be termed as nonuniform QAM (NU-QAM).

To make a QAM constellation non-uniform, a Cumulative Distribution Function (CDF) can be used.

In the case of 64, 256, or 1024 QAM, ie 2^ΛΝ AMs, QAM can be divided into two independent N-PAMs. By dividing the Gaussian CDF into N sections of identical probability and allowing one signal point in each section representing the section, a constellation showing the Gaussian distribution can be realized. In other words, the xj coordinate of the newly defined nonuniform NPAM can be defined as follows:

<img file="PT2323331T_D0001.tif" />

: λ · - ii

LV'2, V^v '' 2Λ⁷ j (Eq.

Fig. 3 is an example of 64QAM transformation of DVB-T to NU-64QAM using the above processes. Figure 3 represents the result of modifying the coordinates of each I axis and Q axis using the above processes and corresponding to the constellation points prior to the

ΡΕ2323331 newly defined coordinates. In the case of 32, 128, or 512 QAM, ie transverse QAM, which is not 2^ΛΝ QAM, by modifying Pj appropriately, can find a new coordinate.

One embodiment of the present invention may modify QAM using BRGC employing the characteristics of BRGC. As shown in Figure 4, the Hamming distance between the reflected pair in BRGC is one because it differs only in one bit that is added in front of each code. Figure 5 shows the characteristics in QAM, where there is the reflected pair for each I axis and Q axis. In this figure, there is the reflected pair on each side of the dotted black line.

By using existing reflected pairs in QAM, an average power of a QAM constellation can be reduced while maintaining the Gray mapping rule in QAM. In other words, in a constellation where an average power is normalized to 1, the minimum Euclidean distance can be increased in the constellation. When this modified QAM is applied to transmission or communication systems, it is possible to implement a more robust system using noise that uses the same energy as a conventional system or a system with the same performance as a conventional system, but which consumes less energy.

Figure 6 shows a process for modifying QAM using a reflected BRGC pair. Figure 6a shows

322323331 is a constellation and figure 6b shows a flowchart for modifying QAM using a reflected pair of BRGC. First, a target point that has the highest power must be found between the constellation points. Candidate points are the points where that target point can move and are the contiguous points closest to the reflected pair of the target point. Then an empty point (ie a point not yet taken up by other points) having the lowest power must be found between the candidate points, comparing the power of the target point and the power of a candidate point. If the power of the candidate point is lower, the target point moves the candidate point. These processes are repeated until an average power of points in the constellation reaches a minimum while maintaining the Gray mapping rule.

Figure 7 shows an example of modified 64/256/1024 / 4096QAM. The corresponding gray values ​​correspond to figures 8 ~ 31 respectively. In addition to these examples, other types of modified QAM may be performed which allow identical power optimization to be performed. This is because a destination point can move to multiple candidate points. The suggested modified QAM can be applied to not only 64/256/1024/4096-QAM, but also transverse QAM, a larger QAM, or modulations using another BRGC other than QAM.

Figure 32 shows an example of mapping of

322323331 modified QAM bits where 256-QAM is modified using BRGC. Figure 32a and Figure 32b show the mapping of the most significant bits (MSB). The points designated as full circles represent the mappings of ones and the points designated as blank circles represent the mappings of zeros. Similarly, each bit is matched as shown in figures (a) to (h) in figure 32 until the least significant bits (Least Significant Bits - LSB) are matched. As shown in Fig. 32, modified QAM can enable bit decision using only the I and Q axes as conventional QAM, except for a bit that is close to MSB (Fig. 32c and Fig. 32d). Using these features, a simple receiver can be realized by partially modifying a receiver for QAM. An efficient receiver can be implemented by checking both I and Q values ​​only in determining the near bit of MSB and calculating only I or Q for the rest of the bits. This process can be applied to approximate LLR, exact LLR, or difficult decision.

Using modified QAM or MQAM, which uses the above BRGC characteristics, non-uniform constellation or NU-MQAM can be performed. In the above equation where Gaussian CDF is used, Pj can be modified to fit MQAM. Like QAM, in MQAM, two PAMs that have the I axis and Q axis can be considered. However, unlike QAM where several points that correspond to a value of each PAM axis are identical, the

ΡΕ2323331 number of points changes in MQAM. If a number of points corresponding to the j value of PAM is defined as nj in an MQAM where there are a total of M constellation points, then Pj can be defined as follows:

<img file="PT2323331T_D0002.tif" />

f = / - l

V

no

11, + (Eg. 2)

By using the newly defined Pj, MQAM can be transformed into nonuniform constellation. Pj can be defined as follows for the 256-MQAM example.

~ J 2.5 10 22 36 51 ^{Pi &amp;} [256 ’256’256’ 256’ 256

6? 84 [02.19.5 136.5 Â54 172 189 205 220 234 246 253.5]

256'256 * 256 '256' 256 * 256 * 256'256'256'256 * 256 * 256 '256 j

Fig. 33 is an example of transforming MQAM into a nonuniform constellation. NU-MQAM performed using these processes can retain MQAM receptor characteristics with modified coordinates of each PAM. In this way, an efficient receiver can be implemented. Additionally, a more robust noise system than the previous NU-QAM can be implemented. For a more efficient broadcast transmission system, MQAM and NU-MQAM hybridization is possible. In other words, a more noise-resistant system can be implemented using MQAM for an environment where a high code rate error correction code is used and otherwise employing NU-MQAM. In such a case, a transmitter may

It is possible for a receiver to have code rate information of a currently used error correction code and a type of modulation currently used such that the receiver can demodulate according to the currently used modulation.

Fig. 34 shows an example of digital transmission system. Entries can include a number of MPEG-TS streams or GSE (General Stream Encapsulation) streams. An input processor module 101 may add transmission parameters to the input stream and execute schedules for a BICM module 102. The BICM module 102 may add redundancy and interleave data for transmission channel error correction. A frame builder 103 can construct frames by adding physical layer signaling information and pilots. A modulator 104 can perform modulation on input symbols in efficient processes. An analog processor 105 may perform various processes for converting digital input signals to analog output signals.

Fig. 35 shows an example of an input processor. The MPEG-TS or GSE input stream can be transformed by the input preprocessor into a total of n streams that will be processed independently. Each of these streams can be a complete TS frame that includes components of various services or a minimal TS frame that includes service component (ie video or audio). In addition, each of these streams can be a GSE stream that

322323331 broadcasts multiple services or a single service.

input interface module 202-1 may assign multiple input bits equal to the maximum data field capacity of a baseband frame (BB). A padding zone can be inserted to complete the capacity of the LDPC / BCH code block. Input stream timing module 203-1 may provide a mechanism for regenerating the transport stream clock (or generic packet stream) at the receiver to ensure constant end-to-end bit rates and delays.

In order to allow transport stream recombination without requiring additional memory at the receiver, incoming transport streams are delayed by delay compensators 204-1 considering the interleaving parameters of the data PLPs in a group and the corresponding common PLP. . Zero-size packet deletion modules 205-1-n can increase transmission efficiency by removing zero-size packets inserted in case of VBR (variable bit rate) service. Cyclic Redundancy Check (CRC) encoder modules 206-1-n can add CRC parity to increase the reliability of BB frame transmission. BB header insert modules 207-1-n may add the frame header to an initial part of the BB frame. The information that can be included in the BB header is shown in figure 36.

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A fusion / cut module 208 may perform the BB frame cut of each PLP by merging BB frames of various PLPs, and scheduling each BB frame within a transmission frame. Therefore, the fusion / cut module 208 can output signaling information L1 regarding the assignment of PLP in the frame. Finally, a BB 209 mixer module can randomize input bitstreams to minimize correlation between bits within bitstreams. The shaded modules in figure 35 are modules used when the transmission system uses a single PLP, the other modules in figure 35 modules used when the transmission device uses multiple PLPs.

Figure 37 shows an example of the BICM module. Figure 37a shows the data path and Figure 37b shows the path L1 of the BICM module. An external encoder module 301 and an internal encoder module 303 may add redundancy to error correction input bitstreams. An external interleaver module 302 and an internal interleaver module 304 can interleave bits to prevent burst error. External interleaver module 302 can be omitted if BICM is specifically for DVB-C2. A bit demultiplexer module 305 can control the reliability of each bit issued by the internal interleaver module 304. A symbol mapping module 306 can match input bitstreams to symbol streams. At this point, it is possible to use any of a conventional QAM, a MQAM that uses the BRGC described above for performance improvement, a NU-QAM that

322323331 uses non-uniform modulation, or a NU-MQAM that uses applied non-uniform modulation BRGC for performance improvement. To build a system that is more robust against noise, combinations of modulations using MQAM and / or NU-MQAM can be considered depending on the error correction code code rate and constellation capability. At this time, the symbol mapping module 306 may use a suitable constellation according to the code rate and constellation capacity. Fig. 39 shows an example of such combinations.

Case 1 shows an example of using NU-MQAM exclusively at low code rate for simplified system deployment. Case 2 shows an example of using optimized constellation at each code rate. The transmitter can send error correction code code rate and constellation capability information to the receiver such that the receiver can use an appropriate constellation. Fig. 40 shows another example of cases where compatibility between conventional systems is considered. In addition to the examples, it is possible to

<td>combinations</td><td>additional</td><td>to optimize</td><td>the system.</td><td></td>

<td> 0</td><td>module of</td><td>insert 307</td><td>of the header</td><td>ModCod</td>

<td>introduced</td><td>in the figure.</td><td>37 can get</td><td>information from</td><td>return</td>

coding and adaptive modulation (ACM) / variable coding and modulation

322323331 (Variable coding and modulation - VCM) and add parameter information used in coding and modulation for an FEC block as a header. The Modulation Type / Code Rate (ModCod) header may include the following information:

* FEC type (1 bit) - long or short LDPC * Code rate (3 bits) * Modulation (3 bits) - up to 64K QAM * PLP identifier (8 bits) interleaver module 308 can perform interlacing in the domain of symbol for additional interlacing effects. Similar processes performed on the data path can be performed on signaling path L1 but with possibly different parameters (301-1 - 308-1). At this point, a shortened / punctured code module (303-1) for internal code may be used.

Fig. 38 shows an example of LDPC encoding using shortening / puncturing. The shortening process can be performed on input blocks that have fewer bits than a required number of LDPC encoding bits since so many zero bits required for LDPC encoding can be placed in padding zones (301c). Zero fill zone input bitstreams can display parity bits through LDPC encoding (302c). At this moment,

With 322323331 for bit streams corresponding to original bit streams, zeros can be removed (303c) and for parity bit streams, punching (304C) can be performed according to code rates. These processed information bit streams and parity bit streams can be multiplexed into original sequences and output (305c).

Fig. 41 shows a frame structure comprising preamble for L1 signaling and data symbol for PLP data. It can be seen that the preamble and data symbols are cyclically generated using a frame as a unit. Data symbols include PLP type 0 which is transmitted using a fixed coding modulation and PLP type 1 which is transmitted using a variable modulation / coding. For PLP type 0, information such as modulation, FEC type, and FEC code rate are transmitted in the preamble (see Fig. 42 frame header insert 401). For PLP type 1, the corresponding information may be transmitted in an FEC block header of a data symbol (see Fig. 37 ModCod header insertion 307). By splitting PLP types, ModCod's complementary information can be reduced by 3 ~ 4% of a total baud rate, for PLP type 0 which is transmitted at a fixed bit rate. At a receiver, for PLP type 0 fixed modulation / PLP encoding, frame header remover r401 shown in Fig. 63 can extract modulation and rate information from the FEC code and provide the extracted information to a module.

BICM decoding ΡΕ2323331. For PLP type 1 PLP variable modulation / encoding, the ModCod, r307 and r307-1 extraction modules shown in Fig. 64 can extract and provide the required parameters for BICM decoding.

Fig. 42 shows an example of a frame builder. A frame header insertion module 401 may form a frame of input symbol streams and may add the frame header in front of each transmitted frame. The plot header can include the following information:

* Number of connected channels (4 bits) * Guard interval (2 bits) * PAPR (2 bits) * Pilot pattern (2 bits) * Digital identification system (16 bits) * Frame identification (16 bits) * Length of frame (16 bits) - number of symbols per frame of Orthogonal Frequency Division Multiplexing (OFDM) * Superframe length (16 bits) - number of frames per superframe * Number of PLPs (8 bits) * To each PLP

PLP identification (8 bits)

Channel bonding id (4 bits)

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PLP start (9 bits)

PLP type (2 bits) - common PLP or others

PLP payload type (5 bits)

MC type (1 bit) - fixed / variable modulation &amp; coding if MC type == fixed modulation &amp; coding

FEC type (1 bits) - long or short LDPC

Coderate (3 bits)

Modulation (3 bits) - up-to 64K QAM end if;

Number of notch channels (2 bits) for each notch,

Notch start (9 bits)

Notch width (9 bits) end for;

PLP width (9 bits) - max number of FEC blocks of PLP

PLP time interleaving type (2 bits) end for;

* CRC-32 (32 bit) channel bonding environment is assumed for information transmitted in the frame header L1, the data corresponding to each data slice being defined as PLP. Therefore, information such as the PLP identifier, channel link identifier, and PLP start address is required for each channel used in the link. One embodiment of the present invention suggests passing the ModCod field in the FEC frame header if the PLP type supports variable modulation / encoding and the ModCod propulsion field in the frame header if the PLP type supports

322323331 fixed modulation / coding to reduce supplementary signaling information. Additionally, if there is a notch band for each PLP, by transmitting the notch start address and its width, the corresponding decoding carriers in the receiver may become unnecessary.

Fig. 43 shows an example of pilot pattern 5 (PP5) applied in a channel bonding environment. As shown, if the SP positions coincide with the pilot preamble positions, an irregular pilot structure may arise.

Fig. 43a shows an example of pilot insert module 404 as shown in Fig. 42. As shown in Fig. 43, if a single frequency band (e.g. 8 MHz) is used, the available bandwidth is 7 61 MHz, but if multiple frequency bands are connected, the guard bands can be removed, so the frequency efficiency can be considerably increased. Fig. 43b is an example of the preamble insert module 504 as shown in Fig. 51 which is transmitted at the front of the frame and even with channel bonding, the preamble has a repetition rate of 7.61 MHz, which is width Ll. This is a structure considering the bandwidth of a tuner that performs the initial channel scan.

322323331

There are pilot patterns for both the preamble and the data symbols. For data symbol, scattered pilot (SP) patterns can be used. Pilot Patterns 5 (PP5) and Pilot Patterns 7 (PP7) of T2 may be good candidates for frequency-only interpolation. PP5 has x = 12, y = 4, z = 48 for GI = 1/64 and PP7 has x = 24, y = 4, z = 96 for GP = 1/128. Additional interpolation time is also possible for better channel estimation. Preamble pilot patterns can cover all possible pilot positions for initial channel acquisition. Additionally, the preamble pilot positions should be coincident with the SP positions, and a unique pilot pattern is desired for both the preamble and the SP. Preamble pilots can also be used for time interpolation and all preambles can have an identical pilot pattern. These requirements are important for scanning C2 detection and necessary for frequency offset estimation with coding sequence correlation. In a channel bonding environment, coincidence in pilot positions should also be maintained for channel bonding because the uneven pilot structure may impair interpolation performance.

In detail, if a distance z between diffused pilots (SPs) in an OFDM symbol is 48 and if a distance y between SPs that correspond to a specific SP carrier along the time axis is 4, an effective distance x after time interpolation becomes

322323331

12 This is when the guard interval fraction (GI) is 1/64. If the GI fraction is 1/128, x = 24, y = 4, and z = 96 can be used. If channel bonding is used, SP positions may be realized coincident with pilot preamble positions by generating non-continuous points in the diffused pilot structure.

At this point, the preamble pilot positions can be coincident with all SP positions of the data symbol. When channel bonding is used, the data slice where a service is transmitted can be determined independently of the 8 MHz bandwidth granularity. However, to reduce the complementary information for addressing the data slice, it can be chosen the transmission from the SP position and ending in the SP position.

When a receiver receives such SPs, if necessary, the channel estimation module r501 shown in Fig. 62 can perform time interpolation to obtain the pilots shown in dotted lines in Fig. 43 and perform frequency interpolation. At that time, for non-continuous points whose ranges are marked 32 in Figure 43, you can implement left and right interpolation separately or perform one-sided interpolation, then perform interpolation on the other side using already interpolated pilot positions, of which the range is 12 as the starting point

Reference. At this time, the data slice width can vary within 7.61 MHz, so you can minimize power consumption by performing channel estimation and decoding only the required subcarriers.

<td></td><td>THE</td><td>figure</td><td>44 shows</td><td>another</td><td>example of</td><td>PP 5</td>

<td colspan="2">applied in</td><td colspan="2">bonding environment</td><td>of channel</td><td colspan="2">or a structure</td>

<td>from SP</td><td>for</td><td>keep</td><td>the distance</td><td>effective</td><td colspan="2">x like 12 to</td>

<td>avoid</td><td colspan="2">the structure</td><td>SP irregular</td><td colspan="2">shown in the figure</td><td> 43</td>

<td>When</td><td colspan="2">it is used</td><td>the binding of</td><td>channel. THE</td><td>figure 44a is</td><td>an</td>

SP structure for the data symbol and Fig. 44b is an SP structure for the preamble symbol.

As shown, if the SP distance is kept consistent in case of channel bonding, there will be no problem in frequency interpolation but the pilot positions between the data symbols and preamble may not match. In other words, this structure does not require additional channel estimation for an irregular SP structure, however, the SP positions used in channel bonding and pilot preamble positions become different for each channel.

The figure to provide mentioned in

Specifically, solving those shows a new SP or PP5 'structure a solution to the two problems above channel bonding environment, a pilot distance of x = 16 may problems. To preserve pilot density or to maintain the same complementary information, a

322323331

ΡΡ5 'may have x = 16, y = 3, z = 48 for GJ = 1/64 and an ΡΡΊ' may have x = 16, y = 6, z = 96 for GJ = 1/128. Frequency-only interpolation capability can still be maintained. Pilot positions are shown in Fig. 45 for comparison with structure PP5.

Figure 46 shows an example of a new SP standard or PP5 'structure in channel bonding environment. As shown in Fig. 46, if a single channel or channel bonding is used, an effective pilot distance x = 16 may be provided. Additionally, because SP positions can be realized coincident with pilot preamble positions, deterioration of channel estimation caused by SP irregularity or non-coincident SP positions can be prevented. In other words, there is no irregular SP position for the frequency interpolator and coincidence between preamble and SP positions is provided.

Accordingly, the proposed new SP standards may be advantageous in that a single SP standard may be used for both single channel and bonded channel; no irregular pilot structure can be caused, thus a good channel estimate is possible; Both preamble positions and SP pilot positions can be kept coincident, pilot density can be kept the same as for PP5 and ΡΡΊ respectively, and interpolation capability can be preserved only.

322323331 of the frequency.

Additionally, the preamble structure may meet the requirements such that pilot preamble positions should cover all possible SP positions for initial channel acquisition; the maximum number of carriers should be 3409 (7.61 MHz) for the initial scan; Exactly the same pilot patterns and coding sequence should be used for C2 detection, and no specific detection preamble such as PI and T2 is required.

In terms of the frame structure, the granularity of the data slice position can be changed to 16 carriers instead of 12, so that less complementary position addressing information can appear, and no other problem is expected in the frame. which refers to the data slice condition, zero slot condition can be expected, etc.

Therefore, in channel estimation module r501 of Fig. 62, pilots in all preambles can be used when SP time interpolation is performed.

<td>of the symbols of</td><td>Dice.</td><td>Per</td><td>this the</td><td colspan="2">acquisition</td><td>of channel and</td>

<td>Estimate of</td><td>channel</td><td>we</td><td>Limits</td><td>gives</td><td>plot</td><td>can be</td>

<td>improved.</td><td></td><td></td><td></td><td></td><td></td><td></td>

<td>Now,</td><td>at the</td><td>what</td><td colspan="2">refers to</td><td>to</td><td>requirements</td>

related to the preamble and the pilot structure, there is

322323331 consensus that the positions of the preamble pilots and SPs should coincide regardless of channel bonding; the total number of carriers in block ll must be divisible by the pilot distance to avoid uneven structure at the end of the band; L1 blocks must be repeated in the frequency domain; and the L1 blocks must always be decodable at the position of the arbitrary tuning window. The additional requirements would be that pilot positions and standards must be repeated for the period of 8 MHz; The correct carrier frequency offset should be estimated without knowledge of channel bonding, and L1 (reordering) decoding is impossible before the frequency offset is compensated.

Fig. 47 shows a relationship between the data symbol and preamble when the preamble structures are used as shown in Fig. 52 and Fig. 53. The L1 block can be repeated for periods of 6 MHz. both frequency deviation and preamble shift pattern. L1 decoding is not possible at the arbitrary tuner position without channel bonding information, and a receiver cannot differentiate between preamble shift value and frequency offset.

Thus, a receiver, specifically for frame header remover r401 shown in Fig. 63 for performing L1 signal decoding, the channel link structure must be obtained. Given that it is

Since the expected amount of preamble shift in two vertically shaded regions is known in Fig. 47, the time / frequency synchronization module r505 in Fig. 62 can estimate carrier frequency offset. Based on the estimate, the signaling path L1 (r308-1-1301-1) in Fig. 64 can decode L1.

Fig. 48 shows a relationship between the data symbol and the preamble when the preamble structure as shown in Fig. 55 is used. The L1 block can be repeated for periods of 8 MHz. For L1 decoding, only the frequency offset must be found, and knowledge of channel bonding may not be required. Frequency drift can be easily estimated using a Pseudo Random Binary Sequence PRBS. As shown in Fig. 48, the preamble and data symbols are aligned, so searching for additional synchronization may become unnecessary. Therefore, for a receiver, specifically for the frame header remover module r401 shown in Fig. 63, it is possible that only the pilot coding sequence correlation peak must be obtained to perform the decoding of the L1 signal. The time / frequency synchronization module r505 in Fig. 62 can estimate the carrier frequency offset from the peak position.

Figure 49 shows an example of profile of

322323331 cable channel delays.

From a pilot design standpoint, the current GI already protects the spread of cable channel propagation times. In the worst case, redesigning the channel model may be an option. To repeat the pattern exactly every 8 MHz, the pilot distance must be a divider of 3584 carriers (z = 32 or 56). A pilot density of z = 32 may increase the pilot complementary information, so z = 56 may be chosen. Slightly shorter propagation time coverage may not be important in the cable channel. For example, it could be 8με for PP5 'and 4με for ΡΡΊ' compared with 9.3με (PP5) and 4.7με (PP7). Significant delays can be covered by both pilot patterns even in the worst case. For the preamble pilot position, no more than all positions are required.

SP on the data symbol.

If the -40 dB delay path can be ignored, the current spread of the propagation times may become 2.5μ3, 1/64 GI = ΙμΞ, or 1/128 GI = 3.5με.

This shows that the pilot distance parameter, z = 56 may be a good enough value. Additionally, z = 56 may be a convenient value for structuring the pilot pattern allowing the preamble structure shown in the figure.

Figure 50 shows a dispersed pilot structure using z = 56 and Z = 112 which is constructed in the module of

322323331 pilot insert 404 in Fig. 42. PP5 '(x = 14, y = 4, z = 56) and ΡΡΊ' (x = 28, y = 4, z = 112) are proposed. End carriers may be inserted for end closure.

As shown in figure 50, the pilots are aligned at 8 MHz from each end of the band, each pilot position and pilot structure being repeated every 8 MHz. Thus, this structure can support the preamble structure shown in figure 48. Additionally, a common pilot structure may be used between the preamble and data symbols. Therefore, channel estimation module r501 in Fig. 62 can perform channel estimation using preamble interpolation and data symbols because no irregular pilot pattern can arise, regardless of the window position that is decided by the slice locations. Dice. At this time, using only frequency interpolation may be sufficient to compensate for the channel distortion of the propagation time dispersion. If time interpolation is additionally performed, a more accurate channel estimation can be performed.

Therefore, in the proposed new pilot pattern, the pilot position and pattern can be repeated based on a period of 8 MHz. A single pilot pattern can be used for both preamble and data symbols. L1 decoding can always be possible without knowledge of channel bonding. Additionally, the proposed pilot pattern may not commonly affect T2 because

322323331 the same pilot strategy of widespread pilot pattern can be used; T2 already uses 8 different pilot patterns, and no significant receiver complexity can be increased by modified pilot patterns. For a pilot coding sequence, the PRBS period may be 2047 (m-sequence); PRBS generation can be reset every 8 MHz, where the period is 3584; the pilot repetition rate of 56 may also be close to 2047; no PAPR problem can be expected.

Fig. 51 shows an example of an OFDM based modulator. Input symbol streams can be time domain transformed by the IFFT 501 module. If necessary, the peak-to-average power ratio (PAPR) can be reduced in the PAPR reducing module 502. For PAPR processes, active constellation extension (ACE) or tone reserve can be used. The GI inserter module 503 may copy a last part of the effective OFDM symbol to fill the guard interval in a cyclic prefix form.

Preamble inserter module 504 can insert preamble in front of each transmitted frame so that a receiver can detect digital signal, frame and acquire time / frequency offset acquisition. At this time, the preamble signal can perform physical layer signaling such as FFT dimension (3 bits) and guard interval dimension (3 bits). The preamble insert module 504 may be omitted if the modulator is

322323331 specifically for DVB-C2.

Fig. 52 shows an example of a preamble channel bonding structure generated in the preamble insert module 504 in Fig. 51. A complete L1 block must always be decodable at any arbitrary 7.61 MHz tuner window position and not any loss of Ll signaling must take place regardless of the position of the tuner window. As shown, L1 blocks can be repeated in the frequency domain for periods of 6 MHz. The data symbol can be linked per channel for each 8 MHz channel. If for L1 decoding a receiver uses a tuner such as tuner r603 shown in Fig. 61 which uses a bandwidth of 7.61 MHz, frame header remover r401 in Fig. 63 needs to rearrange the cyclically displaced received block L1 (Fig. 53) to its original shape. This rearrangement is possible because the L1 block is repeated for each 6 MHz block. Fig. 53a can be rearranged into Fig. 53b.

Fig. 54 shows a process for designing a more optimized preamble. The preamble structure of Fig. 52 uses only 6 MHz of the 7.61 MHz full bandwidth of the L1 decoder tuner. In terms of spectrum efficiency, the 7.61 MHz bandwidth is not fully utilized. Therefore, there may be further optimization in spectral efficiency.

322323331

Fig. 55 shows another example of the preamble structure or symbol preamble structure for full spectrum efficiency generated in header insertion module 401 in frame 42. Like the data symbol, L1 blocks can be frequency domain for periods of 8 MHz. A complete L1 block is still always decodable in any arbitrary position of the 7.61 MHz tuning window. After tuning, data at 7.61 MHz can be viewed as virtually punctured code. Having exactly the same bandwidth for both preamble and data symbols and exactly the same pilot structure for both preamble and data symbols can maximize spectrum efficiency. Other features such as cyclically shifted property and not sending block L1 in case of no data slices can be kept unchanged. In other words, the preamble symbol bandwidth may be identical to the data symbol bandwidth or, as shown in Fig. 57, the preamble symbol bandwidth may be the tuner bandwidth (here is 7.61 MHz). Tuner bandwidth can be defined as a bandwidth that corresponds to a number of total active carriers when a single channel is used. That is, the preamble symbol bandwidth can correspond to the number of total active carriers (here is 7.61 MHz).

Fig. 56 shows a virtually punctured code. The 7.61 MHz data between the 8 MHz Ll block

322323331 can be considered as punctured code. When a tuner r603 shown in Fig. 61 uses a bandwidth of 7.61 MHz for L1 decoding, the frame header remover r401 in Fig. 63 needs to rearrange the received L1 block, cyclically shifted to its original form as shown. At this time, L1 decoding is performed using the full tuner bandwidth. Once the L1 block is rearranged, a spectrum of the rearranged L1 block may have a blank region within the spectrum as shown in the upper right of the

<td>figure 56 because 8 MHz wide</td><td>one size Band</td><td>original</td><td>from block Ll</td><td>features</td>

<td>Like this</td><td colspan="2">that the region in</td><td>white if</td><td>meet</td>

<td>filled with</td><td>zeros, or</td><td>after the</td><td colspan="2">deinterlacing in</td>

symbol domain by the frequency deinterlacing r403 in Fig. 63 or by the symbol deinterlacing r308-l in Fig. 64 or after bit domain deinterlacing by the r306-l symbol demapper, r305-l bit multiplexer, and internal de-interleaver r304- 1 in Fig. 64, the block may have a shape that appears to be punctured as shown in the lower right side of Fig. 56.

This L1 block can be decoded in the punctured / shortened decoding module r303-1 in Fig. 64. By using this preamble structure, the entire tuner bandwidth can be used, thereby increasing spectrum efficiency and gain. gives

322323331 coding. Additionally, an identical bandwidth and common pilot structure may be used for preamble and data symbols.

Additionally, if the preamble bandwidth or the preamble symbol bandwidth is defined as a tuner bandwidth as shown in Fig. 58, (it is 7.61 MHz in the example), a block may be obtained. Ll complete after reorganization even without punching. In other words, for a frame that has preamble symbols, where preamble symbols have at least one layer 1 (Layer-1) block (Ll), it can be said that block L1 has 3408 active subcarriers, corresponding to the 3408 subcarriers active at 7.61 MHz 8 MHz radio frequency band (RF).

In this way, spectrum efficiency and decoding performance Ll can be maximized. In other words, at the receiver, decoding may be performed on the punctured / shortened decoder module r303-1 in Fig. 64, after performing only deinterlacing in the symbol domain.

Accordingly, the proposed new preamble structure may be advantageous in that it is fully compatible with the previously used preamble except that the bandwidth is different; L1 blocks are repeated for periods of 8 MHz; block ll can always be

ΡΕ2323331 decodable regardless of tuner window position; can be used for decoding Ll the full tuner bandwidth; Maximum spectrum efficiency can guarantee more code gain; Incomplete block L1 may be considered as punctured coded; a simple pilot structure can be used and the same for both preamble and data; and identical bandwidth can be used for both preamble and data.

Fig. 59 shows an example of an analog processor. A DAC 601 module can convert a digital signal input to analog signal. After the transmission frequency bandwidth has been converted upwards (602) and analogously filtered (603) the signal can be transmitted.

Fig. 60 shows an example of a digital receiving system. The received signal is converted to digital signal in an analog process module r05. A demodulator r104 can convert the signal to frequency domain data. An rl03 frame analyzer can remove pilots and headers and enable selection of service information that needs to be decoded. A BICM r100 demodulator can correct errors in the transmission channel. A freckle processor 110 may restore the originally transmitted service stream and timing information.

322323331

Fig. 61 shows an example of analog processor used in the receiver. An AGC r603 tuner / module can select the desired frequency bandwidth from the received signal. An r602 downconversion module can restore the baseband. An ADC r601 module can convert an analog signal into a digital signal.

Fig. 62 shows an example of demodulator. A frame detection module r506 can detect the preamble, check for a corresponding digital signal, and detect the start of a frame. A time / frequency synchronization module r505 can perform time and frequency synchronization. At this time, for time domain synchronization, a guard interval correlation can be used. For frequency domain synchronization, correlation may be used or the offset may be estimated from the phase information of a subcarrier that is transmitted in the frequency domain. A preamble remover module r504 can remove the front preamble of the detected frame. An GI removal module, r503, can remove the guard interval. An FFT module, r501, can transform the time domain signal into the frequency domain signal. A channel estimation / equalization module r501 can compensate for errors by estimating transmission channel distortion using the pilot symbol. Preamble removal module r504 can be omitted if the demodulator is specifically for DVBC2.

322323331

Fig. 63 shows an example of frame analyzer. An r404 pilot remover module can remove pilot symbol. A frequency deinterlacing module r403 can perform frequency domain deinterlacing. An OFDM symbol concentrator r402 can restore the data stream of symbol streams transmitted in OFDM symbols. A frame header remover module r401 can extract physical layer signaling from the header of each transmitted frame and remove the header. The extracted information can be used as parameters for the following processes on the receiver.

Fig. 64 shows an example of a BICM demodulator. Fig. 64a shows a data path and Fig. 64b shows a signaling path L1. A symbol deinterleaver r308 can perform deinterlacing in the symbol domain. A ModCod r307 extract can extract ModCod parameters from the front of each BB frame and make the parameters available for the following adaptive / variable demodulation and decoding processes. An r306 symbol demapper can demap input symbol streams into Log-Likelyhood Ratio (LLR) bitstreams. LLR streams of output bits can be calculated using a constellation used in a transmitter symbol mapper 306 as a reference point. At this point, when the aforementioned MQAM or NU-MQAM is used, when calculating both the I axis and Q axis when calculating the nearest bit of MSB and when calculating the I axis or Q axis when

If the residual bits are calculated, an efficient symbol demapper can be implemented. This process can be applied to, for example, approximate LLR, exact LLR, or difficult decision.

When an optimized constellation is used according to the constellation capacity and error correction code code rate in the transmitter's symbol mapper 306, the receiver's symbol demapper r306 can obtain a constellation using the code rate and information about the transmitter's transmitted constellation capacity. The receiver bit multiplexer r305 may perform an inverse function of the transmitter bit demultiplexer 305. The receiver's internal deinterlacing r304 and the receiver's external deinterlacing r302 can perform inverse functions of the transmitter's internal interleaver 304 and external interleaver 302, respectively, to obtain the bit stream in their original sequence. External deinterleaver r302 can be omitted if the BICM demodulator is specifically for DVB-C2.

The receiver's internal decoder r303 and the receiver's external decoder r301 may perform the corresponding decoding processes for the transmitter's internal encoder 303 and external code 301, respectively, to correct errors in the transmission channel. Similar processes performed on the data path can be performed on the signaling path L1, but with different parameters (r308-1 - r301-1). At this point, as

As explained in the preamble part, a shortened / punctured code module r303-1 may be used for decoding the L1 signal.

Fig. 65 shows an example of LDPC decoding using shortening / punching. A demultiplexer r301a can separately output part information and systematic code parity part from input bitstreams. For the information part, a zero fill zone (r302a) can be realized according to a series of LDPC decoder input bitstreams, for the parity part, input bitstreams for (r303a) the decoder LDPC can be generated by puncturing the punctured part. LDPC decoding (r304a) can be performed on generated bit streams, and the zeros in the information portion can be removed and issued (r305a).

Fig. 66 shows an example of output processor. A BB decoder r209 can restore encoded bit streams 209 in the transmitter. A splitter r208 can restore BB frames corresponding to multiple PLP's that are multiplexed and transmitted from the transmitter according to the PLP path. For each PLP path, a BB header remover r207 ~ l ~ n can remove the header that is transmitted in front of the BB frame. A CRC r206-l ~ n decoder can perform CRC decoding and make reliable BB frames available for selection. An r205-l ~ n zero packet insertion module can restore packets

322323331 zero that have been removed for greater transmission efficiency in its original location. A delay recovery module r204-1 can restore a propagation time that exists between each PLP path.

An output clock recovery module r203-1 can restore the original service flow timing from the timing information transmitted from the input flow timing module 203-1. An output interface module r202-1 can restore TS / GS packet data from input bitstreams that are divided into portions in the BB frame. An output postprocessing module r201-l ~ n can restore multiple TS / GS streams by converting them to a complete TS / GS stream if required. The shaded blocks shown in Fig. 66 represent modules that can be used when a single PLP is processed in a period and the rest of the blocks represent modules that can be used when multiple PLPs are simultaneously processed.

Pilot preamble patterns have been carefully designed to avoid PAPR increase, so if the repetition rate L1 increases the PAPR must be considered. The number of information bits L1 varies dynamically according to channel bonding, number of PLPs, etc. In detail, it is necessary to consider things such as the fixed size of block L1 can introduce unnecessary supplementary information; Ll signs must be protected more strongly than

322323331 data symbols; and L1 block time interlacing can improve robustness over channel damage such as the need for impulsive noise.

<td></td><td>For a fee</td><td>block repeat</td><td>8 MHz ll,</td>

<td>such</td><td>as shown in</td><td>Figure 67 is displayed.</td><td>the efficiency</td>

<td>of</td><td>full spectrum</td><td>(26.8% increase</td><td>BW) with</td>

virtual punching but the PAPR can be extended since the bandwidth L1 is the same as that of the data symbols. For the 8 MHz repetition rate, 4K-FFT FVB-T2 frequency interleaving can be used for standardization, and the same pattern can be repeated at an 8 MHz period after interlacing.

For a 6 MHz L1 block repetition rate as shown in Fig. 68, reduced spectrum efficiency can be displayed without virtual punching. A similar PAPR problem may occur as in the case of 8 MHz as the bandwidths of L1 and data symbols share LCM = 24 MHz. For the 6 MHz repetition rate, 4K-FFT FVB-T2 frequency interleaving can be used for smoothing, and the same pattern can be repeated at 24 MHz after interlacing.

<td></td><td></td><td>THE</td><td>figure</td><td> 69</td><td>show</td><td>a new rate of</td><td>repetition</td><td>in</td>

<td>block</td><td>Ll</td><td>in</td><td> 7.61</td><td>MHz</td><td>or one</td><td>bandwidth</td><td>complete</td><td>in</td>

<td colspan="3">tuner</td><td>. An</td><td colspan="2">efficiency</td><td>broad spectrum</td><td>(increase</td><td>in</td>

<td> 26.8%</td><td>in</td><td>BW)</td><td>can</td><td>to be</td><td>obtained</td><td>without punching</td><td>virtual.</td><td>No</td>

322323331 there may be any problem with PAPR since Ll and data symbol bandwidths share LCM ~ 1704 MHz. For the 7.61 MHz repetition rate, 4K-FFT DVB-T2 frequency interleaving can be used for standardization. The same pattern may be repeated at a time of about 1704 MHz after interlacing.

Fig. 70 is an example of L1 signaling that is transmitted in the frame header. Each information in the L1 signaling can be transmitted to the receiver and can be used as a decoding parameter. In particular, the information may be used in the signal path L1 shown in Fig. 64, and PLPs may be transmitted in each data slice. An increase in robustness can be obtained for each PLP.

Fig. 72 is an example of a symbol interleaver 308-1 as shown in the signaling path L1 in Fig. 37 and may also be an example of its corresponding symbol deinterleaver r308-1 as shown in the signaling path L1 in Fig. 64. Blocks with slanted lines represent L1 blocks and solid blocks represent data carriers. L1 blocks can be transmitted not only within a single preamble, but can also be transmitted within several OFDM blocks. Depending on the size of the L1 block, the size of the interlacing block may vary. In other words, num_Ll_sym and amplitude L1 may differ from each other. To minimize supplemental information

Unnecessary dados2323331, data may be transmitted in the rest of the OFDM symbol bearers where the L1 block is transmitted. At this point, full spectrum efficiency can be guaranteed because the L1 block repeat cycle is still a full bandwidth tuner. In Fig. 72, block numbers with slanted lines represent the order of bits within a single LDPC block.

Accordingly, when the bits are written in a row-direction interleaving memory according to a symbol index as shown in Fig. 72 and read in the column direction according to a carrier index, an interleaving effect can be obtained. block. In other words, an LDPC block can be interlaced in the time domain and the frequency domain and then transmitted. Num_L1_sym may be a predetermined value, for example a number between 2-4 may be defined as a series of OFDM symbols. At this point, to increase the granularity of the L1 block dimension, a punctured / shortened LDPC code that has a minimum length of the keyword can be used to protect the L1.

Fig. 73 is an example of a L1 block transmission. Fig. 73 illustrates Fig. 72 in the frame domain. As shown in Fig. 73a, L1 blocks can be generated at full tuner bandwidth or as shown in Fig. 73b, L1 blocks can be partially generated and the rest of

ΡΕ2323331 carriers can be used for data transport. In either case, it can be seen that the L1 block repetition rate can be identical to a full tuner bandwidth. Additionally, for OFDM symbols using L1 signaling including the preamble, only symbol interleaving can be performed until data transmission on those OFDM symbols is allowed. Therefore, for the OFDM symbol used for L1 signaling, a receiver can decode L1 by performing deinterlacing without data decoding. At this point, the block L1 may transmit the current frame signaling Ll or a subsequent frame signaling Ll. At the receiver side, the decoded parameters L1 of the signaling decoding path L1 shown in Fig. 64 can be used for the decoding process for the data frame analyzer data path of the subsequent frame.

Briefly, in a transmitter, the region 11 interleaving blocks can be executed by writing blocks in a memory in an inline direction and reading the written blocks of memory in the column direction. At a receiver, the deinterlacing blocks of region L1 can be executed by writing blocks in a memory in a column direction and reading the written blocks from memory in the row direction. The read and write indications of the transmitter and receiver can be interchanged.

322323331

When the simulation is performed with assumptions such as CR = 1/2 for protection L1 and for T2 equalization; 16-QAM symbol matching; pilot density of 6 in the preamble; the short LDPC number implies that the required amount of punching / shortening is performed, results or conclusions such as preamble only for L1 transmission may not be sufficient; the number of OFDM symbols depends on the amount of block dimension L1; the shortest LDPC keyword (e.g. 192 bits of information) among the shortened / punctured code can be used for flexibility and fine granularity; The filling area may be added, if necessary, with negligible additional information. The result is summarized in figure 71.

Consequently, for a L1 block repetition rate, the full tuner bandwidth without virtual punching can be a good solution and no PAPR problem with full spectrum efficiency arises yet. For L1 signaling, an efficient signaling structure can allow maximum configuration in an environment of 8 channel links, 32 notches, 256 data slices, and 256 PLPs. For L1 block structure, flexible L1 signaling can be implemented according to L1 block size. Time interlacing can be performed for better robustness for T2 uniformity. Less supplementary information may allow data transmission in the preamble.

322323331

For better robustness block interlacing of block Ll can be performed. Interlacing can be performed with a predefined fixed number of symbols L1 (num_L1_sym) and a number of carriers generated by L1 as a parameter (Ll_span). The same technique is used for the preamble P2 interleaving in BVB-T2.

A variable block L1 may be used. The size may be adaptable to the amount of signaling bits L1, resulting in reduced complementary information. Full spectrum efficiency can be achieved without any PAPR problems. A repetition of less than 7.61 MHz may mean that more redundancy may be sent but not used. No PAPR problem can arise due to the 7.61 MHz repetition rate for the Ll block.

Fig. 74 is another example of L1 signaling transmitted within a frame header. This figure 74 is different from figure 70 in which the 12-bit Ll_span field is divided into two fields. In other words, the Ll_span field is divided into a 9-bit Ll_column and a 3-bit Ll_row. The Ll_column represents the carrier index that Ll generates. Since the data slice starts and ends every 12 carriers, which is the pilot density, the 12 bits of supplementary information can be reduced by 3 bits to 9 bits.

322323331

Ll_row represents the number of OFDM symbols in which Ll is fractionated when time interlacing is applied. Consequently, time interlacing can be performed within an area of ​​Ll_columns multiplied by Ll_rows. Alternatively, a total block size L1 can be transmitted such that L1 span shown in Fig. 70 can be used when time interlacing is not performed. For this case, the block size Ll is 11,776 x 2 bits in the example, so 15 bits is enough. Consequently, the Ll_span field can consist of 15 bits.

Fig. 75 is an example of frequency or time interlacing / de-interlacing. Fig. 75 shows a part of a complete transmission frame. Fig. 75 also shows the connection of various bandwidths of 8 MHz. A frame can be composed of a preamble that transmits L1 blocks and a data symbol that transmits data. The different types of data symbols represent data slices for different services. As shown in Fig. 75, the preamble transmits L1 blocks for each 7.61 MHz.

For the preamble, frequency or time interlacing is performed within L1 blocks and not between L1 blocks. That is, for the preamble, it can be said that interlacing is performed at block level Ll. This allows decoding of L1 blocks by transmitting L1 blocks within a tuner window bandwidth.

322323331 even when the tuner window has moved to a random location within a channel bonding system.

For decoding data symbols in a random tuner window bandwidth, interleaving between data slices should not take place. That is, for data slices, it can be said that interlacing is performed at the data slice level. Consequently, frequency interlacing and time interlacing must be performed within a data slice. Therefore, a symbol interleaver 308 on a data path of a transmitter BICM module as shown in Fig. 37 can perform symbol interleaving for each data slice. A symbol interleaver 308-1 on a signal path L1 can perform symbol interleaving for each block.

Ll.

A frequency interleaver 403 shown in Fig. 42 needs to perform interleaving on the preamble and data symbols separately. Specifically, for preamble, frequency interleaving can be performed for each L1 block and for the data symbol, frequency interleaving can be performed for each data slice. At this point, time interlacing in the data path or signal path L1 may not be performed considering the low latency mode.

322323331

Using the suggested processes and devices, among other advantages, it is possible to implement an efficient digital transmitter, receiver and signaling of the physical layer structure.

By transmitting ModCod information on each BB frame header that is required for ACM / VCM and transmitting the rest of the physical layer signaling on a frame header, signaling supplemental information can be minimized.

Modified QAM can be implemented for more energy efficient transmission or a more robust digital noise broadcasting system. The system may include the transmitter and receiver for each example described and combinations thereof.

An improved non-uniform QAM can be implemented for more energy efficient transmission or a more robust digital noise broadcasting system. Also described is a process of using an error correction code code rate of NUMQAM and MQAM. The system may include the transmitter and receiver for each example described and combinations thereof.

The suggested L1 signaling process can reduce the complementary information by 3 ~ 4% by reducing the complementary information during channel bonding.

322323331

It will be apparent to those skilled in the art that various modifications and variations may be made to the present invention without departing from the scope of the invention.

Lisbon, December 20, 2016