Noise-balanced qam detection

Field of the invention

0001The present invention relates to principles of phase modulation coding and decoding, which may be utilised in various types of telecommunication systems. The invention is especially applicable for radio communication systems.

Background of the invention

0002A commonly used principle for transmitting data over a radio channel and for overcoming the signal rate limitation of binary sequence signalling is to make use of four or more unique symbols. Thereby, the bit rate can exceed the maximum signal rate (in bits/s) corresponding to double the pass-band (in Hz) as given by the Nyquist theorem.

0003Quadrature phase shift keying (QPSK) also denoted 4-state quadrature amplitude modulation (4-QAM) involves that two-bit words are coded into four discrete symbols. These symbols can be represented as signal vectors in the complex plane having constant amplitude but four distinct phase values in relation to a reference signal. Detection is carried out by establishing to which quadrant in the complex plane the received signal can be referred.

0004If a higher modulation order is used, the bit rate can be increased further. However, higher requirements are inflicted on the detection stage since it becomes more difficult to distinguish the individual symbols from one another, as they appear closer in the complex plane. The deterioration of the signal as transmitted over a given media also constitutes a limitation to the possible number of symbols being used.

0005Higher order keying is commonly referred to as M'ary QAM, where M = 2^N refers to the number of discrete symbols being available, whereby N bits can be transmitted per symbol. M'ary QAM is also referred to as M'ary APK (amplitude phase shift keying), as both the amplitude and phase may vary for individual symbols.

0006<figref idref="f0001">Figure 1</figref> shows a conventional transmitter and <figref idref="f0001">figure 2</figref> shows a conventional receiver.

0007The transmitter unit comprises a data buffer 1, a mapper 2, baseband filtering unit 3, intermediate frequency (IF) oscillator 6, phase divider 5, adders 7, and summer 4 from which a radio frequency (RF) signal is transmitted.

0008Data stored temporarily in buffer 1 is conveyed to the mapper 2 in accordance with the rate data can be transmitted over the radio interface. The data, which can be seen as a binary bit serial string, is partitioned into symbols by the mapper 2 having an I component and Q component in the complex plane as explained above.

0009The receiver, on the other hand, decodes I and Q components multiplying the incoming signal (RF) with 90 degree phase skewed signals provided by signal oscillator IF12 from divider D11. The signal of IF 12 is typically rendered coherent by means of a carrier recovery PLL (phase locked loop) with the carrier signal from IF 6, such that the RF signal, after being filtered in respective filters 9 and 10, can be decoded back into the complex plane. An error signal 16 corresponding to the deviation of the detected symbol value from an expected symbol value is fed into PLL loop back filter 13 adjusting IF generator IF 12.

0010<figref idref="f0002">Fig. 3 and 4</figref> show a conventional scheme for transmitting data. A frame alignment word F1 consisting of a predetermined sequence of symbols functions as a reference for subsequent frames of traffic data B1, B2..BN-1. For example, the frame-word may have a length of 8 bits. After transmission of a fixed period of frames, the frame alignment word is repeated. Via a frame-aligner 15, in which the predetermined sequence is recovered, the demodulator, can identify the individual frame position for each frame.

0011As is shown in <figref idref="f0002">fig. 4'</figref>, the frame alignment word may comprise a single pilot signal P, which is discernible from the remaining traffic carrying symbols T.

0012An error signal vector E corresponding to the deviation of the detected symbol value D from an expected reference symbol value R is detected in de-mapper 14 and is fed into PLL loop back filter 13, which deviates a control value E', also denoted deviated error signal. For instance the angle ϕ between vectors for points D and R, can be calculated and used as error control signal E'.

0013The latter signal is used to adjust IF generator IF 12, so that the phase of signal from IF12 is rendered coherent with the signal of IF6.

0014Additive noise, which consists mainly of thermal noise in the receiving signal, will typically be transferred to the phase detector output. The noise part of the received signal constitutes a constant area around the transmitted constellation symbol, as the noise part is independent of the given symbol.

0015All QAM schemes larger than 4 have constellations for which the envelopes vary for the individual symbols. Hence, if the error signal E is used directly and unprocessed for QAM schemes larger than 4, the noise transfer from symbols with a small envelope, G, will be much larger than symbols with a large envelope, H. This relation has been indicated in <figref idref="f0003">fig. 5</figref>, which discloses detected symbols for a 16QAM constellation under the influence of thermal noise.

0016Therefore, a need has arisen as to compensate for noise contributions.

0017The optimum with respect to noise transfer would be to "equalise" the phase detector with regard to the envelope, hence to multiply the detected phase error with the envelope of the signal, as expressed below: <maths id="math0001" num="I:"><math display="block"><mi mathvariant="italic">Eʹ</mi><mo>=</mo><mi>ϕ</mi><mo>⋅</mo><mfenced open="|" close="|"><mi>D</mi></mfenced></math><img file="EP1535439B1_D0001.tif" /></maths> where D is the detected signal and ϕ is the angle between the detected signal and the decided symbol reference R (square centre). Please confer <figref idref="f0004">fig. 6</figref>.

0018However, the above calculation requires many programming instructions and is therefore not suitable for some applications.

0019A deviated signal, which is more easily calculated, is given by the expression: <maths id="math0002" num="II:"><math display="block"><mi mathvariant="italic">Eʹ</mi><mo>=</mo><mi mathvariant="italic">D_Q</mi><mo mathvariant="italic">⋅</mo><mi mathvariant="italic">R_I</mi><mo mathvariant="italic">-</mo><mi mathvariant="italic">D_I</mi><mo mathvariant="italic">⋅</mo><mi mathvariant="italic">R_Q</mi></math><img file="EP1535439B1_D0002.tif" /></maths> where D is the detected signal and R is the decided symbol.

0020In the latter case, the noise will get a "square" dependency related to the envelope of the signal. In <figref idref="f0004">fig. 7</figref>, various lines have been shown for given values of control signals E' for relation II above.

0021In order to equalise the detected error with regard to noise, the error should be divided by the actual envelope. Hence, the following expression may be used: <maths id="math0003" num="III:"><math display="block"><mi mathvariant="italic">Eʹ</mi><mo mathvariant="italic">=</mo><mfrac><mrow><mi mathvariant="italic">D_Q</mi><mo mathvariant="italic">⋅</mo><mi mathvariant="italic">R_I</mi><mo mathvariant="italic">-</mo><mi mathvariant="italic">D_I</mi><mo mathvariant="italic">⋅</mo><mi mathvariant="italic">R_Q</mi></mrow><msqrt><msup><mfenced><mi mathvariant="italic">D_Q</mi></mfenced><mn>2</mn></msup><mo mathvariant="italic">+</mo><msup><mfenced><mi mathvariant="italic">D_I</mi></mfenced><mn>2</mn></msup></msqrt></mfrac></math><img file="EP1535439B1_D0003.tif" /></maths>

0022However, also the two above methods of noise balancing require relatively complex algorithms, which then again require extensive processing power in the receiver stage.

0023Prior art document <patcit id="pcit0001" dnum="US5796786A"><text>US-5796786</text></patcit> shows a phase error detection method in which a phase error value for the received data is obtained by subtracting the decided I-channel data and multiplying the sign of the difference by the difference itself and applying a weighting function to the phase error value. This signal is used for phase correction of received data. The weighting function is applied to reduce the wrong detection of a phase error caused by a decision error possibly generated in an adjacent error between symbols.

0024Prior art document <patcit id="pcit0002" dnum="US5684842A"><text>US-5684842</text></patcit> shows a digital receiver for a 16-QAM constellation. The receiver estimates a symbol identical to a transmitted symbol based upon a symbol in the constellation. For correcting the phase error, a phase error detector is used which produces a correction signal which corrects the local oscillator. According to <patcit id="pcit0003" dnum="US5684842A"><text>US-5684842</text></patcit> it is reckoned that the probability of a good decision is enhanced by weighting the correction signal that acts on the local oscillator. One zone is defined for each state which zone contains the state in which the decision made is considered a good decision. Outside the zones, the decisions are considered less good, all the more so as the received symbol shown in the plane by the point P (decision-subjected point) is farther away from the zones containing each state. A weighting factor is defined which is equal to 1 when the decision-subjected point is situated inside the zones which contain the states. The weighting factor γ progressively decreases when the decision-subjected point is farther away from said zones. The sampled received symbol can be written in cartesian coordinates as: p_k =a_k +jb_k. When the decision-subjected point is situated outside the zones which contain the states, the weighting factor is such that: γ =z/(λ*d), where z is equal to the smaller of the absolute values a_k or b_k, for a QPSK modulation and where lambda λ is a zone defining factor. The weighting factor may be stored in a look-up table. The two-part form of claim 1 is based on <patcit id="pcit0004" dnum="US5684842A"><text>US-5684842</text></patcit>.

0025Prior art document <patcit id="pcit0005" dnum="US4683578A"><text>US4683578</text></patcit> deals with automatic gain control (AGC) circuits which monitor the amplitude levels of the received signals and generate an output to the demodulator section of a receiver. In such circuits, the AGC gain is dynamically adjusted by using the RMS value of the incoming signals to try to maintain said output within a pre-selected amplitude range. <patcit id="pcit0006" dnum="US4683578A"><text>US4683578</text></patcit> tracks gain hits by adjusting the modem signal gain by a slight amount every time whenever either the innermost or outermost point in each quadrant of the signal constellation is received. More particularly, when the innermost point is received the signal gain is increased by a slight amount and decreased by a slight amount when an outermost point is decoded.

Summary of the invention

0026It is a first object of the invention to set forth a method, which provides for a robust reception of high bit rate data transmission, which is bandwidth efficient, and which method lessens hardware requirements.

0027This object has been achieved by the subject matter specified in claim 1.

0028It is a further object to set forth a method that provides noise balance with respect to the envelope of the QAM signal.

0029This object has moreover been achieved by the subject matter defined by claim 1.

0030It is a further object to set forth a method, which provides suppression of noise related to QAM decision thresholds.

0031This object has been achieved by claim 4.

0032More advantages will appear from the following detailed description of preferred embodiments of the invention.

Brief description of the drawings

0033<ul id="ul0001" list-style="none"><li><figref idref="f0001">Fig. 1</figref> shows a known transmitter,</li><li><figref idref="f0001">fig. 2</figref> shows a known receiver,</li><li><figref idref="f0002">fig. 3</figref> shows a typical known frame timing diagram,</li><li><figref idref="f0002">fig. 4</figref> shows a detail of <figref idref="f0002">fig. 3</figref>,</li><li><figref idref="f0002">fig. 4'</figref> shows another known frame diagram,</li><li><figref idref="f0003">fig. 5</figref> shows thermal noise in a 16QAM-keying constellation,</li><li><figref idref="f0004">fig. 6</figref> shows details relating to an error vector E,</li><li><figref idref="f0004">fig. 7</figref> shows curves for various derived control error values E' according to a known method in a 64-QAM system,</li><li><figref idref="f0005">fig. 8</figref> shows an exemplary quadrant error correction scheme according to a first embodiment according to the invention,</li><li><figref idref="f0006">fig. 9</figref> shows an exemplary octant error correction scheme according to an alternative first embodiment according to the invention,</li><li><figref idref="f0007">fig. 10</figref> shows an exemplary receiver for parabolic error weighting according to a second embodiment of the invention.</li><li><figref idref="f0008">fig. 11</figref> shows an exemplary first parabolic error weighting function according to the second embodiment of the invention,</li><li><figref idref="f0009">fig. 12</figref> shows an exemplary second parabolic error weighting function according to the second embodiment of the invention, and</li><li><figref idref="f0010">fig. 13</figref> shows an exemplary 128-QAm scheme according to a third embodiment of the invention.</li></ul>

Detailed description of a preferred embodiment of the invention

0034According to a first preferred embodiment of the invention, noise equalisation is achieved by directly using the quadrature error components, E_Q and E_I, which are independent of the signal envelope.

0035According to a first embodiment, as shown in <figref idref="f0005">fig. 8</figref>, the constellation area is divided into 4 sectors for which the deviated error control signal E' is determined as follows in dependency of the detected symbol vector D: <ul id="ul0002" list-style="none" compact="compact"><li>IV: for sector A -<maths id="math0004" num=""><math display="block"><mfenced open="|" close="|"><mi mathvariant="italic">D_Q</mi></mfenced><mo>≥</mo><mo>|</mo><mi mathvariant="italic">D_I</mi><mo mathvariant="italic">|</mo><mo mathvariant="italic">:</mo><mspace width="4em" /><mi mathvariant="italic">Eʹ</mi><mo mathvariant="italic">≡</mo><mi mathvariant="italic">E_Q</mi></math><img file="EP1535439B1_D0004.tif" /></maths></li><li>V: for sector B -<maths id="math0005" num=""><math display="block"><mfenced open="|" close="|"><mi mathvariant="italic">D_Q</mi></mfenced><mo>&lt;</mo><mo>|</mo><mi mathvariant="italic">D_I</mi><mo mathvariant="italic">|</mo><mo mathvariant="italic">:</mo><mspace width="4em" /><mi mathvariant="italic">Eʹ</mi><mo mathvariant="italic">≡</mo><mi mathvariant="italic">E_Q</mi></math><img file="EP1535439B1_D0005.tif" /></maths></li></ul>

0036According to a further aspect of the first embodiment, the constellation area is divided by four lines through the origin into 3 sectors, A, B and C, as shown in <figref idref="f0006">fig. 9</figref>. The definition of these sectors and the deviated error control signal E' in the associated sectors are given as follows: <ul id="ul0003" list-style="none" compact="compact"><li>VI: for sector A -<maths id="math0006" num=""><math display="block"><mfenced open="|" close="|"><mi mathvariant="italic">D_Q</mi></mfenced><mo>≥</mo><mn>2</mn><mo>⋅</mo><mo>|</mo><mi mathvariant="italic">D_I</mi><mo mathvariant="italic">|</mo><mo mathvariant="italic">:</mo><mspace width="4em" /><mi mathvariant="italic">Eʹ</mi><mo mathvariant="italic">≡</mo><mi mathvariant="italic">E_I</mi></math><img file="EP1535439B1_D0006.tif" /></maths></li><li>VII: for sector B -<maths id="math0007" num=""><math display="block"><mfenced open="|" close="|"><mi mathvariant="italic">D_Q</mi></mfenced><mo>&lt;</mo><mi>½</mi><mo>⋅</mo><mo>|</mo><mi mathvariant="italic">D_I</mi><mo mathvariant="italic">|</mo><mo mathvariant="italic">:</mo><mspace width="4em" /><mi>Eʹ</mi><mo mathvariant="italic">≡</mo><mi mathvariant="italic">E_Q</mi></math><img file="EP1535439B1_D0007.tif" /></maths></li><li>VIII: for sector C (non A, non B): <maths id="math0008" num=""><math display="block"><mi mathvariant="italic">Eʹ</mi><mo>≡</mo><mfrac><mrow><mi mathvariant="italic">E_I</mi><mo mathvariant="italic">+</mo><mi mathvariant="italic">E_Q</mi></mrow><mn>2</mn></mfrac></math><img file="EP1535439B1_D0008.tif" /></maths></li></ul>

0037As appears, for sector A and B surrounding the I and Q axis, one of the orthogonal error components E_I or E_Q is used directly and unprocessed once it is detected to which sector the detected error symbol, D, belongs. For sector C in the "corner areas", the mean value of the orthogonal error components E_I or E_Q is used as deviated error control signal E'.

0038The deviation of the error control value E' is for instance carried out in the PLL loop back filter 13 of the receiver shown in <figref idref="f0001">fig. 2</figref>.

0039It appears that the above steps of comparison and calculation can be easily accomplished by very few programming instructions.

0040Another noise contributor is related to the thresholds of the QAM decision device, i.e. the symbol detection performed in de-mapper 14. When the detected symbol approaches the square shaped boundaries of size T, for instance shown in <figref idref="f0003">fig. 5</figref> and <figref idref="f0004">6</figref>, the noise contribution may lead to errors. In those cases where the noise component is larger than the distance to the symbol boundary, the error signal will get a wrong sign that will disturb the PLL tracking heavily.

0041According to the invention, this effect can be reduced by applying the above weighting function that suppresses the detector output for points close to the boundaries.

0042The weighting can be accomplished in a number of ways. However, according to the invention it is a basic objective to reduce the sharp transitions in the detector output when the detected signal D approaches the decision boundaries. Known demodulators have a typical "saw-tooth" like shape of the detector vs. phase error response. It is moreover an objective to subdue responses outside the detector boundaries.

0043According to a preferred second embodiment of the invention the following relation is used for weighting the occurring symbol error: <maths id="math0009" num="VIII:"><math display="block"><mi mathvariant="italic">WE</mi><mo>=</mo><mi mathvariant="italic">Eʹ</mi><mrow><mo>(</mo><mn>1</mn><mo>-</mo><mfrac><mrow><mn>2</mn><mo>⁢</mo><mi>W</mi></mrow><mi>T</mi></mfrac><mo>)</mo><mi>for</mi><mo>-</mo><mi>½</mi><mo>⁢</mo><mi>T</mi><mo>≤</mo></mrow><mi mathvariant="italic">Eʹ</mi><mo>≤</mo><mi>½</mi><mo>⁢</mo><mi>T</mi></math><img file="EP1535439B1_D0009.tif" /></maths> where T corresponds to the boundary size as indicated in <figref idref="f0009">fig. 12</figref> and E' corresponds to deviated control error and W = Max {abs(E_I); abs(E_Q)}.

0044In principle, no values of E' will occur outside the above range of +- ½T once it has been detected to which symbol boundary E_Q and E_I belongs. Hence one alternative is to render the weighting function zero outside the above interval. <maths id="math0010" num="IX:"><math display="block"><mi mathvariant="italic">WE</mi><mo>=</mo><mn>0</mn><mspace width="1em" /><mi mathvariant="italic">for</mi><mspace width="1em" /><mi mathvariant="italic">Eʹ</mi><mo>&lt;</mo><mo>-</mo><mi>½</mi><mo>⁢</mo><mi>T</mi><mo>∨</mo><mi>½</mi><mo>⁢</mo><mi>T</mi><mo>&lt;</mo><mi mathvariant="italic">Eʹ</mi></math><img file="EP1535439B1_D0010.tif" /></maths>

0045Generally, it applies, that the weighted error signal (WE) should approach zero for error signals (E) approaching zero. The weighting function should also produce a positive value for positive values in the area close to zero and a negative value for negative values in the area close to zero. Moreover, the weighting function approaches zero when the error signal vector approaches the symbol boundaries of the detected symbol.

0046In <figref idref="f0007">fig. 10</figref>, one embodiment for implementing the above method has been shown, wherein a weighting filter 17 performs the weighting of the derived error control signal E' and provides the feedback weighted control value WE for IF generator IF12. As appears from <figref idref="f0007">fig. 10</figref> an optional signal 18 from de-mapper 14 may enable or bar the weighting of some specific symbols.

0047The above weighting will result in a parabolic shape of the detector vs. phase error response. The error weighting has been illustrated by line 26 in <figref idref="f0008">fig. 11</figref>.

0048As appears from <figref idref="f0008">fig. 11</figref>, the false phase errors, which otherwise would have occurred, if no weighting function were utilised are substantially reduced. Consequently, the possibilities for erroneously adjusting the PLL loop have been reduced.

0049In <figref idref="f0009">fig. 12</figref>, an alternative weighting function using discrete values has been shown. As appears from the figure the curve shape of the weighted function is similar to the curve-shape of <figref idref="f0008">fig. 11</figref>. However, a number of discrete values are used for simplicity.

0050As mentioned above it is an objective to subdue responses near the decision boundaries. However, according to a further embodiment of the invention, this only applies for the central symbols in the I/Q plane.

0051In <figref idref="f0010">fig. 13</figref>, an exemplary excerpt of a 128-QAM scheme has been shown. Eight outer signalling points 32 in the constellation define corners on a borderline to an outer area 34. If signals are detected in the outer corner area 34, the deviated error control signal is not weighted, WE=E'. On the other hand, if signals fall outside the symbol boundaries along the Q and I axes as indicated by the area 36, the weighting function WE=0 is applied. This leads to an improved burst error performance.

0052Hence, according to the invention an improved demodulator has been accomplished which lessens errors from occurring and enhances tracking capabilities although noise should occur in the demodulator.