A system for dc restoration of serially transmitted binary signals

21i~7~Q BP File No. 0374-155 ~: A SYSTEM FOR DC RESTORATION OF SERIALLY TRANS~l l~ BINARY SIGNALS 5 FIELD OF THE INVENTION This invention relates to direct current (DC) restoration of serially transmitted binary signals.

More particularly, the invention relates to a hysteretic bipolar clamp DC restorer circuit particularly suited for use with automatic co-axial cable equalizers.

BACKGROUND OF THE INVENTION In signal transmission applications where the average or DC component of a serially transmitted binary signal is non-zero, the DC component of the signal must be restored at the receiver in order 15 to achieve accurate data reproduction.

Where the transmission medium is a coaxial cable which includes AC coupling methods for signal transmission, there are two steps required to recover the transmitted data.

The first step involves high frequency equalization to compensate for losses during transmission in the cable.

The second 20 step involves DC restoration of the equalized signal to recover the low frequency and DC components which are lost through the coupling circuits.

Quantized Peedback (QFB) is the standard technique for DC restoration of an AC coupled binary signal.

The Quantized 25 Peedback method has been utilized in practical implementations for cable equalizers for serial digital transmission applications.

However, the Quantized Peedback method is sensitive to variations in the amplitude and rise time of the input waveform.

As will be appreciated by one skilled in the art, this sensitivity places considerable emphasis 30 on the performance of the high frequency equalizer because as the length of the transmission cable increases, the signal losses become more severe and precise equalization will eventually be impossible.As -2- ~17~7~ a result of this sensitivity, the Quantized Feedback method can introduce jitter into the restored waveform.

SUMMARY OF THE INVENTION The present invention provides a new approach to DC restoration which, unlike the Quantized Feedback technique, can operate correctly over a wide variation of input signal quality, and therefore is inherently better suited for the role of DC restorer in cable equalizers.

In a first aspect, the present invention provides a device for restoring a binary signal having a DC value and having one or more pulses each with a rising edge and a falling edge, the binary signal being coupled to said device by a capacitor having first and second terminals, said device comprising: (a) an input port coupled to 15 the first terminal of the capacitor for receiving the binary signal and producing a coupled binary signal; (b) clamping means for clamping the coupled binary signal when said coupled binary signal exceeds a predetermined range, said clamping means having an input for receiving said coupled binary signal and being connected to the second 20 terminal of the capacitor; (c) means for generating a signal component for controlling the rising and falling edges of the pulses in the coupled binary signal, said means for generating having an output for said signal component and said output being coupled to the second terminal of the capacitor; (d) means for combining said signal 25 component with the coupled binary signal to produce a restored binary signal and said restored binary signal having pulses with rising and falling edges dependent on said signal component, said means for combining being connected to the output of said means for generating and to the second terminal of the capacitor; and (e) means for 30 outputting the restored binary signal, said means for outputting having an input connected to said means for combining and an output connected to the input of said means for generating.

-3- ~ ~ q~ ~i 7 ~ ....

In a second aspect, the present invention provides a device for restoring a binary signal having a DC value and having one or more pulses each with a rising edge and a falling edge, said device being suitable for use with a high frequency equalizer circuit in a cable 5 equalizer system, said device comprising: (a) an input port for receiving the binary signal, said input port being formed from a first terminal of a capacitor for producing a coupled binary signal; (b) clamping means for clamping the coupled binary signal when said coupled binary signal exceeds a predetermined range, said clamping 10 means having an input for receiving said coupled binary signal and said input being connected to a second terminal of said capacitor; (c) means for generating a signal component for controlling the rising and falling edges of the pulses in the coupled binary signal, said generating means having an output for said signal component and 15 said output being coupled to the second terminal of said capacitor; (d) means for combining said signal component to the coupled binary signal to produce a restored binary signal and said restored binary signal having pulses with rising and falling edges dependent on said signal component, said means for combining being connected to the 20 second terminal of said capacitor and to the output of said generating means; (e) output means for outputting said restored binary signal, said output means having an input connected to said means for combining and an output connected to the input of said means for generating; and (f) gain control means for producing a gain control 25 signal, said gain control means having an input coupled to said clamping means, and said clamping means including means for producing an amplitude signal responsive to the amplitude of said coupled binary signal, and said gain control means including means for generating said gain control signal from said amplitude signal 30 received at said input.

In a third aspect, the present invention provides a method for restoring a binary signal having a DC value and having ~,~ ,.i, ,~ 3A ~ ~ ~ 3 ~ ~ _ one or more pulses each with a rising edge and a falling edge, the binary signal being coupled through a capacitor, said method comprising the steps of: (a) inputting the binary signal through the capacitor to produce a coupled binary signal; (b) clamping said coupled 5 binary signal when said coupled binary signal exceeds a predetermined range; (c) generating a signal component responsive to said coupled binary signal for controlling the rising and falling edges of the pulses in said coupled binary signal; and (d) combining said signal component with said coupled binary signal to produce a restored 10 binary signal.

BRIEF DESCRIPTION OF THE DRAWINGS For a better understanding of the present invention, and to show more clearly how it may be carried into effect, reference will 15 now be made, by way of example, to the accompanying drawings which show a preferred embodiment of the present invention.

Figure 1 shows a typical co-axial transmission system in simplified block diagram form;

Figure 2(a) shows a typical binary signal (input waveform) prior to AC coupling;

Figure 2(b) shows the binary signal (of Figure 2(a)) after AC coupling;

Figure 3 shows a known system utilizing the Quantized Feedback method (prior art);

Figure 4 shows in block diagram form a hysteretic bipolar clamp DC restorer according to the present invention;

Figure 5 shows in schematic form the hysteretic bipolar clamp DC restorer of Figure 4;

Figure 6(a) shows a binary signal which has been subject 30 to coaxial cable transmission and AC coupling;

Figure 6(b) shows the intermediate restoration of the binary signal (of Figure 6(a)) according to the present invention;

r ~, ~ . .

~ -3B- ~1 ~ 3 ? ~ Figure 6(c) shows the complete restoration of the binary signal (of Figure 6(a)) according to the present invention;

_,_ } ,j .

~, , ~ 4 2113~ Figure 7 shows in block form an improved cable equalizer utilizing a second embodiment of a hysteretic bipolar clamp DC restorer according to the present invention;

Figure 8 shows a circuit for the DC restorer of Figure 7 5 which is integrated with the automatic gain controller; and Figures 9(a)-(e) show the various voltage and current waveforms associated with the clamping circuit of the hysteric bipolar clamp DC restorer of Figure 8.

10 DETAILED DESCRIPIION OF THE PREFERRED EMBODIMENTS The transmission of serial encoded binary data requires a transmission medium with a wide bandwidth, since the frequency content of such signals is broad.

Co-axial cables are utilized as the transmission medium in typical data transmission systems.

Figure 1 shows a typical known transmission system 1 in simplified block diagram form.

The system 1 comprises a transmitter 2 and a receiver 4.

The transmitter 2 and receiver 4 are coupled together by a co-axial cable 6 which serves as the transmission medium.

In known manner, the transmitter 2 includes electronic circuitry (not 20 shown) for encoding the information to be transmitted into serial binary data.

Similarly, the receiver 4 includes circuitry (not shown) for decoding (e.g. demodulating) the transmitted serial binary data.

In a system 1 where the co-axial cable 6 is AC coupled, the receiver 4 will include a cable equalizer circuit 7 which comprises a restoration circuit 25 8 and a high frequency equalizer 9.

The function of the restoration circuit 8 is to recover the low frequency and DC components of the serial binary signal.

The low frequency and DC components of the serial signal are lost through the AC coupling circuits (not shown) as will be understood by one skilled in the art.

Before describing the 30 restoration circuit 8 according to the present invention, a typical known restoration method and system will be discussed.

'~ 5 2113~6~ The Quantized Feedback method is a known standard technique for DC restoration of an AC coupled binary signal.

In the following description, DC and AC refer to direct current and alternating current respectively.

As will be shown below, the 5 Quantized Feedback (QFB) method is sensitive to variations in the amplitude and rise time of the input binary signal waveform.

This sensitivity places considerable demands on the high frequency equalizer 9, which as is known compensates for losses in the cable by restoring the amplitude of the binary signal.

It will be appreciated by 10 one skilled in the art that as the length of the co-axial cable increases, the signal losses become more severe and precise performance by the high frequency equalizer 9 is no longer possible.

Thus, there will be a performance loss for the QFB method when utilized in cable equalizer 7 applications.

Before explaining the details of the present invention, 15 the Quantized Feedback method and its sensitivity to the shape of the input waveform will be considered in greater detail.

The serial binary signal (not shown in Figure 1) which is transmitted by the co-axial cable 6 is received by the receiver 4 as an input binary signal.

Figure 2(a) depicts an input binary signal VIN prior 20 to AC coupling and Figure 2(b) shows an AC coupled binary signal waveform VAC which represents the input binary signal VIN subsequent to AC coupling.

It will be appreciated by those skilled in the art that AC coupling techniques involve using a coupling capacitor (not shown) and such techniques are common features of co-axial 25 tran~mi~sion systems.

Referring to Figure 2(a), the input binary signal VIN is shown as having a logic '1' defined by the voltage level +Vl, and a logic '0' defined by a voltage level -V1, and a threshold voltage VTH.

The threshold voltage VTH is the voltage level which defines the 30 transition between the logic 1 level and the logic 0 level or vice versa.

As shown in Figure 2(a), the threshold voltage VTH for the input binary signal VIN is taken as the mid-point between the logic 1 and 211376~ logic 0 levels, which in this case is zero volts.

The input binary signal VIN also has a period T and a duty cycle D.

The duty cycle is defined as the duration of the logic '1' pulse, denoted by TH, divided by the period of the binary signal VIN, denoted by T.

A consequence of AC coupling is the loss of the low frequency and the DC components in the input binary signal VIN.

The loss of the DC component can lead to two other effects.

First, the loss of the DC component due to AC coupling can change the duty cycle D which results in jitter.

Secondly, the AC coupling can also lower the 10 noise margin of the input binary signal VIN.

The effects of AC coupling on the duty cycle and noise margin will be considered with reference to the AC coupled binary signal shown in Figure 2(b).

In Figure 2(b), AC coupling of the input binary signal V]N introduces a DC offset voltage VoS.

The DC offset voltage Vos is the 15 difference between the threshold voltage VTH of the input binary signal VIN and the mid-way point of the AC coupled binary signal VAC.

Furthermore, as a result of the DC offset voltage, the logic '1' level for the AC coupled binary signal is defined by a voltage level which is the sum of +Vl and VoS.

Similarly, the logic '0' level for the AC coupled 20 binary signal is defined by a voltage level which is the sum of -Vl and Vos.

As shown in Figure 2(b), the DC offset voltage Vos also changes the duty cycle of the AC coupled binary signal by altering the duration TH of the logic '1' pulse.

The effect of AC coupling and the resultant DC offset 25 voltage Vos will be illustrated by considering the steady state condition.

In the steady state condition, the net current flowing into the coupling capacitor (not shown) must be zero.

This condition can also be expressed by saying that the integral of the AC coupled binary signal VAC (Figure 2(b)) over one cycle is zero.

Using this property of 30 the steady state condition, an expression (1) for determining the off-set voltage Vos as a function of the duty cycle D can be derived.

The off-set voltage Vos is the difference between the original threshold voltage ~7~ 2113rl~13 VTH (i.e. zero volts in this case) and the threshold voltage of the AC coupled binary signal VAC of Figure 2(b). (Taken as the mid-point between the logic '1' and logic '0' levels.) Vos = Vl ~ (1- 2~D) (1) In the above expression, the AC coupled binary signal VAC is expressed as a function of the duty-cycle D.

As is known, it is desirable to eliminate the off-set voltage Vos because it can lower the noise 10 margin.

Furthermore, in practical systems where there are finite rise and fall times, the off-set voltage Vos changes the duty-cycle of the input binary signal VIN which results in jitter.

The jitter problem can be illustrated by considering the duty-cycle D of the original input binary signal VIN and the AC 15 coupled binary signal VAC shown in Figures 2(a) and 2(b) respectively.

Assuming that the input binary signal VIN has equal rise and fall times, denoted by Ts, then the following expression (2) for the dutycycle D' of the AC coupled binary signal VAC can be derived.

D' = D + (vos/vl) ~ (Ts/T) (2) Substituting expression (1) for the off-set voltage Vos derived above, the duty-cycle D' for the AC coupled binary signal VAC can be rewritten as:

D' = D + (1 - 2*D) ~ (Ts/T) (3) It can be seen from the above expressions (2) and (3) that there will be zero jitter (i.e. duty-cycle D = duty-cycle D'), if either the rise-time Ts is 30 zero, or the duty-cycle D of the input binary signal VIN is 0.5.

It will therefore be appreciated that a small rise-time Ts in the input binary signal VIN can help reduce the jitter caused by AC coupling.

However, -8- 21137~û a small rise-time Ts has no effect on the diminished noise margin which is another by-product of the off-set voltage VoS.

The Ouantized Feedback Method (Prior Art) The Quantized Feedback (QFB) method is one known technique which is used for DC restoration of serially transmitted binary signals.

The QPB achieves DC restoration by reducing the offset voltage Vos to as small a voltage as practically possible.

Figure 3 shows a typical known Quantized Feedback (QFB) system indicated by 10 refer~llce 20.

The QFB system 20 restores the DC component in the AC coupled binary signal VAC to produce a DC restored output signal VQ.

The following discussion will demonstrate that while the QFB sy~le"~ of Figure 3 reduces the offset voltage VoS~ it can contribute to jitter in the restored output signal VQ, and the jitter is proportional to both the 15 error between the amplitude of the input binary signal Vl and the amplitude of the comparator output signal V2, and the rise time Ts of the input signal.

The Quantized Feedback system 20 shown in Figure 3 comprises a voltage comparator 22, an input RC network 24, a feedback 20 loop RC network 26, and a summation unit 28.

The input RC network 24, provides one of the inputs to the summation unit 28 and couples the binary signal VIN to the QFB system 20 through a coupling capacitor 30.

The other input for the summation unit 28 is derived from the feedback loop RC network 26, which in turn is coupled to the 25 output of the voltage comparator 22.

In known manner, the voltage comparator 22 produces an output voltage signal Vc.

The feedback loop RC network 26 then filters the output voltage Vc to produce a feedback voltage signal VFB which provides the other input to the summation unit 28.

The summation unit 28 adds the voltage feedback 30 signal VFB to the AC coupled signal VAC that is derived from the input binary signal VIN (i.e. the input binary signal VIN has been integrated using the input RC network 24).

The output of the summation unit 28 ~ -9 211376~ is the restored voltage signal VQ which is the sum of the AC coupled voltage signal VAC and the feedback voltage signal VFB.

Therefore, the function of the prior art QFB system shown in Figure 3 is to reduce the off-set voltage Vos by adding the voltage feedback signal VFB to the AC 5 coupled binary signal VACThe following discussion will demonstrate that theknown Quantized Feedback system 20 shown in Figure 3 introduces jitter.

The jitter is proportional to both the error between the amplitude of the input binary signal VIN and the comparator output 10 signal Vc, and the rise time Ts of the input binary signal VIN.

In the Quantized Feedback system 20 shown in Figure 3, the input RC network 24 has the same time constant as the feedback loop RC network 26 (which provides a low pass filter function).

To simplify the following analysis, the time constant for the input RC 15 network 24 and the feedback loop RC network 26 are assumed to be large so that the feedback voltage signal VFB can be considered to be a DC quantity.

In known manner, the feedback voltage signal VFB can be expressed in terms of the output voltage Vc of the comparator 22 and the duty-cycle DQ of the DC restored output signal VQ from the 20 summation unit 28 as follows:

VFB = VC * (2 " DQ -1) where DQ is the duty-cycle of the DC restored output signal VQ.

As shown previously, the steady state AC coupled binary signal VAC can be expressed as the sum of the original binary signal VIN and the offset voltage VoS~ and using this property the following expression 30 for the DC restored output signal VQ can be written:

VQ = VIN + VOS + VFB (5) 21137~ - 10 (i.e. VAC = VIN + VOS) and the duty-cycle DQ of the DC restored output signal VQ can also be expressed as follows: (where V1 is the amplitude of VIN; V2 is the 5 amplitude of Vc) DQ = D + ((Vos + VFB)/V1) ~ (TS/T) (6) Substituting expression (1) for the offset voltage Vos and expression (4) 10 for the feedback voltage VFB into expression (6) above, expression (7) can be derived for the duty-cycle DQ of the DC restored output signal VQ:

DQ = D ~ (1-2~T~/T) + (1-V2/V1) * T~/T (l-(V2/Vl) * (2~Ts/T)) (7) From the two above expressions for duty-cycle DQ, i.e. (6) and (7), it can be seen that the condition for zero jitter, i.e. DQ = D, is either the requirement that the amplitude V1 of the input binary signal VIN is 20 equal to the amplitude V2 of the comparator output signal Vc, or that the rise time Ts of the input binary signal VIN is zero.

It follows that minimum jitter and maximum noise margin in the DC restored output signal VQ will result when the offset voltage VoS is cancelled exactly by the feedback voltage VFB.

This corresponds to the condition 25 of the amplitude V1 of the input binary signal VIN being equal to the amplitude V2 of the DC restored voltage signal VQ.

Conversely, if equal amplitudes (i.e.

Vl = V2) cannot be maintained, the above expression shows that the Quantized Feedback system 20 (shown in Figure 3) will introduce jitter into the DC restored output signal VQ 30 and the jitter will be proportional to both the error between the amplitudes of the input binary signal VIN (i.e. V1) and the comparator 21~75~ 11 output signal Vc (i.e. V2) and the rise time of the input binary signal VIN (i.e- Ts).

A HYSTERE1~C BIPOLAR CLAMP DC RESTORER (THE PRESENT 5 INVENTION) Reference is now made to Figure 4 which shows in block form the elements of a hysteretic bipolar clamp DC restorer 30 according to the present invention.

The clamp DC restorer 30 comprises a clamping circuit 32, a voltage comparator 34, and a 10 feedback network 36.

The input binary signal VIN is coupled to the DC restorer 30 through a coupling capacitor 38.

One terminal of the coupling capacitor 38 is connected to a summing node 40.

The summing node 40 is formed from the input terminal of the clamping circuit 32, the positive input terminal of the voltage comparator 34 15 and the output terminal of the feedback network 36.

Reference is next made to Figure 5 which shows in detail the circuit elements that comprise the DC restorer 30.

The voltage comparator 34 provides a digital data output 42 for a DC restored output signal VDATA.

The output 42 of the voltage comparator 34 also 20 feeds into the positive terminal of a differential buffer 44.

The differential buffer 44 produces a feedback output current IFB which flows into the summing node 40.

Referring still to Figure 5, the clamping circuit 32 comprises first and second diodes Dl and D2.

The anode of the first 25 diode Dl is connected to the cathode of the second diode D2 and to the summing node 40.

The cathode of diode Dl is connected to the positive terminal of a first clamping voltage reference Ul.

The negative terminal of the clamping voltage reference Ul is connected to signal ground and to the positive terminal of a second clamping 30 voltage reference U2.

The negative terminal of the second voltage reference U2 is connected to the anode of the second diode D2 to complete the clamping circuit 32.

21~376~J - 12 The function of the first voltage reference Ul is to set the clamping voltage for the first diode Dl, that is, the voltage level at which diode Dl will turn-on in response to a signal being present at node 40.

In other words, diode Dl clamps the positive peak of the AC 5 coupled binary signal VAC when the peak voltage value exceeds the value of the first voltage reference Ul.

In a similar fashion, the second voltage reference U2 sets the voltage value for clamping the negative peak of the AC coupled binary signal VAC at the node 40.

Accordingly, the clamp circuit 32 clamps both the positive and negative peaks of the 10 AC coupled binary signal VACAs shown in Figure 5, the voltage comparator 34, thefeedback network 36 and the summing node 40 form a positive feedback loop.

The function of this feedback network 36 is two-fold.

First, the feedback network 36 uses the voltage comparator 34 to 15 "latch" the logic state of the AC coupled binary signal VAC following the last transition of the signal VAC.

The output of the comparator 34 remains latched in this state until there is another transition in the AC coupled binary signal VAC.

The output of the comparator 34 is tapped to provide the output terminal 42 for the DC restored output 20 signal VDATA.

It will be appreciated that the voltage comparator 34 converts the AC coupled binary signal VAC (analog format) into a digital format data output signal, i.e. the positive peak of the DC restored output signal VDATA corresponds to a logic HIGH and the negative peak corresponds to a logic LOW value as determined by the 25 supply settings of the comparator 34.

The second function of the feedback network 36 is to produce the feedback current IFB for restoring the AC coupled binary signal VAC at the summing node 40.

The feedback current IFB is generated by the buffer 44 which takes the output from the voltage 30 comparator 34 as its positive input.

The output of the buffer 44 is connected to the summing node 40 thereby completing the positive 21~37~0 - 13 feedback loop for the feedback current IFB.

At the summing node 40, the feedback current IFB is added to the AC coupled binary signal VAC.

As was discussed above, the DC restorer circuit 30 according to the present invention reduces "jitter" in the AC coupled 5 binary signal VAC.

The implementation of this function will be explained by referring to both Figures 5 and 6.

Figure 6(a) shows the input binary signal VIN which is coupled to the DC restorer circuit 30 by the coupling capacitor 38.

The input binary signal VIN as shown in Figure 6(a) exhibits jitter as a result of unequal rise and fall times, 10 which are denoted by the rising and falling slopes SR and SF respectively.

The amplitude of the input binary signal VIN is 2V1 (i.e.

+Vl to-Vl) peak-to-peak.

One way to achieve zero jitter in the input binary signal VIN is by making the rise time equal to the fall time, i.e. slope SR = 15 slope SF.

This property can be illustrated by considering the characteristics of the input binary signal VIN shown in Figure 6(a).

In Figure 6(a), the bit-width of the input binary signal VIN (as transmitted) is denoted by TH.

The bit-width TH is the interval between the starting points of the low to high and high to low transitions.

The 20 effective or actual bit-width TH' which is the interval between the points where the transitions (i.e. rising slope SR and falling slope SF) cross the threshold voltage VTH (i.e. mid-level voltage value).

Referring to Figure 6(a) the effective bit-width TH' of the input binary signal VIN can be expressed in terms of the bit-width TH and respective 25 delays (denoted by TDR and TDF in Figure 6(a)) between the start of the rising and the falling edges SR, SF and the points where the input binary signal VIN crosses the logic threshold VTH.

Using simple addition, the effective bit-width TH' is:

TH'= TH + TDF - TDR (8) - 14 - 2 1 L 3 7 6 ~ The rise and fall delay times (i.e. TDR and TDF) can be expressed in terms of the rising and falling slopes SR and SF and the amplitude V of the input binary signal VIN, as follows:

TDF = _V1/SF TDR = V1/SR (10) Substituting the above expressions (9) and (10) for the rise and fall 10 time delays TDR and TDF into expression (8) yields the following result for the effective bit-width TH':

TH' = TH - V1/SF - V1/SR (11) = TH - V1 ~ (1/SF + 1/SR) Therefore according to expressions (8) and (11), there will be zero jitter (i.e. TH' = TH) if the rise delay time TDR equals the fall delay time TDF in expression (8), and this requires that the rising slope equal the falling slope, i.e. SR = SF Reference is next made to Figure 6(b) which shows the AC coupled binary signal VAC restored according to the present invention to produce a DC restored binary (i.e. analog) signal denoted by the reference VDC.

The DC restored binary signal VDC is produced at the summation node 40 by the operation of the clamping circuit 32 and the feedback current IFB.

The DC restored binary signal VDC is produced by clamping the AC coupled binary signal VAC and using the feedback current IFB to control the rise and fall times.

Referring to Figure 6(b), the positive peak of the DC restored binary signal VDC is given by the voltage +VCLAMP.

The 30 voltage value +VCLAMP corresponds to the peak value of the AC coupled binary signal VAC which is clamped by diode Dl and voltage reference Ul.

Similarly, the negative value of the DC restored binary ~ - 15 - 2 i ~37 ~ ~ signal VDC is given by the voltage _VCLAMP, which corresponds to the negative peak of the AC coupled binary signal VAC which is clamped by diode D2 and voltage reference U2.

When the AC coupled binary signal VAC is clamped, the feedback current IFB flows into the 5 respective diode, i.e. either Dl or D2.

For the purposes of the following analysis, the diodes Dl and D2 are considered to be ideal, i.e. the forward voltage-drop is negligible.

Another property of the clamping circuit 32 for this analysis is the very high input impedance for voltages within the range +VCLAMP to-VcLAMp.

This high input 10 impedance forces virtually all of the feedback current IFB to flow into the coupling capacitor 38 when the AC coupled binary signal VAC is within the clamping voltage range, i.e. the diode Dl or D2 is not conducting.

The operation of the clamping circuit 32 can be 15 demonstrated by considering the rising edge SR of the DC restored binary signal VDC.

The polarity of the feedback current IFB is negative as long as the input binary signal VIN is less then zero volts.

The polarity of the feedback current IFB is controlled by the voltage comparator 34 and buffer 44 (Figure 5).

At the start of the transition 20 period indicated by reference 46 in Figure 6(b), diode D2 is turned off which causes the feedback current IFB to flow into the coupling capacitor 38.

Since by design the feedback current IFB has a constant value, the effect on the voltage Vc of the capacitor 38 is a linear charging with a rate of IFB/C, where C is the capacitance of capacitor 38.

25 However, this change in the voltage Vc of the coupling capacitor 38 will be opposed by the change in the input binary signal VIN as long as the AC coupled binary signal VAC is negative (i.e. the AC coupled binary voltage VAC controls the polarity of the feedback current IFB)- It follows that the DC restored binary signal VDC will have a slope that is 30 the difference between the transition rate of the input binary signal VIN and the rate of change of the voltage Vc of the coupling capacitor 38.

This is shown in Figure 6(b), where the DC restored binary signal -16- 2i~7~)~ VDC has rising and falling slopes denoted by SRA and SFA respectively.

The rising slope for the DC restored signal VDC can be expressed as:

SRA = SR - IFB/C (12) where SR is the rising slope of the input binary signal VIN and the second term is the charging capacitor voltage Vc resulting from the feedback current IFB.

Similarly, the falling slope for the DC restored binary signal VDC can be expressed as follows:

SFA = _(SF + IFB/C) (13) where SF is the falling slope of the input binary signal VIN (Figure 6(a)) and the second term is charging rate of the coupling capacitor 38 by the feedback current IFB The rising and falling slopes, SRA and SFA, for the DC restored binary signal VDC can be converted to corresponding rising and falling edge delays, TDRA and TDFA~ by using the clamped voltage values +VCLAMP and _VCLAMP of the references U1 and U2 for the peak amplitudes of the DC restored binary signal VDC, as follows:

TDRA = VCLAMP/SRA = VCLAMP/ (SR - IFB/C) (14) TDFA = _VCLAMP/ - SFA = VCLAMP/(SF+IFB/C) (15) Referring still to Figure 6(b), the DC restored binary signal VDC has a bit-width THA which can be expressed in terms of the bitwidth TH of the input binary signal VIN and the rise and fall time delays TDRA and TDFA as follows:

THA = TH + TDFA - TDRA (16) ~ -17- 211376~ Substituting the above expressions (14) and (15) for TDRA and TDFA into expression (16) yields the bit-width of the DC restored binary signal VDC in terms of the voltage value VCLAMP of the reference U1 and U2, the feedback current IFB and the rising and falling slopes i.e. SR and SF:

THA = TH - VCLAMP (1/(SR IFB/C) + 1/(SF+IFB/C)) (1 SR ~ C (or _SF ~ C) >> IFB (18) 10 It then follows from expression (17) above that the rising and falling slopes, SRA and SFA, of the DC restored binary signal VDC are only slightly less steep then those of the input binary signal VIN, i.e. SR ~ SRA and SF ~ SFA because the feedback current IFB is small.

By comparing the expressions (11) and (17) for TH' and THA respectively, it 15 can be seen that any jitter error in the input binary signal VIN is reduced in the DC restored binary signal VDC by the amount approaching the ratio V1/VCLAMP.

This follows from the clamping of the AC coupled binary signal VAC to +VCLAMP and _VCLAMP as discussed above for Figure 6(b).

In the preferred embodiment of the 20 invention, this ratio is typically four or more which produces a reduction in jitter of at least two or three times.

It will be appreciated by one skilled in the art that noise margin considerations will limit the jitter reduction achievable by reducing the clamping voltages values +VCLAMP and _VCLAMP for the references U1 and U2.

Referring next to Figure 6(c), the "latching" function of the voltage comparator 34 will be considered.

The "latching" function of the comparator 34 converts the DC restored binary signal VDC to a DC restored digital output signal indicated by reference VDATA.

The DC restored digital output signal VDATA has logic level "1" and logic level 30 "0" which are set by the voltage comparator 34 output characteristics (i.e. positive and negative voltage rails for comparator 34).

The threshold voltage level for converting the DC restored analog signal 2113~ - 18 VDC to the restored digital signal VDATA is determined by the reference voltage level connected to the negative terminal of the voltage comparator 34, which in this case is set to zero volts.

Therefore, the output of the voltage comparator 34 will remain high (i.e. logic "1") 5 until the DC restored binary signal VDC makes a transition which is less than the reference voltage level on the negative terminal of the comparator 34.

The input characteristics (e.g. high open loop gain) of the voltage comparator 34 can also be used to detect weak AC coupled binary signals VAC, for example signals which have been attenuated as 10 a result of being transmitted over long distances over coaxial transmission lines.

AUTOMATIC GAIN ADJUSTMENT FOR HIGH FREQUENCY EOUALIZATION As discussed for the known system shown in Figure 1, the binary signal VIN which is received from the co-axial cable 6 will have experienced attenuation or amplitude losses.

For accurate data reproduction, the AC coupled binary signal VAC must undergo both high frequency equalization and low frequency equalization (i.e. DC 20 restoration).

To compensate for these losses, the cable equalizer 7 includes the DC restorer 8 and the high frequency equalizer 9 (as shown in Figure 1).

As discussed in detail above, the DC restorer 8, according to the present invention, restores the DC and low frequency components of the AC coupled binary signal VAC.

The high frequency 25 equalizer 9, on the other hand, restores or equalizes the high frequency component of the AC coupled binary signal VA C .

In a second embodiment of the present invention, the DC restorer 8 described above has been modified and integrated with an automatic gain controller for controlling the operation of the high frequency 30 equalizer.

As will be discussed below, the automatic gain controller allows optimum operation of both the modified DC restorer and the high frequency equalizer 9.

~ -19- 2~ ~37 60 Reference is next made to Figure 7 which shows in block form the cable equalizer 7 incorporating a modified DC restorer 50 and an automatic gain controller 52 according to the second embodiment of the present invention.

The automatic gain controller 52 is coupled 5 between the modified DC restorer 50 and the high frequency equalizer 9.

As will be discussed in detail below, the automatic gain controller 52 produces a voltage control signal VAGC which is a function of the difference between a reference current IREF (Figure 8) and a signal current ISIG.

The signal current ISIG is proportional to the amplitude of 10 the AC coupled binary signal VAC.

Reference is next made to Figure 8 which shows in detail the DC restorer 50 and the automatic gain controller 52 according to the second embodiment of the present invention.

Corresponding reference numerals indicate elements corresponding to those of Figure 15 5.

The automatic gain controller 52 (indicated by the broken outline) comprises a current source 54 which produces the reference current IREF, an integrating capacitor CAGC and a voltage output line 56.

The automatic gain controller 52 is coupled to the DC restorer 50 through a current input line 58 which is connected to the current source 54 at 20 node 60.

The magnitude of the reference current IREF should be chosen such that the equalized output of the high frequency equalizer 9 (Figure 7) stabilizes at a level which represents optimal high frequency equalization as will be within the understanding of one skilled in the art.

For a typical high frequency equalizer 9, the reference current IREF 25 should be in the range 200 to 300 micro-amperes.

The automatic gain controller 52 produces a control current IAGC which flows into the integrating capacitor CAGC.

The capacitor CAGC integrates the control current IAGC to produce a voltage control signal VAGC which is fed to the high frequency equalizer via 30 line 56.

The gain control current IAGC is the difference between the reference current IREF and the signal current ISIG taken at node 60.

The signal current ISIG is a current component which flows into the DC ,,~ -20- 21~ 3 ~ restorer 50 via line 58.

The signal current ISIG is produced from the difference between the feedback current IFB and a clamping current ICLAMP~ where the clamping current ICLAMP is a current flowing in the conducting clamping diode Dl or D2.

In Pigure 8, the clamping current 5 ICLA~P corresponds to current ID1 if diode Dl is conducting and current ID2 if diode D2 is conducting.

Recall that diode Dl conducts if the coupled binary signal VAC is positive and diode D2 conducts if the coupled binary signal VAC is negative.

In other words, the automatic gain controller 52 compares the reference current IREF to the difference 10 between the clamped current ID1 or ID2 and the feedback current IFB (i.e.

IsIG=IDl(or ID2)-IFg).

The difference between the reference current IREF and the signal current ISIG is the current IAGC which is integrated by the capacitor CAGC to produce the voltage control signal VAGC.

The voltage control signal VAGC is then applied as negative feedback to 15 control the gain of the high frequency equalizer 9.

Since the current flowing in the clamping diodes Dl or D2 is proportional to the amplitude of the binary signal VAC, the DC restorer 52 provides an elegant implementation for automatic gain control as will become clear from the following explanation.

In the second embodiment of the DC restorer 50 shown in Figure 8, the clamping diodes Dl and D2 (of Figure 5) have been replaced by respective transistors Ql and Q2.

The transistors Q1 and Q2 permit the clamping circuit 32 to be integrated with a current sensor 62 as shown in Figure 8.

The current sensor 62 senses the current flowing 25 in the diode-connected transistor Ql when it is conducting, i.e. the AC coupled binary signal VAC is positive.

When the AC coupled binary signal VAC is negative the diode current ID2 is a direct component of the signal current ISIG The current sensor 62 comprises the combination of 30 diode-connected transistor Q1 (i.e. diode Dl) and a transistor Q1A.

The base of transistor Q1A is connected to the base of diode-connected transistor Ql to form a current mirror as will be recognized by one -21- 21137fi~ skilled in the art.

The current sensor 62 is coupled to the current output line 58 from the automatic gain controller 52.

The clamping current ICLAMP is represented as either current ID1 or current ID2~ depending on which transistor Ql or Q2 is conducting.

The current ID2 5 flows as collector current through transistor Q2 when it is forward biased or conducting.

The other diode current ID1 flows through diodeconnected transistor Ql when it is forward-biased by the AC coupled binary signal VAC present at the summing node 40.

The sensing transistor Q1A has a collector current which is denoted by IC1A. BY the 10 operation of the current mirror comprising diode-connected transistor Ql and sensing Q1A, the collector current IC1A of the sensing transistor Q1A will be equal to the diode current ID1 (i.e. the clamping current ICLAMP) which flows in the diode-connected transistor Ql when it is forward-biased.

When diode-connected transistor Ql is reverse biased 15 or not conducting, the clamping current ICLAMP comprises current ID2 which flows into the collector of transistor Q2 which is forward-biased or conducting because the AC coupled binary signal VAC at the summing node 40 is negative.

Referring still to Figure 8, the clamping current ICLAMP 20 (i.e. current ID1 or current ID2) flows into the summing node 40.

The clamping current ICLAMP which is in excess of the feedback current IFB is due to the clamping of the input binary signal VIN by the clamping circuit 32.

Furthermore, the clamping current ICLAMP (i.e current ID1 or current ID2) is a measure of the amplitude and rise time of the input 25 binary signal VIN.

Therefore, the signal current ISIG is the current which results from a voltage change on the coupling capacitor 38.

The magnitude or pulse height of the signal current ISIG is proportional to the rate of rise of the input binary signal VIN.

The duration (i.e. pulse width) of the signal current ISIG, on the other hand, is proportional to 30 the amount by which the input binary signal VIN exceeds the clamping level (i.e. voltage reference Ul or U2), and inversely proportional to the rate of rise of the input signal VIN.

Accordingly, the average of the -22- 21 137 ~ ~ signal current ISIG, (i.e. pulse height x pulse width), provides an approximate measure of the amplitude of the input binary signal VIN which is independent of the rise time.

These relationships are clearly shown in Figure 9.

Figure 9 shows the waveforms for the currents ID1 and ID2 which flow in the collectors of transistors Ql and Q2 as a result of the input binary signal VIN being clamped at the summing node 40.

The feedback current IFB is the positive feedback current which flows into the clamping circuit 32 ~Figures 5 and 8).

For the input binary 10 signal VIN shown in Figure 9(a), consider the rising edge SR.

The positive rising edge SR of the input signal VIN coupled at node 40 causes diode-connected transistor Ql to conduct, i.e. current ID1 will flow in diode Dl along with the feedback current IFB.

The net current flowing in diode-connected transistor Ql (i.e. ID1 -IFB) is proportional 15 to the amplitude and rise time of input binary signal VIN.

The current ID1 can be calculated from the product of the coupling capacitor 39 capacitance C times the rate of change for the input signal, i.e. ID1 = CxdVIN/dt.

The signal amplitude current ISIG which flows when diode-connected transistor Ql is forward-biased is equal to the current 20 ID1 (i-e- ICLAMP) minus the feedback current IFB.

Since the signal amplitude current ISIG results from the voltage change (i.e. d VIN/dt) across the coupling c~a~t~:3~, the amplitude of the signal amplitude current ISIG is proportional to the rise time of the input binary signal VIN (i.e. rate of change).

On the other hand, the width of the pulse for 25 the signal amplitude current ISIG shown in Figure 9(c) is proportional to the amount by which the input binary signal VIN exceeds the voltage reference U1, and inversely proportional to the rate of rise SR of the input signal VIN.

The second property follows because the faster the rise of a signal, the larger the magnitude which means the 30 narrower the pulse.

Therefore, the average of the signal amplitude current ISIG, i.e. pulse height x pulse width, provides an approximate measure of the amplitude of the input binary signal VIN. A similar 23 - 21 i3 ~ ~ ~ analysis follows for the falling edge (denoted by reference SF in Figure 9(a)) for the input binary signal VIN As will be appreciated by one skilled in the art, the second embodiment of the present invention provides a DC restorer 50 which 5 can be merged with the automatic gain controller 52 to elegantly couple with the high frequency equalizer 9.

The negative feedback (i.e.

VAGC) control of the high frequency equalizer 9, in turn, relaxes the requirement for precision matching and performance in the equalizer 9.

Although various preferred embodiments of the present invention have been described in detail, it will be appreciated by those skilled in the art, that variations may be made to the present invention without departing from the spirit of the invention or the scope of the appended claims.