Transmission system comprising a transmitter and a receiver for transmitting information in a prescribed frequency band, and transmitters and receivers to be used in said system

Beginning of the patent period: October 15, 1969.

The invention relates to a transmission system comprising a transmitting and receiving device for transmitting information in a prescribed frequency band, the information to be transmitted as a whole originating from a main information source and from an associated auxiliary information source having a smaller information content than the main information source. 5 The transmission of the information signals can be directly or after modulation, for example

Amplitude modulation, frequency modulation or pulse modulation, done.

The information to be transmitted can be varied, such as call signals, telegraphy signals u. Likewise, while the associated auxiliary information is more indicative of the character of a sync signal, address signal, notification signal, and the like. Like. Has. Although the information content of the auxiliary information signal 10 is very small, it is usually necessary to devote special attention to the transmission of the auxiliary information signal for a good transmission of the main information signal. For example, in time-division multiplexing systems, part of the available time interval is used exclusively to transmit the auxiliary information in the form of sync pulses, whereas in frequency division multiplexing systems, most of the available frequency band is used exclusively to transmit the auxiliary information in form 15 of a number of pilot frequencies to synchronize the locally generated carrier frequencies.

The invention aims to provide another design of a transmission system of the type mentioned, in which the transmission of the auxiliary information takes place without frequency separation and without time separation in the frequency band intended for transmission of the main information.

The transmission system according to the invention initially mentioned type has the characteristic that the

Auxiliary information signal is formed by a periodic pulse pattern which is within the frequency band of the main information signal and uncorrelated with the main information signal and originates from the auxiliary information source formed as a pulse pattern generator, which pulse pattern in the transmitting device is combined with the main information signal in a linear combining device without frequency separation and without time separation; while in the receiving device the main information signal and the pulse pattern located within its frequency band lie together on a modulation device to which at the same time the locally generated pulse pattern has been applied, originating from a local pulse pattern generator corresponding to the pulse pattern generator in the transmitting device, wherein the output of the modulation device is connected to a smoothing filter in the form of an integrating network whose output signal for automatic phase stabilization is applied to a frequency-determining element of the local pulse pattern generator.

- 2 No. 281924. If the pulse pattern is denoted by a (t), its period T and the main information signal s (t), then the uncorrelation of a (t) and s (t) will be understood to mean that the integral

I (T)

<img file="AT281924B_D0001.tif" />

s (t) a (t - t) dt (D for all values ​​of τ is practically zero, in the formula:

Ι (Τ) ~ O; - »&lt;T &lt;» (Π)

The invention and its advantages will now be explained in more detail with reference to the drawings.

1 shows a transmission system according to the invention, while FIG. 2 shows some timing diagrams for explaining the transmission system in FIG. 1; FIGS. 3 and 4 show variants of the transmission system according to FIG. 1; Figures 5 and 6 show transmission systems according to the invention arranged to transmit a plurality of main information signals over a common time division multiplexed transmission path; FIG. 7 a transmission system according to the invention adapted to transmit a main information signal in the form of pulses; 8 shows an advantageous variant of the transmission system according to FIGS. 7 and 9, some frequency diagrams for explaining the transmission system according to FIG. 8.

In Fig. 1 is a transmission system according to the invention with a transmitting device and a receiving device for direct transmission of a call signal to which a frequency band of, for example, 0 to 3400 Hz is assigned. In this transmission system, the transmission signal originating from a microphone -1- is transmitted to a transmission line ~4 after amplification in a transmission amplifier 2 via a low-pass filter - 3--, while the transmitted conversation signal is received at the receiving end after amplification in a reception amplifier. a playback device -6- is supplied.

In addition to the call signal, an address signal is transmitted at the same time, this for establishing a connection between the transmitting device and the receiving device characterized by a certain address, which exclusively upon receipt of its own address by means of a switch -7 ™ the reproducing device -6- with the Empfangsveistärker -5- combines. Thus, the information to be transmitted in total consists of the call signal originating from a main information source in the form of a microphone -1- and the address signal originating from an auxiliary information source in the form of an address transmitter, the information content of the address signal being much lower than that of the call signal ,

In order to obtain a particularly effective transmission in the described transmission system according to the invention, the auxiliary information signal is formed by a lying within the frequency band of the main information signal and uncorrelated with the main information signal periodic pulse pattern resulting from the auxiliary information source formed as a pulse pattern generator -8-, which pulse pattern in the transmitting device in a linear assembly device - 9 - without frequency separation and without time separation is assembled with the main information signal.

In the in Fig. 1 In the embodiment shown in which the auxiliary information source acts as an address generator, the pulse pattern generator -8- is provided as a feedback shift register -10- with a number of shift register elements -11, 12, 13, 14, 15- whose content is fed to the shift register -10 - connected clock pulse generator -16- with a constant shift period D is pushed, andmeans a modulo-2 sum generator -17 -matched between the third and fourth shift register element -13 and -14, the output of the shift register being connected, on the one hand, to the second input of the modulo-2 sum generator -17 and, on the other hand, to the input of the first Shift register is -10-connected, while at the input of the shift register -10-further a start pulse source -18- is connected. As is known, a modulo-2 sum generator supplies an output pulse only if pulses of different value occur simultaneously at both inputs, and no output pulse if the two simultaneously occurring input pulses are equivalent.

Now, when switching on the pulse pattern generator -8- by the start pulse source -18- the shift register -10- a single start pulse is supplied, this pulse from the clock pulse generator -16- will be pushed through the shift register -10-, and via the feedback of the output to the modulo-2 sum generator -17- and returned to the input wer3

No. 281924, whereby the shift register -10- will generate a series of pulses with a respective returning period due to the feedback. In particular, it can be mathematically demonstrated that the occurring pulse pattern when using n cascaded shift register elements and a suitable choice of the location of the modulo-2 sum generator a period (2ⁿ - 1) D, where D is the length of the shift period; For example, in the pulse pattern generator in Fig. 1, where n = 5, the period T of the pulse pattern (2⁵ In the embodiment described, the pulse pattern having a period 31D occurring at the output of the pulse pattern generator -8- has the shape shown in FIG. 2a, which pulse pattern is effective here as an address signal which is used in the linear assembly device. without frequency separation and without time separation the call signal is added within the call volume from 0 to 3400 Hz.

In the receiving device according to the invention, the main information signal and lying within its frequency band, linearly associated with it pulse pattern together to a modulation device -19- laid, to which at the same time the locally obtained pulse pattern has been laid by a the pulse pattern generator -8- in the transmitter corresponding local pulse pattern generator -8¹-, wherein the output of the modulation device is connected to a smoothing filter-20-connected to the automatic phase correction to a frequency-determining member -21- of the local pulse pattern generator -8'- geschlössen.

In the receiving device shown in Fig.l the local pulse pattern generator -8 '- in a similar manner as the pulse pattern generator -8- formed in the transmitting device, wherein corresponding elements with the same, but with an indexed reference numerals are indicated. Further, the modulation device -19- is designed as a product modulator whose one input to the Empfangsveistärker -5- and the other input to the local pulse pattern generator -8 '- is connected, while the output with a smoothing filter in the form of an integrating network connected -20- whose output voltage controls a frequency corrector -21- designed, for example, as a variable reactance, which is connected to an effective as a local clock generator oscillator -16'-.

In this way, the product modulator 19 on the one hand the received from the call signal s (t) and the pulse signal used as an address signal a (t) existing information signal and on the other hand, the locally obtained pulse pattern supplied, which is probably in shape, but not in phase with the pulse side pattern a (t), which local impulse pattern is denoted by a (t-t), where τ indicates the time delay of the local pulse pattern versus the pulse side generated pulse pattern.

At the output of the integrating network -20-, whose time constant is at least the same order of magnitude as the period T of the pulse pattern a (t), then an output voltage will arise with the value s (t) + a (t). a (t - t) dt s (t). a (t - r) dt (t). a (t - T) dt (ΙΙΓ)

Due to the fact that s (t) is not correlated with a (t), the first integral of the right-hand member of (III) is practically zero for all values ​​τ, so that at the output of the integrating network -20- an integration voltage R (r) arises, which is almost equal to:

'T

R (t) = s (t) + a (t) a (t). a (t - r) dt (XV)

In FIGS. 2b and 2c, the pulse pattern a (t-τ) or the profile of the integration voltage R (τ) as a function of the time delay r of the pulse pattern a (t-τ) with respect to the pulse pattern a (t) is shown in FIG , As is apparent from Fig. 2c, the integration voltage R (τ) will increase in proportion to the time delay τ in the interval -D &lt;τ &lt;O and decrease in proportion to the time delay τ in the interval 0 &lt;t &lt;D, the integration voltage R (r) a maximum value assumes for r = O "ie when the two pulse patterns a (t) and a (t - τ) coincide, while the integration voltage R (τ) a

-4Nr. 281924 has constant minimum value in the interval D &lt;r &lt;T - D. Since the pulse pattern a (t) is periodic with a period T, the integration voltage R (r) has the same periodicity. The integration voltage R (τ) shown in Fig. 2c is supplied as a control voltage to the frequency corrector -21- for phase stabilization of the local clock pulse generator -16'- on the phase of the transmission-side generated pulse pattern a (t).

If at a certain point in time, for example when starting the transmission system, the locally generated pulse pattern a (t - r) has a time delay τ in the interval D &lt;τ &lt;T - D compared with the arrived pulse pattern a (t), then, as in FIG FIG. 2c, an equalizing voltage R (t) of constant value occurs at the frequency corrector -21-, causing no phase lag of the local clock pulse generator -16'-, so that due to the always present frequency differences between the transmit side clock pulse generator -16- and the local clock pulse generator 16 '- the pulse patterns a (t) and a (t - τ) will shift each other. At this time, the shift process continues until the time delay of the local pulse pattern a (t-r) comes within the interval - D &lt;τ &lt;D, at which interval the phase adjustment takes place. When the local clock pulse generator -16'-has a lower frequency than that of the transmit-side clock pulse generator -16-, and thereby causes an increase in the integration voltage R (r) over the frequency corrector -21- an increase in the frequency of the local clock pulse generator -16'- result . at a time delay r of the local pulse pattern within the interval - D &lt;τ &lt;D, the frequency of the local clock pulse generator -16 '- will be brought exactly in line with the frequency of the transmit side clock generator -16- by the increase of the integration voltage. There is still a small mutual time shift between the two pulse patterns a (t) and a (t-r), the magnitude of the shift being determined inter alia by the original frequency difference between the clock pulse generators -16 and 16 '.

At the same time, the increase of the integration voltage R (τ) at the integrating network -20-, which in the foregoing constitutes an indication of the stabilization of the local clock pulse generator -16'-, is used to control the switch-7 preceding the reproducing apparatus -6-. For this purpose, the integrating network -20- is connected via a threshold device to the control circuit of the switch -7-. In this way, the connection between the transmitting device and the receiving device is brought about only upon receipt of the address signal characteristic of the receiving device.

By applying the measures according to the invention for the transmission of the address signal not only additional frequency and period is saved, but it is also realized thereby that the influence of the call quality of the address signal can be reduced to a great extent.

As a result of the integration, the magnitude of the integration voltage R (τ) at the output of the integrating network -20- when the pulse patterns a (t) and a (t-r) coincide becomes proportional to the number of pulses present per period T in the pulse pattern a (t) be, because when you collapse each impulse contributes to the integration. In this way it is possible to perceive this coincidence with great certainty even if the pulse pattern a (t) has a very low level, for example 20 dB below the level of the talk signal s (t). The then already slight influence on the call signal s (t) from the address signal a (t) can be further reduced in that in a linear difference generator -22- of the received, from the call signal s (t) and the pulse pattern a (t) existing information signals to subtract the locally obtained pulse pattern a (t - r), whereby the power of the after this difference generation still in the call signal s (t) remaining address signal is greatly reduced. In particular, by subtracting the local pulse pattern a (t-τ) from the original pulse pattern a (t), the differential voltage shown in FIG. 2d is produced, which is the consequence of the small time delay τ of the local pulse pattern a (t-r) always present during phase stabilization original pulse pattern a (t). This differential generation not only reduces the power of the remaining address signal in the call signal, but at the same time relocates the frequency spectrum of the difference signal to that of the original pulse pattern a (t) at the higher frequencies, which further attenuates using a high attenuation network in FIG Form of a de-emphasis network -23- allows. At the output of the de-emphasis network -23- then remains in the call signal s (t) from the original address signal a (t) (see Fig. 2a) only the low residual signal shown in Fig. 2 e left. For the call signal S (t), a corresponding pre-emphasis network -24- must be used on the transmission side.

- 5 No. 281924

In this way, the application of the measures described by the common effect of integration, difference generation and de-emphasis allows a particularly expedient reduction of the influence of the address signal on the call signal, in a simple way a reduction to 50 to 60 db below the level of the call signal achievable is.

In this way, in the transmission system according to the invention, the transmission of the address signal without frequency separation and without time separation in the call volume, while still the call quality of the address signal is almost not affected.

3 shows a variant of the transmission system according to the invention, wherein with Fig. 1 matching elements are indicated by the same reference numerals.

This transmission system differs from the transmission system illustrated in FIG. 1 in the embodiment of the modulation device -19-; Namely, the modulating device -19- here consists of a modulo-2 sum generator -25-, preceded by a limiter -26-, whereby the incoming information signals are converted into a two-valued signal.

The mode of operation of the receiving device essentially corresponds to that of the receiving device according to FIG. 1, in particular, the integration voltage occurring at the output of the integrating network 20 also has the profile shown in FIG. 2c.

However, in practice, the receiving device shown in Fig. 3 is preferred because the modulo-2 summing generator -25-, preceded by a limiter -26, forms a simpler and more reliable modulation device than the product modulator used in Fig. 1.

4 shows a preferred embodiment of the transmission system according to the invention, wherein the matching with Fig. 1 and 3 elements are again indicated by the same Bezugsziffem..Statt a simply running modulation device as shown in Fig. 1 and 3, a double modulation device is used here ,

In the illustrated embodiment, the modulating device -19- consists of two modulo-2 sum generators-27, 28- connected in parallel with their first input to the limiter -26- and their output terminals connected to a linear difference generator -29- are whose output voltage is supplied to the integrating network -20-. The second input of the modulo-2 sum generator -27- is supplied with the premature by a shift period D local pulse pattern a (t - r + D), while the second input of the modulo-2 sum generator -28- delayed by a shift period D. local impulse patterns a (t - τ - D) are supplied, which premature or delayed local pulse patterns are the outputs of the shift register elements-14 'and 11'-removed. At the output of the integrating network -20- will then arise an integration voltage which as a function of the time delay τ has the course shown in Figure 2f with e in a Radialsymmetrie for τ = O. The control of the playback device -6- preceding switch -7- hiebei done by the output voltage of the modulo-2 sum generator -28- via a smoothing filter -30- in the form of an integrator.

In a corresponding manner, as explained in the transmission systems in Fig. 1 and 3, here a phase stabilization of the local clock pulse generator -16'- is achieved on the phase of the transmission-side generated pulse pattern. However, the double implementation of the modulation device 19 here offers the advantage that the course of the integration voltage shown in FIG. 2f enables the phase delay to have a small time delay r of the local pulse pattern a (t-τ) compared to the original pulse pattern a (FIG. t) can now be returned to virtually zero.

The transmission systems according to the invention described above are each set up to transmit only one call signal as the main information signal, while the auxiliary information signal used is an address signal.

In contrast, in the embodiments shown in Figs. 5 and 6, a large number of main information signals are successively time-division multiplexed via a common transmission path, the auxiliary information signal being used as a synchronizing signal in recovering the individual main information signals at the reception side.

The transmission system according to the invention in Fig. 5 is arranged for the transmission of a number of call signals, each from a separate signal source -31.32 ...... 33 -auskommt, andjeeine

Bandwidth of, for example, 0 to 4 kHz, at the transmitting end of this transmission system, each source -31, 32, ... 33- is transmitted on its own line with an analog-to-digital converter 34, 35,.. in the form of a δ-modulator, to one of the inputs-37,

38 .... 39- connected to a commutator -40-, with the aid of which the call signals are transmitted in digital form one after the other in time multiplex over a transmission path -41-. EmpNr. 281924

On the output side, with the aid of a corresponding commutator, each of the call signals is recovered in digital form from the transmitted time-division multiplex signal and sent to one of the commutator outputs -43, 44, 45, ie via its own line, into which a digital-to-analog converter corresponding to the analog-to-digital converter - 46, 47,. 48-, for example in the form of an integrator associated with the δ-modulator, are connected to a separate consumer station-49,50 ".j51-.

For controlling the transmitting side commutator, the clock pulse generator -16-- in the pulse pattern generator -8-, which is implemented in a similar manner to the previous transmission systems, is simultaneously connected to a control circuit -52- of the commutator -40, wherein Control circuit -52- determines which commutator input -37,38 .... 39-is connected to transmission path -41- at a particular time. The initial position of the commutator -40- in which, for example, the first commutator input -37 ™ is connected to the transmission path 41- is coupled to a particular state of the shift register -10- in the pulse pattern generator -8-, which state is known to be only once per period T. the generated periodic pulse pattern occurs. In the illustrated embodiment, the output of each shift register element -11,12 13,14,15- is connected to a separate input of an AND circuit -53-, which only if at the same time at the output of all shift register elements -11,12,13, 14,15- a pulse appears, outputs an output pulse, which returns via the control circuit -52- the commutator -40- each in its initial position.

On the receiving side, the control of the commutator -42- is carried out in the same way as on the transmitting side, wherein in Fig.5 corresponding elements in the devices used with the same, but provided with an index reference numerals are indicated.

For mutual synchronization of the transmitting and receiving side commutators -40 and 42- is transmitted in this transmission system together with the talk signals at the same time a sync signal for which, as already explained above, no additional frequency and time is required.

For this purpose, the pulse pattern occurring at the output of the pulse pattern generator -8- is added to each call signal within the conversation band from 0 to 4 kHz as a synchronizing signal by means of linear combining devices -54,55 .., 56- without frequency separation and without time separation. On the receive side, the recovered information signals consisting of the talk signals and the sync signals added to each of them are combined in a linear combining device, and fed together with the locally obtained pulse pattern to the modulating device -19- as already described in FIG Way is executed, and the output voltage thereof via the integrator -20- controls the frequency corrector -21- connected to the local clock pulse generator -16'-.

In the manner already explained in detail above, here a phase stabilization of the local clock pulse generator -16'- is obtained on the phase of the pulse side generated pulse pattern, this pulse pattern and the local pulse pattern coincide, and thus the states in which the send and receive shift registers -10, 10'-, which are the same at each instant, whereby an exact synchronization of the transmitting and receiving side commutators -40,42- is obtained.

The influence on the quality of the call by the sync signal can be reduced particularly expedient here by using the measures not already described above and in the figure, again subtracting the local pulse pattern from the recovered information signals and recording the de-emphasis networks. In the transmission system according to FIG. 5 In addition, a further reduction is possible, since in the assembly of the recovered information signals at the receiving end, the same sync signals for each call signal systematically contribute to the output signal of the linear merger-57, whereas the mutually independent call signals provide no systematic, but a more random contribution, so that the transmitting side, the already low level of the sync signal compared to the call signals can be further degraded. In particular, the level of the synchronizing signal can be lowered by a factor / m, where m is the number of call signals to be transmitted.

In this way, in a time-division multiplex transmission system, it is achieved that the whole period is available for the transmission of the call signals, while nevertheless, without increasing the frequency band, accurate mutual synchronization of the transmitting and receiving sides

- 7 No. 281924

Commutators -40,42- is effected.

Fig. 6 shows a variant of the transmission system shown in Fig. 5, wherein corresponding elements are indicated by the same reference numerals. The transmission system in Fig. 6, which is arranged to transmit, for example, a number of remote measurement signals, differs from the transmission system of Fig. 5 in the formation of the time division multiplexed signal and the addition of the sync signal to the remote measurement signals.

On the transmission side, a time division multiplex signal is first formed from the Femmeßsignalen, which is then an analog-to-digital converter in the form of a PCM Encoder --58- is fed, while the receiving side using a corresponding PCM decoding device -59- the original time division multiplex signal is recovered from which the individual Fernmeßsignale be recovered. Further, the pulse pattern used for mutual synchronization of the transmitting and receiving side commutators -40,42- in a linear combining device-60 - is added to the entire time division multiplex signal and not to the individual Femmeßsignalen.

These measures offer the advantage that now the whole frequency band of the time division multiplex signal is available for the effective as a synchronizing signal pulse pattern, instead of a frequency band of a single Femsehmeßsignals, thereby a more sensitive synchronization control, and a faster intervention of the phase control is achieved.

In Fig. 7, a transmission system according to the invention is shown, which is adapted for transmission in a prescribed transmission band of a main information signal in the form of bivalent pulses whose presence and absence characterize the main information signal, and their occurrence times with a row equidistant, for example, from the clock pulse generator -16- Pulse Pattern Generator 8 originating clock pulses coincide. Further, the bivalent pulse signals are arranged here in successive pulse groups each consisting of 31 elements, for example, the first 26 elements in a pulse group contain the actual main information, and the following (31-26) = 5 elements form the parity control of the main information, such as for a cyclic (31, 26) code.

In this transmission system, the pulse signals originating from a pulse source -61- are supplied via a low-pass filter -62- having a cut-off frequency equal to half the clock pulse frequency and a pulse amplifier -63- to a transmission path in the form of a cable -64- and transmitted to the receiving device in which an equalization network -65- for equalizing the amplitude and phase characteristic of the transmission path -64-, a pulse amplifier -66-, a Impulsregenerator -67- for the regeneration of the received signal pulses concerning the shape and the time of occurrence, and a consumer point -68- are included.

In order to make the consumer station recognize the start time of each pulse group, in this transmission system, in addition to the main information signal for marking the start time of a pulse group, a group sync signal is transmitted for which, as already explained in detail above, no additional frequency and period is required ,

A certain state of the shift register -10 in the pulse pattern generator -8- is coupled to the start time of a pulse group, for which in this embodiment a group synchronizing pulse occurring at this time at a separate output of the pulse source -61 is applied to all the shift register elements -11, 12, 13, 14 , 15- is supplied via separate inputs to bring the shift register -10- in that state, in which simultaneously appears at the output of all shift register elements -11,12,13,14,15- a pulse. The pulse pattern occurring at the output of the pulse pattern generator is added to the main information signal within the prescribed transmission band at a level of, for example, 20 dB lower than that of the main information signal in a linear combiner -69-without frequency separation and without time separation.

On the receiving side, the gl ce devices that were used in the transmission systems of Figure 5 and 6 for the recovery of the sync signal used to generate the local group sync signal, with corresponding elements are indicated by the same reference numerals. Also, the phase stabilization of the local clock pulse generator -16 '- on the phase of the transmission-side generated pulse pattern, quite in accordance with the manner described in detail above, with the send and receive shift registers -10,10' - are in every moment in the same state, and thus coincide exactly with the output from the AND circuit -53- and the consumer point -68- supplied group sync pulses with the output from the pulse source -61- group sync pulses. The clock pulses of the local clock pulse generator

No. 281924 -16'- are also used here to control the pulse regenerator -67-.

In order to reduce the risk of insufficient or erroneous phase stabilization, it is advantageous to increase the difference between the main information signal and the group sync signal, both of which are formed by two-valued pulse trains in the transmission system described so far, by converting one of these two-valued pulse trains into a multi-valued pulse train. For this purpose, for example, the bivalent pulse series of the main information signal by means of a transcoder --70 - be converted into a trivalent pulse series. In particular, in this transmission system, the transcoder -70- is transmit side formed by a modulo-2 sum generator -71- with a subsequent linear difference generator-72, the output of the modulo-2 summator being connected via a delay network -73 with a delay time of 2 Clock pulse periods connected to the interconnected second inputs of the modulo-2 sum generator -71- and the linear difference generator -72- is connected, while at the receiving end the original bivalent pulse series is contained by means of a full wave rectifier --74--. To explain the code conversion are in Fig, 7 a pulse train -75- at the input of the transcoder -70- and the associated pulse trains -76,77- at the output of the transcoder. -70- or the full-wave rectifier -74- shown.

Fig. 8 shows an advantageous variant of the transmission system shown in Fig. 7, wherein corresponding elements are indicated with the same Bezugsziffem, are.

In the transmission system of Fig. 8, the delay network -73- accommodated in the transcoder 70 is constituted by a shift register -78- having more than two cascaded shift register elements-79, 80,... 81-whose contents are fed to the shift register - 10 of the pulse pattern generator -8- connected clock pulse generator -16- is pushed further. If the delay network now has a total delay time V, then in the frequency spectrum of the trivalent pulse series at the output of the transcoder -70-spectral zero points occur, k

at the frequencies f = - with k = 0,1,2,3 .... In the preferred embodiment shown in Fig. 8, the number N of shift register elements is -79,80 .... 81- in the transcoder -70- equal to the number of shift periods D occurring per period T of the generated pulse pattern, which means a number of shift register elements N * 31 and a total delay time V = ND = 31 D in the relevant period of the pulse pattern T = 31D.

In this choice of the number N of the shift register elements occur, as shown at a in Fig. 9, for a main information signal s (t) with pulses having a width D in the frequency spectrum of the trivalent pulse series of coded main information signal c (t) simple spectral zero points on the kk

Frequencies f = - = _p with k = 0,1,2,3 ...., while the periodic pulse pattern a (t) with a

Period T = 31 D, as shown at b in Fig. 9 for pulses also having a width D, a Lik k nienspektrum excluding frequency components at the frequencies f = - = - with k = 0,1,2,3. . ,,, so that the frequency components of the periodic pulse pattern a (t) coincide exactly with the simple zero points in the spectrum of the coded main information signal c (t). Further, in Fig. 9, nor the cutoff frequency f_Q = - of the low-pass filter -62- shown.

Characterized in that the shift registers -78 and 10- in the code converter -70 and pulse pattern generator-8-connected in common to the same clock pulse generator -16- is also hei fluctuations in the clock pulse frequency, which are expressed in fluctuations of the disk period D, in the Frequency spectrum avoided any shift of the zero points in the coded main information signal c (t) and the spectrum components of the pulse pattern a (t) with respect to each other.

By applying these measures, a particularly accurate phase stabilization of the local clock pulse generator -16'- is obtained on the phase of the pulse side generated pulse pattern. This is also apparent from the fact that the perturbation element, which indicates the influence of the main information signal s (t) on the control voltage for the phase stabilization occurring at the output of the integrator -20-, in the devices according to FIGS. 1,3,4,5,6 and 7 is given by the integral given in formula (I),

<img file="AT281924B_D0002.tif" />

s (t). a (t - τ) dt,

I (T) =

No. 281924, which is practically zero due to the fact that s (t) and a (t) are virtually uncorrelated, as indicated in formula (Π):

I (t) »0; -« &lt;r &lt;oo, whereas in the device according to Fig. 8, as a result of the transposition of s (t) into c (t), the perturbation element now represented by the integral (τ)

T c (t). a (t - τ) dt O, for all values ​​of r is exactly zero, in formula

I (τ) = Ο, -μ &lt;τ &lt;oo c

The very precise phase stabilization of the local clock pulse generator -16 * - due to the virtually completely interference-free phase control voltage is maintained even with fluctuations in the clock pulse frequency, since the frequency components in the line spectrum of a (t) even then with the simple zero points in the spectrum of c (t) still exactly coincide.

In this simple manner, in the transmission of master information signals in the form of a bivalent pulse train in which the times of occurrence of the pulses coincide with a series of equidistant clock pulses, an influence on the phase stabilization of the local clock pulse generator 16'- by the main information signals to be transmitted is returned to zero and consequently a causes particularly accurate phase stabilization.

Optionally, the level at which the pulse pattern in the transmitting device is supplied to the main information signal can be further lowered without adversely affecting the phase stabilization so that the influence of the pulse pattern on the main information signal is further attenuated. It turns out in practice, however, that such a reduction is not necessary, since the disturbing influence of the pulse pattern on the pulse regeneration in the pulse generator -67- is also already very low as a result of the very well at exact phase stabilization Takti'mpulsfrequenz.

For a different number of shift register elements N in the shift register -78- of the transcoder -70- than that corresponding to the number of times per period T of the pulse pattern occurring shift period D multiplied by an integer m = 1,2,3 .... falls a part of the spectrum components of the pulse pattern together with the spectral zero points of the coded main information signal, which already some improvement of the phase stabilization is achieved. However, the given designation of the number of shift register elements N, given by ND = m T with m = 1, 2, 3,..., Leads to optimum results, because then all the spectrum components of the pulse pattern coincide with the spectral zero points of the coded main information signal coincide.

For the sake of completeness it should be noted that in the transcoder -70- instead of a linear difference generator -72- and a modulo-2 summation generator -71- there is also a linear sum generator and a modulo-2 difference generator consisting of an inverter and a modulo-2 sum generator , are usable. Even then, however, not all of the spectrum components of the pulse pattern coincide with the spectral zeros of the coded main information signal, since these zero points correspond in their frequency position to a series of odd numbers, while the spectrum components of the pulse pattern correspond in their frequency position to a series of natural numbers.

Of course, the pulse pattern shown in Fig. 2a can also be obtained in other ways, for example by means of so-called word generators, which are used to test telegraphy connections. At the same time, the number of shift register elements-11. ... 15- to pulse patterns a (t), as already explained, with a suitable implementation of the feedback via modulo-2 sum generators -17- between the shift register elements -11,... with a period T = (2ⁿ-l) D leads, wherein the integration voltage R (t) in a simple embodiment of the modulation device-19-one of Fig. 2c corresponding course.

Except for the phase stabilization particularly suitable periodic pulse patterns a (t), in which the integration voltage R (r) exclusively for τ- O and τ = T a maximum and in the time-10. 281924 tervall between τ = D and τ = T - D has a constant minimum value, periodic pulse patterns a (t) can also be used in which the integration voltage R (r) again has a maximum for τ = 0 and τ = T, however in the time interval between τ = D and τ = T - D some peaks of smaller amplitudes occur. Any interference with phase stabilization by these smaller amplitude peaks can be easily avoided by incorporating behind the integrator - 20 - a threshold device that does not pass these lower amplitude peaks.