Ultra-wideband communication systems and methods

ULTKA-WIDEBAND COMMUNICATION SYSTEMS AM)METHODS BACKGROUND OFTHEINVENΗON

1. Field of the Invention

The invention relates generally to ultra-wideband αanmunicalioris, and more particularly to systems and methods for cornmumcationiEing ultra-wideband technology.

2. Background

Wireless comraunication systems are proliferating at the ^"Wide Area Network (WAN), Local Area Network (LAN), andPersonal AreaNetwork(PAN) levels. These wireless commiidcationsystemsuseavarietyoftechniquesto allow simultaneous access to multiple users. The most common of these techniques are Frequency Division Multiple Access (EDMA), which assigns specific frequencies to each user, Time Division Multiple Access (TDMA), which assigns particular time slots to each user, and Code DivisionMultiple Access (CDMA), which assigns specific codes to each user. But these wireless communication systems and various modulation techniques are afflicted by a host of problems that limit the capacity and the quality of service providedtoihe users. The Mowirigparagi^lisMefly describe a few of theseproblems for thepurpose of illustration.

One problem that can exist in a wireless communication system is multipath interference. Multipara interference, or multipath, occurs because some of the energy in a transmitted wireless signal bounces off of obstacles, such as buildings or mountains, as it travels tram source to destination. The obstacles in effect create reflections of the teansrnitted signal and the more obstacles there are, the more reflections they generate. The reflections then travel along their own transmission paths to the destination (or receiver). The reflections will contain the same information as tie original signal; however, because of the differing transmission path lengths, the reflected signals will be out of phase with the original signal As a result, they will often combine destructively with the original signal in the receiver. This is referred to as lading. To combat fading current systems typically try to estimate the multipalh effects and then compensate for them in the receiver using an equalizer. In practice, however, it is very difficult to achieve effective multipalh compensatioα

A secondproblemthat can affect the operation of wireless communication systems is interference from adjacent cornrnunication cells within the system. Bi FDMA/TDMA systems, this type of interference is prevent through a frequency reuse plan. Under a frequency reuse plan, available communication frequencies are allocated to communication cells within the communication system such that the same frequency will not be used in adj acent cells. Essentially, the available frequencies are split into groups. The number of groups is tenned the reuse factor. Then the communication cells are grouped into clusters, each cluster containing the same number of cells as there are frequency groups. Each frequency group is then assigned to a cell in each cluster. Thus, if a frequencyreuse factor of7 is used, for example, then a particular communication frequency will be used only once in every seven communication cells. Thus, in any group of seven communication cells, each cell can only use I//¹ of the available frequencies, Le., each cell is only able to use!//¹ ofthe available bandwidth. Ih a CDMA communication system, each cell uses the same wideband cctnmunication channel. Ih order to avoid interference with adjacent cells, each communicafion cell uses a particular set of spread spectrum codes to differentiate cornrnunications wifliin the cell from those originating outside of the cell Thus, CDMA systems preserve the bandwidth in the sense that they avoid reuse planning. But as will be discussed, there are other issues that limit the baidwidthin CDMA systems as well. Thus, in overcoming interference, system bandwidlh is oim sacrificed Bandwidth is becoming a very valuable commodity as wireless communication systems continue to expand by adding more and more users. Therefore, trading offbandwidth for system performance is a costly, albeit necessary, proposition that is inherent in all wireless cxsmmunicationsystems.

The foregoing are just two examples of the types of problems that can affect conventional wireless communication systems. The examples also illustrate that there are many aspects of wireless cornmunication system performance that can be improved through systems and methods that, for example, reduce interference, increase bandwidth, or both.

Not only are conventional wireless cαnmunication systems eflktedby problems, such as those describedinthe preceding paragraphs, but also different types of systems are effected in different ways and to different degrees. ^'Wireless communication systems can be split into three types: 1) line-of-sight systems, which can include point-to-point or point- to-multipoint systems; 2) indoor non-line of sight systems; and3) outdoor systems such as wireless WANs. Line-of-sight systems are least affected by the problems described above, while indoor systems are more affected, due for example to signals bouncing off of building walls. Outdoor systems aie by far the most affected of tie three systems. Because these types of problems are limiting factors in the design of wireless transmitters and receivers, such designs must be tailored to the specific types of system in which it will operate. Ih practice, each type of system implements unique communication standards float address the issues unique to the particular type of system. Even if an indoor system used the same cornmurάcatton protocols andmodulation techniques as an outdoor system, for example, 1he receiver designs would still be different because multipath and other problems are unique to a given type of system and must be addressed with unique solutions. This would not necessarily be the case if cost efficient and effective methodologies can be developed to combat suchpioblems as described above fhatbiiiMmprogtammaM types of systems and still maintain superior performance.

SUMMARYOFTHEINVEMION

In order to combat the above problems, systems and methods of ultra-wideband communicalion are provided. In one ultra-wideband communication method, a communication channel is divided into apluralify of non-overlapping sub-channels by dividing a single serial message intended for an ultra-wideband communication device into a plurality of parallel messages. Each of the plurality of parallel messages are then encoded onto at least some of floe plurality of sub¬ channels, andthen transmitted.

Anoflier embodiment ultra-wideband communication system may operate similar to the one described above, however apluralify of serial messages maybe divided into apluralify ofparallel messages, rather than dividing one serial message into aplurality of parallel messages. Each oflhe plurality of parallel messages may fhenbe encoded onto at least some oflhe plurality of sub-channels, andflientaismitted,

These and otier features and advantages of the present invention will be appreciated from review of the following Detailed Description of Hie Preferred Embodiments, along with the accompanying figures in which like reference numerals are used to describe the same, similar or correspondmgpattsinύieseveralviewsofthe drawings.

BRΓEFDESCRIFΠONOFTHEDRAWINGS

Preferred embodiments of the present inventions taught herein are illustrated by way of example, and not by way of limitation, inthe figures of the accompanying drawings, in which:

Figure IA is a diagram illustrating an example embodiment of a wideband channel divided into a plurality of sub^haτmelsinacoDidancewifliflieinventioiτ;

Figure IB is a diagram illustrating the effects ofmuMpalhinawireless communication system;

Figure 2 is a diagram illustrating another example embodiment of a wideband communication channel divided into aplurality of sub^rjamielsmaccordancewilhihemverilion;

Figure3isadiagιmLiUustra1ingtheappHc^ 1 and2;

Figure 4A is a diagram illustrating Hie assignment of sub-channels tor a wideband communication channel in accordance with the invention;

Figure 4B is a diagram illustrating the assignment of time slots for a wideband communication channel in accordance with the invention;

Figure 5 is a diagram illustrating an example embodiment of a wireless communication in accordance with the invention;

Figure 6 is a diagram illustrating the use of synchronization codes in the wireless communication system of figure 5 in accordance withthe invention;

Figure 7 is a diagram illustrating a correlator that can be used to conelate synchronization codes in the wireless communication system of figure 5;

Figure 8 is adiagramfflustrating synchronization code ∞rreMcninaccoiriancewiiiihe invention;

Figure 9 is a diagram illustrating the cross-correlation properties of synchronization codes configured in accordance withthe invention;

Figure 10 is a diagram illustrating another example embodiment of a wireless communication system in accordance with lie invention;

Figure 1 IA is a diagram illustrating how sub-channels of a wideband communication channel according to the present invention canbe grouped in accordance with the present invention;

Figure 1 IB is a diagram iUnstrating the assignment of the groups of sub- channels of figure 1 IA in accordance wife the invention;

Figure 12 is a diagram iUustratingihe group asεigriments of figure 1 IB in the time domain; Figure 13 is a flow chart illustrating fee assignment of sub-channels base don SIR measurements in fee wireless communication system of figure lOinaccordancewifefeeinvention;

Figure 14 is a logical block diagram of an example embodiment of transmitter configured in accordance wife fee invention;

Figure 15 is a logical block diagram of an example embodiment of a modulator configured in accordance wife feepresentinventionforuseinfeeirarisrrito 14;

Figure 16 is a diagram illustrating an example embodiment of a rate controller configured in accordance wife fee inventionforuse in feemoαϊilator offigure 15;

Figure 17 is a diagram illustrating another example embodiment of a rate controller configured in accordance wife fee inventionforuseinfeemodulator of figure 15;

Figure 18 is a diagram iUustrating an example embodiment of a irequency encoder configured in accordance wife fee invsitionforuse in fcemodulator offigure 15;

Figure 19 is a logical block diagram of an example embodiment of a TDMFDM block configured in accordance wife fee inventionforuseinfeemodulatoroffigure 15;

Figure 20 is a logical block diagram of another example embodiment of a TDMFDM block configured in accordan∞wifefeeinventionibruseinfeemodulatoroffigure 16;

Figure 21 is alogical block diagram of an example embodiment of afi^quencyshifler configuiedinaccoκiance wife fee inventionforuseinfeemodulator offigure 15;

Figure 22 is a logical block diagram of a receiver configured in accordance wife fee invention;

Figure 23 is a logical block diagram of an example embodiment of a demodulator configured in accordance wife fee invention for use in fee receiver offigure 22;

Figure 24 is a logical block diagram of an example embodiment of an equalizer configured in accordance wife fee present invention for use in fee demodulator offigure 23;

Figure 25 is a logical block diagram of an example embodiment of a wireless communication device coiTfiguredinaixordancewifefeeinventiorT;

Figure 26 illustrates different cx)mmunicationmethods; aid

Figure 27 illustrates two ultra-widebandpulses.

DETAIIJEI⁾DESCRImONOFTHEPREFERRED EMBODIMENTS

1. Introduction

Ih fee following paragraphs, fee present invention will be described in detail by way of example wife reference to fee attached drawings. While this invention is capable of embodiment in many different forms, there is shown in the drawings and willhereinbe described in detail specific emlx>diments,wifefee understanding fe^ to be considered as an example of the principles of fee invention and not intended to limit fee invention to fee specific embodiments shown and described That is, throughout this description, the embodiments and examples shown should be considered as exemplars, rather ton as limitations on the present invention. As used herein, Hie "present invention" refers to any one of the embodiments of the invention described herein, and any equivalents. Furthermore, reference to various feature® of the "present invention" throughout this document does not mean that all claimed embodiments or methods must include the referenced, features).

In order to improve wireless ccmmunication system performance and allow a single device to move from one type of system to another, while still maintaining superior performance, the systems and methods described herein provide various communication methodologies that enhance perfomiance of transmitters and receivers with regard, to various common problems that afflict such systems and that allow the transmitters and/or receivers to be reconfigured for optimal performance in a variety of systems. Accordingly, the systems and methods described herein define a channel access protocol that uses a common wideband ∞mmunication channel for all communication cells. The wideband channel, however, is then divided into aplurality of sub-channels. Different subKiharmels are theriassignedtoone or more users within each cell But the base station, or service access point, within each cell transmits one message that occupies the entire bandwidth of the wideband channel. Each user's communication device receives the entire message, but only decodes those portions of the message that reside in sub-channels assigned to the user. For a point-to-point system, for example, a single user maybe assigned all sub-channels and, therefore, has the full wide band channel available to them. In a wireless WAN, on the other hand, Hie sub-channels maybe divided among aplurality of users.

In the descriptions of example embodiments that follow, implementation differences, or unique concerns, relathgtoclhϊei^ types of systems wfflte and methods described herein are applicable to any type of cxammunication systems. In addition, terms such as communication cell, base station, service access point, etc. are used interchangeably to refer to the common aspects of networks at these cϋfrerent levels. To begin iUustrating the advantages of the systems and methods described herein, one can start by looking at the multipath effects for a single wideband communication channel 100 of bandwidth B as shown in figure IA ttmmunications sent over channel 100 in a traditional wireless communication system will comprise digital data bits, or symbols, that are encoded and modulated onto a RF carrier that is centered at frequency^ and occupies bandwidth^. Generally, the width of the symbols (or the symbol duration) Tis defined as 1/B. Thus, if the bandwidth^ is equal to lOQMHz, then the symbol duration Tis definedby the following equation:

T=l/B = l/100megaJwrteβdHZ) = 10nmose∞nώ(ιis). (1)

When a receiver receives the communication, demodulates it, and then decodes it, it will recreate a stream 104 ofdatasymbols 106 as illustrated in figure IB. But the receiver will also receive multipath versions 108 ofthe same data stream. Because multipath data streams 108 are delayed in time relative to the data stream 104 by delays dl, d2, d3, and d4, for example, theymay combine destructively with data steam 104.

A delay spread d_s is defined as the delay from reception of data stream 104 to the reception ofthe last muMpaih data stream 108 that interferes with the reception of data stream 104. Thus, in the example illustrated in figure IB, the delay spread cl_s is equal to delay d.4. The delay spread d_s will vary for different environments. An environment with a lot of obstacles will create a lot of multipafh reflections. Thus, the delay spread d_s will be longer. Experiments have shown that for outdoor WAN type environments, the delay spread 4 canbe as long as 20 microseconds. Usrngthe lQπs symbol duration of equation (1), this translates to 2000 symbols. Thus, with a very large bandwidth, such as 100MHz, multipath interference can cause a significant amount of interference at the symbol level for which adequate compensation is difficult to achieve. This is true even for indoor environments. For indoor LAN type systems, the delay spread d_s is significantly shorter, typically about 1 microsecond. For a 1 Qns symbol duration, this is equivalent to 100 symbols, which ismoreiΗanageablebutsfillsignificant By segmenting the bandwidthi? into aplurality of sub-channels 202, as illustrated in figure 2, and generating a distinct data stream for each sub-channel, the muMpafli effect can be reduced to a much more manageable leveL For example, if the baidwidth b of each sub-channel 202 is 500KHz, then the symbol duration is 2 microseconds. Thus, the delay spread d_s for each sub-channel is equivalent to only 10 symbols (outdoor) or half a symbol (indoor). Thus, by breaking up amessagetliat occupies the entire bandwidth^ into discrete messages, each occupying the bandwidth b of sub-channels 202, a very wideband signal that suffers from relatively minor multipalh effects is created.

Before discussing further features and advantages of using a wideband communication channel segmented into a plurality of sub-channels as described, certain aspects of the sub-channels will be explained in more detail Referring back to figure 2, the overall bandwidth B is segmented into Nsub-channels center at frequencies^ to Thus, the sub¬ <img file="WO2006041947A2_D0001.tif" /><img file="WO2006041947A2_D0002.tif" /> channel 202 that is immediately to the right of/e is oβktfrσmfc by b/2, where b is the bandwidth of each sub-channel 202. The next sub-channel 202 is offset by 3b/2, the next by 5b/2, and so on. To the left of/e, each sub-channel 202 is αfβetby <img file="WO2006041947A2_D0003.tif" /> etc.

Referably, sub-channels 202 are non-overlapping as this allows each sub-channel to be processed independently in the receiver. To accomplish this, a roll-off factor is preferably applied to the signals in each sub-channel in apulse-shaping step. The effect of such apulse-shaping step is illustrated in figure 2 by the non-rectangular shape of the pulses in each subchannel 202. Thus, 1he bandwidth b of each subchannel canbe representedby an equation such as the following: b = (l+r)/T; (2) wherer=theroll-offfactor,and T= the symbol duration.

Without the roll-off factor, i.e., b = IfI, the pulse shape would be rectangular in the frequency domain, which corresponds to a(sin x)ά function in the time domain<img file="WO2006041947A2_D0004.tif" /> 3 in order to fflustratefheproblems associated with arectangularpulse shape andfheneεdto use aroU-off factor.

As canbe seen, main lobe 402 comprises almost all of signal 400. But some of the signal also resides in side lobes 404, which stretch out indefinitely in both directions from main lobe 402. Side lobes 404 make processing signal 400 much more difficult, which increases the complexity of the receiver. Applying a roll-off factor r, as in equation (2), causes signal 400 to decay faster, reducing the number of side lobes 404. Thus, increasing the roll-off factor decreases the length of signal 400, i.e., signal 400 becomes shorter in time. But including the roll-off factor also decreases the available bandwidth in each sub-channel 202. Therefore, r must be selected so as to reduce the number of side lobes 404 to a sufficientnumber, e.g., 15, wMeεtiHmaximizingiheavailablebandwic^

Thus, the overall bandwidthi? for communication channel 200 is givenby

Hie following equation: B=N(IMjIT; (3) or

B=MZT; (4) where M=(IMjN. (5)

For efficiency purposes related to transmitter design, it is preferable tot r is chosen so that Min equation (5) is an integer. Choosing r so thatMis an integer allows for more efficient liansmitters designs using, for example, Inverse Fast Fourier Transform (IFF¹1) techniques. <img file="WO2006041947A2_D0005.tif" /> andNisdwaysaninteger,fliismeanstormustbechosen so fhatNβ) is an integer. Generally, it is preferable for r to be between 0.1 and 0.5. Therefore, ifNis 16, for example, then 0.5 could be selected for r so that N(r) is an integer. Alternatively, if a value for r is chosen in the above example so that N(r) is not an integer, B can be made slightly wider than MfT to compensate. Ih this case, it is still preferable that r be chosen so thaUV^ is approximately an integer. 2. Example Embodiment of a Wireless Communication System

With the above in mind, figure 5 illustrates an example communication system 600 comprising a plurality of cells 602 that each use a common wideband communication channel to communicate with cαiimunication devices 604 within each cell 602. The common communication channel is a wideband communication channel as described above. Each communication cell 602 is defined as the coverage area of a base station, or service access point 606 within the cell One such base station 606 is shown for illustration in figure 6. For purposes of this specification and the claims that follow, the term base station will be used genericaUy to refer to a device that provides wireless access to the wireless communication system for a plurality of communication devices, whether the system is a line of sight, indoor, or outdoor system.

Because each cell 602 uses the same comiΗunication channel, signals in one cell 602 must be distinguishable from signals in adjacent cells 602. To differentiate signals fiom one cell 602 to another, adjacent base stations 606 use different synchronization codes according to a code reuse plan. Si figure 6, system 600 uses a synchronization code reuse factor of 4, although the reuse factor can vary depending on the application.

Preferably, the synchronization code is periodically inserted into a communication from abase station 606 to a communication device 604 as illustrated in figure 6. After apredetemiined number of datapackets 702, in this case two, the particular synchronization code 704 is inserted into the information being transmitted by each base station 606. A synchronization code is a sequence of data bits known to bofti the base station 606 and any communication devices 604 wiHi which it is communicating. The synchronization code allows such a ∞mmunication device 604 to synchronize its timing to that ofbase station 606, which, in turn, allows device 604 to decode the datapropαly. Thus, in cell 1 (see lightly shaded cells 602 in figure 6), for example, synchronization code 1 (SYΝC1) is inserted into data stream 706, which is generated by base station 606 in cell 1, after everytwo packets 702; in cell 2 SYΝC2 is inserted after every two packeis 702; in cell 3 SYNC3 is inserted; andin cell 4 SYNC4 is inserted Use offhe synctonizationcxxles is discussed inmorc detailbelow.

In figure 4A, an example -wideband communication channel 500 for use in communication system 600 is divided into 16 sub-channels 502, centered at frequencies fo to fe A base station 606 at the center of each communication cell 602 transmits a single packet occupying Hie -whole bandwidth B of wideband channel 500. Such a packet is illustratedby packet 504 in figure 4B. Packet 504 comprises sub-packets 506 that are encodedwith afiequency onset corresponding to one of sub-channels 502. Sub-packets 506 in effect define available time slots in packet 504. Similarly, sub-channels 502 canbe said to define available frequency bins in communication channel 500. Therefore, the resources available in communicatioii cell 602 are time slots 506 and frequency bins 502, -which can be assigned to different communication devices 604 within each cell 602. Thus, for example, frequency bins 502 and time slots 506 can be assigned to 4 different communication devices 604 within a cell 602 as shown in figure 4. Each communication device 604 receives the entire packet 504, but only processes those frequency bins 502 and/or timeslots 506 fliat are assigned to it Preferably, each device 604 is assigned non-adjacent frequency bins 502, as in figure 4. This way, if interference corrupts the information in a portion of communication channel 500, then Hie effects are spread across all devices 604 within a cell 602. Hopefully, by spreading out ύie effects of interference in this manner Ihe effects are rninimized and the entire information sent to each device 604 can still be recreated from the unaffected information received in other frequency bins. For example, if interference, such as fading, corrupted the information in binsj^ then each user 14 loses one packet of data But each user potentially receives three unaffected packets from the other bins assignedtofliemHopefu%,theunanecte^ message for each user. Thus, frequency diversity can be achieved by assigning non-adjacent bins to each of multiple users.

Ensuring ttiat the bins assigned to one user are separated by more than Ihe coherence bandwidth ensures frequency diversity. As discussed above, the coherence bandwidth is approximately equal to l/ck For outdoor systems, where ds is typically 1 microsecond, l/d_s = 1/1 microsecond = 1 megahertz (MHz). Thus, the non-adjacent frequency bands assigned to a user are preferably separated by at least IMHz. It is evenmore preferable, however, if the coherence baidwidthplus some guard band to ensure sufficient frequency diversity separate the non-adjacent bins assignedto each user. For example, it is preferable in certain implementations to ensure that at least 5 times the coherence bandwidth, or 5MHz in the above example, separates the non-adjacent bins.

Another way to provide frequency diversity is to repeat blocks of data in frequency bins assignedto aparticular user that are separated by more than the coherence bandwidth. In other words, if 4 sub-channels 202 are assigned to a user, then data block a can be repeated in the first and third sub-channels 202 and data block b can be repeated in the second and fourth sub-channels 202, provided the sub-channels are sufficiently separated in frequency. Si this case, the system canbe said to be using a diversity length factor of 2. The system can similarly be configured to implement other diversity lengths, e.g., 3, 4,..., /. It should be noted that spatial diversity can also be included depending on the embodiment Spatial diversity can comprise transmit spatial diversity, receive spatial diversity, or both. Ih transmit spatial diversity, the transmitter uses a plurality of separate transmitters and a plurality of separate antennas to transmit each message. Ih other words, each trarjsmittertransniitstiiesamemessagehpar^ the receiver. Because the parallel transmissions travel different paths, if one is affected by lading, ύie others will likely not be affected. Thus, when they are combinedin the receiver, themessage shouldbe recoverable evenif one ormore ofthe other transmissionpaths experienced severe lading.

Receive spatial diversity uses a plurality of separate receivers and a plurality of separate antennas to receive a single message. If an adequate distance separates the antennas, then the transmission path for the signals received by the antennas will be different Again, this difference in the transmission paths will provide irnperviousness to lading when theagnalsrromthereceiversarecombmed Transmit and receive spatial diversity can also be combined within a system such as system 600 so that two antennas are used to transmit and two antennas are used to receive. Thus, each base station 606 transmitter can include two antennas, for transmit spatial diversity, and each oommunication device 604 receiver can include two antennas, forreceive spatial diversity. Ifonlytransmit spatial diversityisirnplemented in system 600, then it can be implemented in base stations 606 or in cx)mmunicatioii devices 604. Similarly, if only receive spatial dversity is included in system 600, then it canbe implemented inbase stations 606 or communication devices 604.

The number of ∞imiunication devices 604 assigned frequency bins 502 and/or time slots 506 in each cell 602 is preferably programmable in real time. Si other words, the resource allocation within a communication cell 602 is preferably programmable in the face of varying external conditions, ie., multipath or adjacent cell interference, and varying requirements, i.e., bandwidth requirements for various users within the cell Thus, if user 1 requires the whole bandwidthto download alarge video file, for example, then the allocation ofbins 502 canbe adjusttoprovideuser 1 with more, or even all, ofbins 502. Qαce user 1 no longer requires such large amounts ofbandwidth, the allocation ofbins 502 canbe readjusted among all of users 14.

It should also be noted that all ofthe bins assigned to a particular user canbe used for both the forward and reverse Mc Alternatively, some bins 502 can be assigned as the forward link and some can be assigned for use on the reverse link, depending on the implementation.

To increase capacity, the entire bandwidth^ is preferably reused in each communication cell 602, with each cell 602 being duTermtiated by aurήque synchronic Thus, system 600 provides increased immunity to multipath and fading as well as increasedband width duetothe elimination of irequencyreirø 3. Synchronization

Figure 7 illustrates an example embodiment of a synchronization code comslator 800. When a device 604 in cell 1 (see figure 5), for example, receives an incoming communication firm the cell 1 base station 606, it compares the incoming data with SYNQ in correlator 800. Essentially, the device scans the homing data trying to coπelate the data with the known synchronization code, in this case SYNQ Once correlator 800 matches the incoming data to SYNQ it generates a conelation peak 804 at fbe output Multipath versions of tie data will also generate correlation peaks 806, al1faough1hesepealffi 806areg perform channel estimation, which allows the device to adjust for tie multipath using an equalizer. Thus, in cell 1, if correlator 800 receives a data stream comprising SYNCl, it will generate correlation peaks 804 and 806. Iζ on the oilier hand, the data stream comprises SYNC2, for example, then no peaks will be generated and Ihe device will essentially ignore tie incoming communication.

Even though a data stream that comprises SYNC2 will not create any correlation peaks, it can create noise in correlator 800 that canprevent detection of correMonpeaks 804 and 806. Seveial steps canbetakentopreventihislrom occurring. One way to minimize tie noise created in correlator 800 by signals from adjacent cells 602, is to configure system 600 so that each base station 606 transmits at the same time. This way, tie synchronization codes can preferably be generated in such a manner that only tie synchronization codes 704 of adjacent cell data streams, e.g., streams 708, 710, and 712, as opposed to packets 702 within those streams, will interfere with detection of tie correct synchronization code 704, e.g., SYNCl . The synchronization codes cantienbe further configured to eliminate or reduce tie interference.

For example, the noise or interference caused by an incorrect synchronization code is a function of tie cross correlation of tiat synchronization code with respect to tie correct code. The better the cross correlation between tie two, the lower the noise level When tie cross correlation is ideal, tien tie noise level will be virtually zero as illustrated in figure 8 by noise level 902. Therefore, a preferred embodiment of system 600 uses synchronization codes tiat exhibit ideal cross correlation, i.e., zero. Preferably, tie ideal cross correlation of the synchronization codes covers aperiod /that is sufficient to allow accurate detection of multipath 906 as well as multipath correlationpeaks 904. This is important so tiat accurate channel estimation and equalization can take place. Outside of period /, the noise level 908 goes up, because the data in packets 702 is random and will exhibit low cross correlation with the synchronization code, e.g,, SYNCl. Preferably, period / is actually slightly longer then the multipath length in order to ensure that the multipath can be detected, a Synchronization code generation

Conventional systems use orthogonal codes to achieve cross correlation in correlator 800, Ih system 600 tor example, SYNCl, SYNC2, SYNC3, and SYNC4, corresponding to cells 1-4 (see lightly shaded cells 602 of figure 5) respectively, will all need to be generated in such a manner that they will have ideal cross correlation with each other, ii one embodiment, if the data streams involved comprise high and low data bits, then the value "1" can be assigned to tie highdatabits and "-1" to tie low data bits. Orthogonal data sequences are tientiosethatproduce a "0" outputwhentiey are exclusively ORed (XORed) together in correlator 800. The following example illustrates this point for orthogonal sequences 1 and2: sequence 1: 1 1 -1 1 sequence^ 1 1 1 -1

1 1 -1 -1 = 0

Thus, when the results ofXQRing eachbitpair are added, tieresultis "0". But in system 600, for example, each code must have ideal, or zero, cross correlation with each of tie other codes used in adjacent cells 602. Therefore, in one example embodiment of a method for generating synchronization codes exhibiting the properties described above, the process begins by selecting a "perfect sequence" to be used as Hie basis for the codes. A perfect sequence is one that when correlated with itselfproduces a number equal to the number of bits in the sequence. For example:

Perfect sequence 1: 1 1 -1 1 1 1 -1 1 1 1 1 1 = 4 But each time aperfect sequence is cyclically sbifedby one bit, the new sequence is orthogonal with the original sequence. Thus, for example, if perfect sequence 1 is cyclically shifiedby one bitand then correlated with the original, the correMonproduces a "0" as inthe following example;

Perfect sequence 1: 1 1 -1 1 1 1 1 -1 1 1 -1 -1-0 If the perfect sequence 1 is again cyclically shitted by one bit, and again correlated with the original, then it will produce a "0". In general, you can cyclically shift aperfect sequence by any number ofbits up to its length and correlate the shifted sequence with the original to obtaina "0".

Once aperfect sequence of the correct length is selected, the first synchronization code is preferably generated in one embodiment by repeating the sequence 4 times. Thus, ifperfect sequence 1 is being used, then a first synchronization codejwouldbeihe following:: y= 11-1 1 11-11 11-11 11-11.

Qr in generic form: y-x(0)x(l)x(2)x(3)x(0)x(l)x(2)x(3)x(0)x(lK2)x(3)x(0)x(lK2K3).

ForasequenceoflengthL:^=x(0)xφ.. x(L) x(0)x®.. x(L)x(0)x(l).. Λ(L)X(0)X(1).. x(L).

Repeating the perfect sequence allows correlator 800 abefter cpportraty to detect the synchronization code and allows generation of other uncorrelated frequencies as well. Repeating has the effect of sampling in the frequency domain. This effect is illustrated by the gtaphs in figure 9. Thus, in trace 1, which corresponds to synchronization code;;, a sample 1002 is generated every fourth sample bin 1000. Each sample bin is separated by 1/(4LxJ), where J is the symbol duration. Thus, in the above example, where L = 4, each sample bin is separated by 1/(16x2) in the frequency domain Traces 2-4 illustrate the next three synchronization codes. As can be seen, the samples for each subsequent synchronization code are shifted by one sample bin relative to the samples for the previous sequence. Therefore, none of sequences interfere with each other.

To generate the subsequent sequences, correspondingtotraces2-4, sequence ymustbesbiftedin frequency. This canbe accomplisliedusingthe following equation: £(m) —y(m) *exp(j*2 *τf^r*m/(n *L)), (6) <img file="WO2006041947A2_D0006.tif" /> (time); and where: I(m)= each subsequent sequence; ^z)=thefirstsequence;and n=thenurnberoftimesthesequence is repeated.

Swill be understood that multiplying by an exp(j27i(r*m/ISI)) factor, whereN is equal to the number of times the

I l sequence is i^peatednmultipliedby the length of tie underlyingpeifect sequence L, in the time domain iesults in a shift in the frequmcy domain. Equation (6) results in the desired shift as fflustrated in figure9fcr each of syjκhronization codes 2-4, relative to synchronization code 1. The final step in generating each synchronization code is to append the copies of the last M samples, where M is the length of the multipath, to tie front of each code. This is done to make the convolution with themultipati cyclic andto allow easier detection ofthe multipath.

It should be noted that synchronization codes can be generated from more than one perfect sequence using tie same methodology. For example, a perfect sequence can be generated and repeated for times and then a second perfect sequence can be generated and repeated four times to get an factor equal to eight The resulting sequence can then be shifted as described above to create tie synchronization codes, b. Signal Measurements Using Synchronization Codes

Therefore, when a communication device is at the edge of a cell, it will receive signals from multiple base stations and, tiierefore, will be decoding several synchronization codes at the same time. This can be illustrated witi tie help of figure 10, which illustrates another example embodiment of a wireless communication system 1100 comprising communication cells 1102, 1104, and 1106 as well as communication device 1108, which is in communication with base station 1110 of cell 1102 but also receiving communication from base stations 1112 and 1114 of cells 1104 and 1106, respectively. If communications from base station 1110 comprise synchronization code SYNCl and communications from base station 1112 and 1114 comprise SYNC2 and SYNC3 respectively, tiien device 1108 will effectively receive the sum of tiese three synchronization codes. This is because, as explained above, base stations 1110, 1112, and 1114 are configured to transmit at the same time. Also, the synchronization codes arrive at device 1108 at almost the same time because they are generated in accordance with the description above.

Again as described above, the synchronization codes SYNCl, SYNC2, and SYNC3 exhibit ideal cross conelation. Therefore, when device 1108 correlates the sumx of codes SYNCl, SYNC2, and SYNC3, the latter two wil not interfere with proper detection of SYNCl by device 1108. lmportantiy,1hesumx can also be used to determine important signal characteristics, because tie sum x is equal to the sum of tie synchronization code signal in accordance mMieMowngecpsiioD: (J) <img file="WO2006041947A2_D0007.tif" />

Therefore, when SYNCl is removed, the sum of SYNC2 and SYNC3 is left, as shownin the following: (8) <img file="WO2006041947A2_D0008.tif" />

The energy computed from the sum (SYNC2 + SYNC3) is equal to the noise or interference seen by device 1108. Since the purpose of correlating the synchronization code in device 1106 is to extract tie energy in SYNC 1, device 1108 also has tie energyin the signal from base station 1110, i.e., the energy represented by SYNCl. Therefore, device 1106 can use tie energy of SYNCl and of (SYNC2 + SYNC3) to perform a signal-to-interference measurement for the communication channel over which it is communicating with base station 1110. The result ofthe measurement is preferably a signal-to-interference ratio (SIR). The SIR measurement can then be communicated back to base station 1110 for puφosestiatwillbediscussedbelow. The ideal cross correlation of the synchronization codes, also allows device 1108 to perform extremely accurate determinations of flie Channel Impulse Response (CIR), or channel estimation, from the correlationproducedby correlator 800. This allows for highly accurate equalizationusing low cost, low complexity equalizers, thus overcoming a significant drawback of conventional systems.

4. Sub-Channel Assignments

As mentioned, the SR as determinedby device 1108 canbe cαmmurricatedbackto base station 1110 for use in the assignment of channels 502. Ii one embodiment, due to the feet that each sub-channel 502 is processed independently, Hie SIR for each sub-channel 502 canbe measured and cornmunicatedback to base station 1110. Ia such an embodiment, therefore, sub-channels 502 can be divided into groups and a SIR measurement for each group can be sent to base station 1110. This is illustrated in figure 12A, which shows a wideband communication channel 1200 segmented into sub-channels fo tofe Sub-channels/) to fe are ύien grouped into 8 groups Gl to G8. Thus, in one embodiment, device 1108 andbase station 1110cornmunicateovαachannelsuchasclτaiτnel 1200.

Sub-channels in tie same group are preferably separated by as many sub-channels as possible to ensure diversity. IQ figure HA for example, sub-channels within the same group are 7 sub-channels apart, e.g, group Gl comprises^ andy§. Device 1102 reports a SIRmeasurement for each ofthe groupsGl toG8. These SlRmeasurements are preferably conparedwilhalhreshold value to detemine which sub-channels groups are useable by device 1108. This comparison can occur in device 1108 or base station 1110. If it occurs in device 1108, flien device 1108 can simply report tobase station 1110 which sub-channel groups are useable by device 1108.

SIR reportingwill be simultaneously occurring for a plurality ofdevices within cell 1102. Thus, figure HB illustrates Hie situation where two communication devices corresponding to user 1 and user 2 report SIR levels above the threshold for groups Gl, G3, G5, and G7. Base station 1110 preferably tiien assigns sub-channel groups to used and user 2 based on the SIR reporting as illustrated in Figure HB. When assigning the "good" sub-channel groups to user 1 and user 2, base station 1110 also preferably assigns them based on the principles of frequency diversity. In figure HB, 1herefore,user 1 and user2are alternately assigned every other "good" sub-channeL

The assignment of sub-channels in the fiequency domain is equivalent to the assignment of time slots in the time domain. Therefore, as illustrated in figure 12, two users, used and user2, receive packet 1302 transmitted over communication channel 1200. Figure 12 also illustrated the sub-channel assignment of figure 1 IB. While figure 11 and 12 illustrate sub-channel/time slot assignment based on SIR for two users, the principles illustrated can be extended for aiy number of users. Thus, a packet within cell 1102 can be received by 3 or more users. Although, as the number of available subchannels is reduced due to high SIR, so is Hie available bandwidth. Ih other words, as available channels are reduced, Hie number of users feat can gain access to communic^cαi channel 1200 is also reduced

Poor SIR can be caused for a variety of reasons, but fiequently it results from a device at the edge of a cell receiving communication signals fiom adjacent cells. Because each cell is using the same bandwidth B, Ihe adjacent cell signals will eventually raise the noise level and degrade SIR for certain sub-channels. In certain embodiments, flierefore, sub-channel assignment canbe coordinated between cells, such as cells 1102, 1104, and 1106 in figure 10, in order to prevent interference iram adjacent cells.

Thus, if communication device 1108 is near Hie edge of cell 1102, and device lllδis near flieedgp of cell 1106, then the two can interfere with each other. As a result, the SIRmeasuremenls that device 1108 and 1118 report backto base stations 1110 and 1114, respectively, will indicate that tie interference level is too high. Base station 1110 can then be configured to assign only the odd groups, i.e., Gl, G3, G5, etc., to device 1108, while base station 1114 can be configured to assign the even groups to device 1118. The two devices 1108 and 1118 will then not interfere wMi each other due to the coordinated assignment of sub-channel groups.

Assigning the sub-channels in this manner reduces the overall bandwidth available to devices 1108 and 1118, respectively. In Ihis case the bandvvάdth is reduced byafactoroftwo_^ closer to each base station 1110 and 1114, respectively, will still be able to use all channels if needed. Thus, it is only devices, such as device 1108, that are near the edge of a cell that will have tie available bandwidthreduced Cbntrast this with a CDMA system, for example, in which the bandwidth for all users is reduced, due to tiie spreading techniques used in such systems, by approximately a factor of 10 at all times. I ranbe seen, 1herefόre,ftat1he systems andmefhods for wireless communication over a wide bandwidth channel using a plurality of sub-channels not only improves the quality of service, but can also increase the available bandwidth significantly.

When there are three devices 1108, 1118, and 1116 near the edge of their respective adjacent cells 1102, 1104, and 1106, the sub-channels canbe divided by three. Thus, device 1108, for example, canbe assigned groups GJ, G4, etc, device 1118 canbe assigned groups G2, G5, etc., and device 1116 canbe assigned groups G3, G6, etc. In this case the available bandwidth for these devices, i.e., devices nearthe edges of cells 1102, 1104, andllO6, is reduced byafactor of 3, but this is still better thanaCDMA system, for example.

The manner in which such a coordinated assignment of sub-channels can work is illustratedby the flow chart in figure 13. First in step 1402, a communication device, such as device 1108, reports Ihe SIR for aU sub-channel groups Gl to G8. The SlRs reported are then compared, in step 1404, to a threshold to determine if the SIR is sufficiently low for each group. Alternatively, device 1108 can make the determination and simply report which groups are above or below the SIR threshold. If the SIR levels are good for each group, then base station 1110 can make each group available to device 1108, in step 1406. Periodically, device 1108 preferably measures the SIR level and updates base station 1110 in case the SIR as deteriorated. For example, device 1108 may move from near the center of cell 1102 towaκi the edge, where interference from an adjacent cell may affect the SIR for device 1108.

If the comparison in step 1404 reveals that the SIR levels are not good, then base station 1110 can be preprogrammed to assign either the odd groups or the even groups only to device 1108, which it will do in step 1408. Device 1108 thenieportsiheSlRmeasurementsfortheoddoreven groups itisassignedinstep 1410, and they are again comparedto aSIRthresholdinstep 1412.

Itis assumed thatthepoorSIRlevelisduetoihefactthat device 1108 is operating atthe edge of cell 1102andis therefore being interfered with by a device such as device 1118. But device 1108 will be interfering with device 1118 at the same time. Therefore, the assignment of odd or even groups in step 1408 preferably corresponds with Ihe assignment of tie opposite groups to device 1118, by base station 1114. Accordingly, when device 1108 reports the SIR measurements for whichever groups, odd or even, are assigned to it, the comparison in step 1410 should reveal that the SIR levels ae now below the threshold level. Thus, base station 1110 makes the assigned groups available to device 1108 in step 1414. Again, device 1108 preferably periodically updates the SlRmeasurements by returning to step 1402.

It is possible for the comparison of step 1410 to reveal that the SIR levels are still above the threshold, which should indicate that a third device, e.g., device 1116 is still interfering with device 1108.Miriisc^ be preprogrammed to assign every third group to device 1108 in step 1416. This should correspond with the corresponding assignments of non-interfering channels to devices 1118 and 1116 by base stations 1114 and 1112, respectively. Thus, device 1108 should be able to operate on the sub-channel groups assigned, Le., Gl, G4, etc., without undue interference. Again, device 1108 preferably periodically updates the SlRmeasurements by returning to step 1402. Optionally, athird comparison step (not shown) canbe implemented after step 1416, to ensure that the groups assigned to device 1408 posses an adequate SIR level for proper operation. Moreover, if there are more adjacent cells, i.e., if it is possible for devices in a 4¹¹ or even a 5^to adjacent cell to interfere with device 1108, then the process of figure 13 would continue and the sub-channel groups would be divided even further to ensure adequate SIR levels on the sub-channels assigned to device 1108.

Even though the process of figure 13 reduces the bandwidth available to devices at the edge of cells 1102, 1104, and 1106, the SlRmeasurements canbe used in such amarmer as to increase the data rate and therefore restore or even increase bandwidth. To accomplish this, the transmitters andreceivers used inbase stations 1102, 1104, and 1106, andin devices in communication therewith, e.g., devices 1108, 1114, and 1116 respectively, must be capable of dynamically changing the symbol mapping schemes used for some or all of the sub-charrneL For example, in some embodiments, the symbol mapping scheme canbe dynamically changed among BPSK, QPSK, 8PSK, 16QAM, 32QAM, etc. As the symbol mapping scheme moves higher, i.e., toward 32QAM, the SIR level required for proper operatiαnmoves higher, i.e., less and less interference can be withstood Therefore, once the SIR levels are detemώied for each group, the base station, e.g, base station 1110, can then determine whatsymbolmapping scheme ranbesuppcαted for each sub-channel group and can change the modulation scheme accordingly. Device 1108 must also change the symbol mapping scheme to conespond to that of the base stations. The change can be effected for all groups imfcαmty, or it can be effected for individual groups. Moreover, the symbol mapping scheme canbe changed on just the forward link, just the reverse link, or both, depending on the embodiment

Thus, by maintaining the capability to dynamically assign sub-channels and to dynamically change the symbol mapping scheme used for assigned sub-channels, the systems and methods described herein provide the ability to maintain higher available bandwidths with higher performance levels than conventional systems. To fully realize the benefits described, however, the systems and methods desαribed thus far must be capable of implementation in a cost effect and convenient manner. Moreover, the implementation must include reconfigurability so that a single device can move between different types of communication systems and still maintain optimum performance in accordance with the systems and methods described herein. The Mowing descriptions detail example high level embodiments of hardware implementations configured to operate in accordance with the systems and methods described herein in such a manner as to provide the capabilityjust described above.

5. Sample Transmitter Embodiments

Figure 14 is logical block diagram illustrating an example embodiment of a transmitter 1500 configured tor wireless communication in accordance with the systems and methods described above. The transmitter could, for example be within a base station, e.g, base station 606, or within a communication device, such as device 604. Transmitter 1500 is provided to illustrate logical components that can be included in a transmitter configured in accordance with the systems and methods described herein. 1 is not intended to limit the systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels to any particular transmitter configuration or any particular wireless communication system.

With this inmind, it can be seen that transmitter 1500 comprises a serial-to-parallel converter 1504 configured to receive a serial data stream 1502 comprising a data rate R Serial-to-parallel converter 1504 converts data stream 1502 into iV parallel data streams 1504, where N is the number of sub-channels 202. 1 should be noted that while the discussion that fellows assumes that asingle serial data stream is used, more than one serial data stream can also be used if required or desired. ]ri any case, the datarate of eachparallel data stream 1504 is theniWV Each data stream 1504 is 1hen sentto ascrambler, encoder, and interleaver block 1506. Scrambling encoding, and interleaving are common techniques implemented in many wireless communication transmitters and help to provide robust, secure rommunication. Examples of these techniques will be briefly explained for illustrativepurposes.

Scrambling breaks up the data to be transmitted in an effort to smctofli out fee spec^ density of Ihe transmitted data. For example, if the data comprises along string of Ts, there will be a spike in the spectral density. This spike can cause greater interference within ftie wireless comminication system By brealdrig up smoothed outto avoid any suchpeaks. Often, sαamblmg is acMevedby XOI^g the data wilharandom sequence.

Encoding, or coding the parallel bit streams 1504 can, for example, provide Forward Error Correction (FEQ. The purpose of EEC is to improve the capacity of a rorrmuπication channel by adding some carefully designed redundant information to the data being transmitted through the channel. The process of adding this redundant information is known as channel coding. Convolutional coding and block coding are the two major forms of channel coding. Convolutional codes operate on serial data, one or a few bits at a time. Block codes operate on relatively large (typically, up to arøupleofliundredbytes)messageblocl&lt;s. There are a variety of useful convolutional andblock codes, and a variety of algorithms for decoding the received coded infonnation sequences to recover the original data For example, convolutional encoding or turbo coding with Viterbi decoding is aFEC technique that is particularly suited to a channel in which the transmitted signal is corrupted mainly by additive white gaussian noise (AWGN) or even a channel that simply experiences fading. Convolutiσnal codes are usually described using two parameters: the code rate and the constraint length. The code rate, kfn, is expressed as a ratio of the number of bite into the convolutional encoder (k) to the number of channel symbols (n) output by the convolutional encoder in a given encoder cycle. A common code rate is 1/2, which means that 2 symbols are produced fcr every 1-bit input into the coder. The constraint length parameter, K, denotes the "length" of the convolutional encoder, i.e. how many Mat stages are available to feed tie combinatorial logic that produces tie output symbols. Closely related to K is the parameter m, which indicates how many encoder cycles an input bit is retained and used for encoding afier it first appears at the input to the convolutional encoder. The m parameter can be thought of as the memory lenglhof the encoder.

Interleaving is used to reduce the effects of fading. Interleaving mixes up the order of the data so that if a fade interferes with a portion of the transmitted signal, the overall message will not be effected This is because once the message is de-interleaved and decoded in the receiver, the data lost will comprise non-contiguous portions of the overall message. In other words, the fade will interfere with a contiguous portion of the interleaved message, but when the message is de-interleaved, the interfered with portion is spread throughout the overall message. Using techniques such as EEC, the missing information canthenbe filled in, or the impact offhe lost datamayjust be negligible.

After blocks 1506, each parallel data stream 1504 is sent to symbol mappers 1508. Symbol mappers 1508 apply the requisite symbol mapping, e.g., BPSK, QPSK, etc., to each parallel data stream 1504. Symbol mappers 1508 are preferably programmable so that the modulation applied to parallel data streams can be changed, for example, in response to the SIR reported for each sub-channel 202. It is also preferable, that each symbol mapper 1508 be separately programmable so that the optimum symbol mapping scheme for each sub-channel can be selected and applied to each parallel data stream 1504.

After symbol mappers 1508, parallel data streams 1504 are sent to modulators 1510. Important aspects and features of example embodiments of modulators 1510 are described below. Aftermodulators 1510, parallel data streams 1504 are sent to summer 1512, which is configured to sum the parallel data streaiiis and Ihereby generate a single serial data stream 1518 comprising each of the individually processed parallel data streams 1504. Serial data stream 1518 is then sent to radio module 1512, where it is modulated with an RF carrier, amplified, andliansmitted via antenna 1516 according to known techniques.

The ttansmitted signal occupies the entire bandwidth^ of communication channel 100 and comprises each of the discrete parallel data streams 1504 encoded onto their respective sub-channels 102 within bandwidth B. Encoding parallel data steams 1504 onto the appropriate sub-channels 102 requires that eachparallel data stream 1504 be shifted in <img file="WO2006041947A2_D0009.tif" /> Figure 15 is a logical block diagram of an example embodiment of a modulator 1600 in accordance with the systems and methods described herein. Importantly, modulator 1600 takes parallel data streams 1602 performs Time Division Modulation (TDM) or Frequency Division Modulation (FDM) on each data stream 1602, filters them using filters 1612, and then shifts each data stream in fiequencyusing frequency shifter 1614 so that they occupy the appropriate sub-channel. Filters 1612 apply the required pulse shaping, i.e., they apply the roll-off factor described in section 1. The fequency shifted parallel data streams 1602 are then summed and tansmitted. Modulator 1600 can also include rate controller 1604, frequency encoder 1606, and interpolator 1610. All of the components shown in figure 15 are described in more detail in the following paragraphs andin conjunction withfigures 17-23.

Figure 16 illustrates one example embodiment of a rate controller 1700 in accordance with the systems and methods described herein. Rate control 1700 is used to control the data rate of each parallel data stream 1602. Ia rate controller 1700, the data rate is halved by repeating data streams d(0) to d(7), for example, producing streams α(0) to α(15) in which α(0) is the same as α(8), α(l) is the same as α(9), etc. Figure 16 also illustrates that the effect of repeating the data streams in this manner is to take the data streams that are encoded onto the first 8 sub-channels 1702, and duplicate them on the next 8 sub-channels 1702. As canbeseαi, 7 sub-diarmelssφarate subchannels 1702 comprising the same, or duplicate, data streams. Thus, if fading effects one sub-channel 1702, for example, the other sub-channels 1702 carrying the same data will likely not be effected, i.e., there is frequency diversity between Hie duplicate data streams. So by sacrificing data rate, in this case half the data rate, more robust transmission is achieved. Moreover, the robustness provided by duplicating the data streams d(0) to d(8) can be further enhanced by applying scrambling to the duplicated data streams via scramblers 1704.

It should be noted that the data rate can be reduced by more than half, e.g., by four or more. Alternatively, the data rate can also be reduced by an amount other than half For example if information from n data stream is encoded onto m sub-channels, where m &gt;n Thus, to decrease the rate by 2/3, information from one data stream can be encoded on a first sub-channel, information from a second data stream can be encoded on a second data channel, and the sum or difference of the two data streams can be encoded on a third channel. In which case, proper scaling will need to be applied to the power in the third channel. Otherwise, for example, ύiepowerin the third cliannelcanbe twice Ihepower inthe first two.

Preferably, rate controller 1700 is programmable so that the data rate can be changed responsive to certain operational factors. For example, if the SIR reported for sub-channels 1702 is low, then rate controller 1700 can be programmed to provide more robust transmission via repetition to ensure that no data is lost due to interference. Additionally, different types of wireless αmnunication system, e.g, indoor, outdoor, line-of-sight, may require varying degrees of robustness. Thus, rate controller 1700 can be adjusted to provide the minimum requited robustness for the particular type of communication system. This type of programmability not only ensures robust αmnunication, it can alsobeusedto aHowasingledevicetomovebetwem communication systems andmaiiTlainsuperiorperfbrrnance.

Figure 17 illustrates an alternative e^ andmethods described. In rate controller 1800 the datarate is increased instead of decreased This is aαx)^ serial-te- parallel converters 1802 to convert each data streams d(0) to d(15), for example, into two data streams. Delay circuits 1804 then delay one of the two data streams generatedby each serial-to-parallel converter 1802 by 1/2 a symbol. Thus, data streams d(0) to d(l 5) are transformed into data streams α(0) to α(31). The data streams generated by a particular serial-to-parallel converter 18Q2 and associate delay circuit 1804 must then be summed and encoded onto the appropriate sub-channel For example, data streams a(0) and a(l) must be summed and encoded onto the first sub- channeL Preferably, the data streams are summed subsequentto each data streambeingpulsedshapedby afilter 1612.

Thus, rate controller 1604 is preferably programmable so that the data rate can be increased, as in rate coniroller 1800, or decreased, as in rate controller 1700, as required by a particular type of wireless communication system, or as required by the communication channel conditions or sub-channel conditions. In the event that the data rate is increased, filters 1612 are also preferablyprogrammable so that they can be configured to applypulse snapping to data streams a(0) to aβl), for example, and then sum the appropriate streams to generate the appropriate number of parallel data streams to sendto frequency shifter 1614.

The advantage of increasing the data rate in the manner illustrated in figure 17 is that higher symbol mapping rates can essentially be achieved, without changing the symbol mapping used in symbol mappers 1508. Once the data streams are summed, the summed streams are shifted in frequmcy so that they reside inihe appropriate sub^^ because the number of bits per each symbol has been doubled, the symbol mapping rate has been doubled. Thus, for example, a 4QAM symbol mapping can be converted to a 16QAM symbol mapping even if tie SIR is too high for 16QAMsymbolmapping to otherwise be applied. foo1herwords,programmingrate controller 1800 to increase the data rate in the manner illustrated in figure 17 can increase the symbol mapping even when channel conditions would otherwise not allow it, which in turn can allow a communication device to maintain adequate or even superior performance regardless of the type of communication system

The draw back to increasing the data rate as illustrated in figure 17 is that interferαice is iiicreased, as is receiver complexity. The former is due to the increased amount of data. The latter is due to the fact that each symbol cannot be processed independently because of the 1/2 symbol overlap. Thus, tiese concerns must be balanced against the increase symbol mapping ability when implementing a rate controller such as rate controller 1800.

Figure 18 illustrates one example embodiment of a frequency encαler 1900 in aαxmlance with tie systems and methods described herein. Similar to rate encoding, frequency encoding is preferably used to provide increased communication robustness. In frequency encoder 1900 the sum or difference of multiple data streams are encoded onto each sub-channel This is accomplished using adders 1902 to sum data streams d(0) to d(7) with data streams d(8) to d(15), respectively, while adders 1904 subtract data streams d(0) to d(7) from data streams d(8) to d(15), respectively, as shown Thus, data streams a(0) to a(15) generated by adders 1902 and 1904 comprise information related to more than one data streams d(0) to d(15). For example, a(0 comprises the sum of d(0) and d(8), i.e., d(0) + d(8), while a(8) comprises d(8)-d(0). Therefore, if either a(0) aca(8) isnotreceiveddueto fading, for example, fhenboth of data streams d(0) and d(8) can stillbe retrieved from data stream cι(8).

Essentially, the relationship between data stream d(0) to ^i4)anda(t^toa(2^isamatrkreMonship. Thus,if the receiver knows the correct matrix to apply, it can recover the sums and differences of d(0) to d(15) fkma(0) to a(15). Preferably, frequency encoder 1900 is programmable, so that it can be enabled and disabled in order to provided robustness when required Preferable, adders 1902 and 1904 are programmable also so that different matrices can be appkedtod(0)tod(15).

After fequency encoding if it is included, data streams 1602 are seπttoTDMZFDMblocks 1608. TDMZFDM blocks 1608 perform TDM or EDM on the datastreams as required bythepaώcular embodiment Figure 19 illustrates an example embodiment of aTDMZFDMblock2000 configured toperfomiTDMmaα^streaniTDMZFDMblock 2000 is provided to illustrate the logical components that can be included in a TDMZFDM block configured to perform TDM on a data stream Depending on the actual implementation, some of Hie logical components may or may not be included TDMM)M block 2000 comprises a sub-block repeater 2002, a sub- block scrambler 2004, a sub-block terminator 2006, asub-blockrepeater 2008, andasync inserter 2010.

Sub-block repeater 2002 is configured to receive a sub-block of data, such as block 2012 αmprisingbits a(0) to a(3) for example. Sub-block repeater is tlien configured to repeat block 2012 to provide repetition, wHchin turn leads to more robust ∞mmuncation. Thus, sub-block repeater 2002 generates block 2014, which comprises 2 blocks 2012. Sub-block scrambler 2004 is then configured to receive block 2014 and to scramble it, thus generating block 2016. One method of scrambling can be to invert half ofblock 2014 as illustrated inblock 2016. But other scrambling methods can also be implemented depending on the embodiment

Sub-block terminator 2006 takes block 2016 generated by sub-block scrambler 2004 and adds a termination block 2034 to the fiont ofblock 2016 to form block 2018. Termination block 2034 ensures fcat each block can be processed independently in the receiver. Without termination block 2034, some blocks maybe delayed due to multipath, for example, and they would therefore overlap part ofthenextblock of data But byincludingterminationblock2034,tlie delayedblockcanbeprevented from overlapping any ofthe actual datainthenext block

Termination block 2034 can be a cyclic prefix termination 2036. A cyclic prefix termination 2036 simply repeats the last few symbols ofblock 2018. Thus, for example, if cyclic prefix teπnination 2036 is three symbols long, then it would simply repeat the last three symbols ofblock 2018. Alternatively, termination block 2034 can comprise a sequence of symbols that ate known to both the transmitter and receiver. The selection of what lype ofblock termination 2034 to use can impact what type of equalizer is used in the receiver. Therefore, receiver complexity and choice of equalizers must be considered whmdetermirmgwliat type ofterniinationblock 203

After sub-block terminator 2006, TDMZFDM block 2000 can include a sub-block repeater 2008 configured to perform asecondblockiepetition step in whichblock2018 is repeated to formblock2020. In certain ernbodimerits, sub- block repeater can be configured to perform a second block scrambling step as welL After sub-block repeater 2008, if included, TDM/FDM block 2000 comprises a sync inserter 210 configured to periodically insert an appropriate synchronization code 2032 after a predetermined number of blocks 2020 and/or to insert known symbols into each block ThepurposeofsynckOnizationcoά^2032isdiscussedinsection3.

Figure 20, on the other hand, illustrates an example embodiment of a TDMZFDM block 2100 configured for FDM, which comprises sub-block repeater 2102, sub-block scrambler 2104, block coder 2106, sub-block tansformer 2108, sub-block terminator 2110, and sync inserter 2112. As with TDMZFDM block 2000, sub-block repeater 2102 repeats block 2114 and generates block 2116. Sub-block scrambler Hien scrambles block 2116, generating block 2118. Sub-block coder 2106 takes block 2118 and codes it, generating block 2120. Coding block correlates the data symbols together and generates symbols b. This requires joint demodulation in tie receiver, which is more robust but also more complex. Sub-block transformer 2108 then performs a transformation on block 2120, generating block 2122. Preferably, the transfcrmation is an IBFT of block 2120, which allows for more efficient equalizers to be used in the receiver. Next, sub-block terminator 2110 temώiates block 2122, generating block 2124 and sync inserter 2112 periodicallyinserts asyQchronizationcode2126 after acertamnurriberofblocks2124 and/or insert known symbols into each block Preferably, sub-block terminator 2110 only uses cyclic prefix temώiation as described above. Again this allows formore efficient receiver designs.

TDMZEDM block 2100 is provided to illustrate the logical compmenls that can be included in a TDM/TOM block configured to perform EDM on a data stream Depending on the actual implementation, some of the logical components may ormaynot be included Moreover, TDM/FDMblock2000 aid2100 areprefeiablyprograinnableso that the appropriate logical components can be included as required by apatticular implementation. This allows a device tiat incorporates one ofblocks 2000 or 2100 to move between different S}«teniswi1ho^erent requirements. Further, it is preferable that TDMZEDM block 1608 in figure 15 be programmable so that it can be programmed to tjerfomi TDM, such as described in conjunction with block 2000, or EDM, such as described in conjunction with block 2100, as requiredby aparticular communication system.

After TDMZEDM blocks 1608, in figure 15,theparaMdatastieamsarepreferablypasse^ 1610.

After Interpolators 1610, the parallel data streams are passedto filters 1612, which apply the pulse shapping described in ∞njunctionwithiheroU-offfactorof equation (2) insection 1. Then the parallel data streams are sent to trequencyshiiter 1614, which is configured to shift each parallel data stream by the frequency offset associated with the sub-channel to which theparticularparallel data streamis associated. Figure 21 illustrates an example embodiment of a frequency shifter 2200 in accordance with the systems and methods described herein. As can be seen, frequency shifter 2200 comprises multipliers 2202 configured to multiply each parallel data stream by the appropriate exponential to achieve the required frequency shift. Each exponential is of the form: eψ(j27φtT/rM), where c is the corresponding sub-channel, e.g., c = 0 to N-I, and n is time. Preferably, frequency shifter 1614 in figure 5 is trogrammable so that various chanrielZsub-channel configurations can be accommodated for various different systems. Alternatively, an IFFT block can replace shifter 1614 and filtering canbe done after the IFFT block This type of implementation canbe more efficient depending on the implementation.

After the parallel data streams are shifted, they ais summed, e.g., in summer 1512 of figure 14. The summed data stream is then transmitted using the entire bandwidth B of the cxarrrnunication channel being used. But tie transmitted data stream also comprises each of the parallel data streams shifted in frequency such that they occupy the appropriate sub-channeL Thus, each sub-channel may be assigned to one user, or each sub-channel may carry a data stream intended for different users. The assigιmieπtofsub&lt;;hatmels is described in section 3h channels are assigned, however, each user will receive the entire bandwidth, comprising all the sub-channels, but will only decode those sub-channels assigned to the user.

6. Sample Receiver Embodiments

Figure 22 illustrates an example en±xxliment of areceiver 2300 that can be configured in accordance with the present invention Receiver 2300 cαnprises an antenna 2302 configured to receive a message transmitted by a tansmitter, such as transmitter 1500. Thus, antenna 2302 is configured to receive a wide band message rømprising the entire bandwidth B of a wide band channel that is divided into sub- channels of bandwidth b. As described above, the wide band message comprises a plurality of messages each encoded onto each of a corresponding sub-channel All of tie sub-channels may or may not be assigned to a device that includes receiver 2300; Therefore, receiver 2300 may or nmynotberequiredtodecOdeallofthesub-channels. Afierthe message is received by antenna 2300, it is sent to radio receiver 2304, which is configured to remove the carrier associated with the wide band ccranunication channel and extract a baseband signal comprising the data stream tHismitted by the transmitter. The baseband signal is ten sent to correlator 2306 and demodulator 2308. Correlator 2306 is configured to correlated with a synchronization code inserted in the data stream as described in section 3.1 is also preferably configured to perform SIR aMmiitipathe described in section 3(b). Demodulator 2308 is configured to extract the parallel data streams fiom each sub-channel assigned to the device ∞mprising receiver 2300 and to generate a single data stream therefrom.

Figure 23 illustrates an example embodiment of a demodulator 2400 in accordance with the systems and methods descnbed herein Demodulator 2402 comprises a fiequency shifter 2402, which is configured to apply a fiequency offset to the baseband data stream so that parallel data streams comprising Hie baseband data stream can be independently processed in receiver 2400. Thus, the output of frequency shifter 2402 is a plurality of parallel data streams, which are then preferably filtered by filters 2404. Filters 2404 apply a filter to each parallel data stream that corresponds to the pulse shape applied in the transmitter, e.g., transmitter 1500. Atematively, an IFFT block can replace shifter 1614 and filtering can be done after the IFFT block. This type of implementation canbemore efficient depending on the implementation.

Next, receiver 2400 preferably includes decimators 2406 configured to decimate the data rate of Ihe parallel bit streams. Sampling at higher rates helps to ensure accurate recreation of Hie data But the higher the data rate, the larger and more complex equalizer 2408 becomes. Thus, the sampling rate, and therefore the number of samples, can be reduced by decimators 2406 to an adequate level that allows for a smaller andless costly equalizer2408.

Equalizer 2408 is configured to reduce the effects of multipath in receiver 2300. Its operation will be discussed more fully below. After equalizer 2408, the parallel data streams are sent to de-scrambler, decoder, and de-rnterleaver 2410, which perform the opposite operations of scrambler, encoder, and interleaver 1506 so as to reproduce the original data generated in the transmitter. The parallel data streams are then sent to parallel to serial converter 2412, which generates a single serial data stream from the parallel data streams. Equalizer 2408 uses the multipath estimates provided by correlator 2306 to equalize Hie effects of rnultipafh in receiver 2300. In one embodiment, equalizer 2408 comprises Single-In Single-Out (SBO) equalizers operating on each parallel data stream in demodulator 2400. Jn this case, each SISO equalizer comprising equalizer 2408 receives a single input and generates a single equalized output Alternatively, each equalizer can be aMuMple-SiMultiple-Out (MMO) or a Multiple-lh Single-Out (MSO) equalizer. Multiple inputs can be required for example, when a frequency encoder or rate ∞ntroller, such as frequency encoder 1900, is included in the transmitter. Because frequency encoder 1900 encodes information from more than one parallel data stream onto each subchannel, each equalizers comprising equalizer 2408 need to equalize more than one sub-channel. Thus, for example, if a parallel data stream in demodulator 2400 comprises d(l) + d(8), then equalizer 2408 will need to equalize both d(l) and d(8) together. Equalizer 2408 can then generate a single output corresponding to d(l) or d(8) (MSO) or it can generate both d(l) and d(8) (MMO).

Equalizer 2408 can also be a time domain equalizer (IDE) or a frequency domain equalizer (EDE) depending on Hie embodiment Gmerally, equalizer 2408 is a TDE if 1he modulator in Iheiransmife^ data streams, and aEDE if the modulator performs EDM But equalize 2408 canbe an EDEevenifTDMis used in Ihe transmitter. Therefore, the preferred equalizer type should be taken into consideralion when deciding what type ofblock termination to use in flie transrnitter. Because ofpower requiremeiits, it is oflenpreferable to use EDM on the forward link and TDM on the reverse link in a wireless ∞mmunication system

As with transmitter 1500, tie various components comprising demodulator 2400 arepreferablyprogramrnable, so that a single device can operate ina plurality of different systems and still maintain superior performance, which is a primary advantage of the systems and methods described herein. Accordingly, the above discussion provides systems and methods for implementing a channel access protocol that allows the transmitter and receiver hardware to be l^rogrammedsHghtlydependhgonthe∞mmumcationsystiaii

Thus, when a device moves from one system to another, it preferably reconfigures the hardware, i.e. transmitter and receiver, as required and switches to a protocol stack corresponding to the new system. An important part of reconfiguring the receiver is reconfiguring, or programming, the equalizer because multipaih is a main problem for each type of system The multipath, however, varies depending on Hie type of system, which previously has meant that a different equalizer is required for different types of cxjmmunication systems. The channel access protocol described in the preceding sections, however, allows for equalizers to be used that need only be reconfigured slightly for operation in various systems. a Sample EcrualizαRiibodiment

Figure 24 illustrates an example embodiment of a receiver 2500 illustrating one way to configure equalizers 2506 in accordance with the systems andmethods describedherein Before discussing the configuration ofreceiver 2500, it should be noted that one way to configure equalizers 2506 is to simply include one equalizer per channel (for the systems and methods described herein, a channel is the equivalent of a sub-channel as described above). A correlator, such as coπelator 2306 (figure 22), can then provide equalizers 2506 with an estimate of the number, amplitude, and phase of any multipafhs present, up to some maximiπn number. This is also known as tie Channel Impulse Response (CIR). The maximum number of multipaths is determined based on design criteria for a particular implementation. The more multipafhs included in the CIR the more path diversity the receiver has and Hie more robust communication in Hie system willbe. Path diversity is discussed alMemore fully below.

If there is one equalizer 2506 per channel, Hie CIR is preferably provided directly to equalizers 2506 tram the correlator (not shown). E such a correlator configuration is used, then equalizers 2506 can be run at a slow rate, but the overall equalizationprocess is relatively fast For systems with arelatively small number of channels, such a configuration is therefore preferable. The problem, however, is that there is large variances in the number of channels used in different types of communication systems. For example, an outdoor system can have has many as 256 channels. This would require 256 equalizers 2506, which would make the receiver design too complex and costly. Thus, for systems with a lot of channels, the configuration illustrated in figure 25 is preferable. Si recdver25C)0,multrρlecrjarine]s share each equalizer 2506. For example, each equalizer can be shared by 4 channels, e.g., Chl-Ch4, Ch5-Ch8, etc., as illustrated in figure 25. In which case, receiver 2500 preferably comprises a memory 2502 configured to store information arriving on each channel.

Memory 2502 is preferably divided into sub-sections 2504, which are each corώguredto store information fora particular subset of channels. Information for each channel in each subset is then alternately sent to the appropriate equalizer 2506, which equalizes the information based on the CIR provided for that channel Ih this case, each equalizer must run much faster than it would if there was simply one equalizer per channel. For example, equalizers 2506 would needtomn4 ormore times as fast in orderto effectively equalize 4 channels as opposedto 1. Ih addition, exteamemory 2502 is required to buffer the channel information. But overall, the complexity of receiver 2500 is reduced, because there are fewer equalizers. This should also lower the overall cost to implement receiver 2500.

Preferably, memory 2502 and the number of channels that are sentto aparticular equalizer is programmable. Si this way, receiver 2500 can be reconfigured for the most optimum operation for a given system. Thus, if receiver 2500 were moved from an outdoor system to an indoor system with fewer channels, then receiver 2500 can preferably be reconfigured so that there are fewer, even as few as 1, channel per eqiializer. The rate at wrich equalizers 2506 are ruti is also prefαablyprogiHrrαiable such that equalizers 2506 canberun at the optimum rate for the number of channels being equalized.

In addition, if each equalizer 2506 is equalizing multiple channels, then the CIR for those multiple paths must alternately be provided to each equalizer 2506. Preferably, therefore, a memory (not shown) is also included to buffer the <img file="WO2006041947A2_D0010.tif" /> each equalizer firm the QRmernory (not shown) when the corresponding channel information is being equalized. The CIR memory (not shown) is also preferably programmable to enstire optimum oper^on

Returning to the issue of path diversity, the number of paths usedbyeqiralizers 2506 must account for the delay spread d_s in the system. For example, if the system is an outdoor system operating in the 5gigahertz (GHz) range, the communication channel can comprise a bandwidth of 125 megahertz (MHz), e.g, the channel can extend from 5.725 GHzto 5.85GHz. Ifthe channel is dividedinto 512 sulMiiannels wilharoU-offfactorrof .125, then each subchannel will have a bandwidth of approximately 215 kϋohertz (KHz), which provides approximately a 4.6 microsecond symbol duration Since the woistease delay spread d_s is 20 microseconds, the number of paths usedby equalizers 2504 canbe set to amaximum of 5. Thus, there would be a first path Pl at zero microseconds, asecondpathP2 at 4.6 microseconds, a thirdpathP3 at 92 microseconds, a fourthpathP4 at 13.8 microseconds, and fiffhpathP5 at 18.4 microseconds, whichis close to the delay spread cb Ih another embodiment, a sixth path can be included so as to completely cover the delay spread d^ however, 20 microseconds is the worst case. Bi feet, a delay spread d_s of 3 microseconds is a more typical value. Ih most instances, therefore, the delay spread d_s will actually be shorter and an extra path is not needed Alternatively, fewer sub-chanriels can be used, thus providing a larger symbol duration, instead of using an extra path. But again, this would typically not be needed

As explained above, equalizers 2506 are preferably configurable so fliat fliey can be reconfigured for various communication systems. Thus, for example, the number of paths used must be sufficient regardless of the type of communication system. But this is also dependent on the number of sub-channels used If, for example, receiver 2500 went fiom operating in the above described outdoor system to ai indoor system, where the delay spread d_s is on the order of 1 microsecond, then receiver 2500 can preferably be reconfigured for 32 sub-channels and 5 paths. Assuming the same overall bandwidth of 125 MHz, the bandwidth of each sub-channel is approximately 4MHz and the symbol dm^onisapproximately250nanoseconds.

Therefore, there will be a first path Pl at zero microseconds and subsequent paths P2 to P5 at 250ns, 500ns, 750ns, and 1 microsecond, respectively. Thus, the delay spread ds should be covered for the indoor environment Again, the 1 microsecond delay spread d_s is worst case so the 1 rrflcroseconddelayspread^providedin the above example will often be more than is actually required This is preferable, however, for indoor systems, because it can allow operation to extend outside of the inside environment, e.g., just outside the building in which the inside environment operates. For campus style environments, where auser is likely tobetravelingbetweenbuildings, this canbe advantageous. 7. Sample Embodiment of a Wireless Cbmmunication device

Figure 25 illustrates an example embodiment of a wireless communication device in accordance with 1he systems and methods described herein Device 2600 is, for example, a portable communication device configured for operation in aplurality of indoor and outdoor communication systems. Thus, device 2600 comprises an antenna 2602 for toismitting and receiving wireless communication signals over a wireless communication channel 2618. Duplexor 2604, or switch, can be included so that transmitter 2606 and receiver 2608 can both use antenna 2602, while being isolated fiom each other. Duplexors, or switches used for this purpose, are well known and will not be explained herein.

Transmitter 2606 is a configurable transmitter configured to implement Hie channel access protocol described above. Thus, transmitter 2606 is capable of tiHismitting and encoding a wideband communication signal αmprising a plurality of sub-channels. Moreover, transmitter 2606 is configured such that the various sub-components that comprise teansrnitter 2606 can be reconfigured, or programmed, as descώedmsecΛcm5. SimiMy,iecdva2608isconfigtnEdto implement tie channel access protocol described above and is, therefore, also configured such tiat the various sub¬ components cornprisingreceiver 2608 canbe reconfigured, orreprograrnmed, as described in section 6.

Transmitter 2606 andreceiver2608 are interfaced witiprocessor 2610, which can ∞mprise various processing, coniroEer, and/or Digital Signal Processing (DSP) circuits. Processor 2610 controls the operation of device 2600 including encoding signals to be transmitted by transmitter 2606 and de∞ding signals recdved by receiver 2608. Device 2610 can also include memory 2612, which can be configured to store operating instructions, e.g., firmware/software, usedbyprocessor 2610 to control the operation of device 2600.

Processor 2610 is also preferably configured to reprogram teansrnitter 2606 and receiver 2608 via control interfaces 2614 and 2616, respectively, as required by the wireless communication system in which device 2600 is operating. Thus, for example, device 2600 can be configured to periodically ascertain the availability is a preferred ∞rnmunication system. If the system is detected, then processor 2610 can be configured to load the corresponding operating instruction fiom memory 2612 and reconfigure Iransrnitter 2606 and receiver 2608 for operation in the preferred system.

For example, it may preferable for device 2600 to switch to an indoor wireless LAN if it is available. So device 2600 may be operating in a wireless WAN where no wireless LAN is available, while periodically searching for the availability of an appropriate wireless LAN. Once the wireless LAN is detected, processor 2610 will load the operating instructions, e.g., the appropriate protocol stack, for tie wireless LAN environment and will reprogram transmitter 2606 and receiver 2608 accordingly. In this manner, device 2600 can move tram one type of communication system to another, while maintaining superior performance.

It sliouldbe noted that abase station configured in accordance with tie systems and methods herein will operate in a similar manner as device 2600; however, because tie base station does not move from one type of system to another, tiere is generally no need to configure processor 2610 to reconfigure transmitter 2606 and receiver 2608 for operation in accordance with the operating instruction for a different type of system. But processor 2610 can still be configured to reconfigure, or reprogram the sub-components of transmitter 2606 and/or receiver 2608 as required by the operating conditions within the system as reported by communication devices in communication with the base station Moreover, suchabase station can be configured in accordance witi the systems and methods described herein to implement more than one mode of operation. In which case, controller 2610 canbe configuredtoreprogram1ransrritter2$ 2608 to implement the appropriate mode of operation

With reference to FIGS.26-27, additional embodimenls of tie present invention will now be described The embodiments desmT^ed below employ ultra-wideband ccm Referring to FIGS.26 and 27, one type of ultra-wideband (UWB) communication technology employs discrete pulses of electromagnetic energy that are emitted at, for example, nanosecond orpicosecond intervals (generally lens of picoseconds to hundreds of nanoseconds induration). Forthis reason, 1his type ofultra-widώandisoto Tnatis,theUWBpulsesrnaybe transmitted without modulation onto a sine wave, or a sinusoidal carrier, in contrast with conventional carrier wave communication technology. Thus, UWB generally requires neither an assigned frequency nor apower amplifier.

Inαm1iasf,theembc&lt;fimentsdesc^^ 1-25 descnT^eammumcation using a sinusoidal signal, for example, in connection with HG. IB, a very large bandwidth signal occupies 100 MHz, with sub- cliannelsof500KHz.

Anotier example of sinusoidal carrier wave communication technology is illustrated inHG.26. IKKH 802.1 Ia is a wireless local area network (LAN) protocol, which transmits a sinusoidal radio frequency signal at a 5 GHz center frequency, with aradio frequency spread of about 5 MHz. As defined herein, a carrier wave is an electromagnetic wave of a specified frequency and amplitude that is emitted by a radio tønsmitterinoiderto <img file="WO2006041947A2_D0011.tif" /> The 802.11 protocol is an example of a carrier wave communication technology. The carrier wave comprises a substantially continuous sinusoidal waveform having a specific narrow radio frequency (5 MHz) tot has a duration that may range from seconds to minutes.

In contrast, an ultra-wideband (UWB) signal may have a 2.0 GHz center frequency, with a frequency spread of approximately4 GHz, as shown inPIG.27, which illustrates two typical UWB pulses, or signals. HG.27 illustrates that the shorter the UWB signal in time, the broader the spread of its frequency spectrum. This is because bandwidth is inversely proportional to tie time duration of tie signal A 600-picosecond UWB pulse can have about a 1.8 GHz center frequency, with a frequency spread of approximately 1.6 GHz and a 300-picosecond UWB pulse can have about a 3 GHz center frequency, with a frequency spread of approximately 3.2 GHz. Thus, UWB signals generally do not operate wiihin a specific frequency, as shown in HG.26. Ether of the pulses shown in HG.27 may be frequency shifted, for example, by using heterodyning, to have essentially Hie same bandwidth but centered at any desired frequency. And because UWB signals are spread across an extremely wide frequency range, UWB communication systems allow α5mmunicationsatveryhighdatarates,suchas 100 megabitsper second or greater.

Also, because the UWB signals are spread across an extremely wide frequencyrange, thepower sampled in, for example, a one megahertz bandwidth, is very low. For example, UWB pulses of one nano-second duration and one milliwatt average power (0 dBm) spreads tie power over tie entire one gigahertz frequency band occupiedby the signal. The resulting power density is thus 1 milliwatt divided by the 1,000 MHz pulse bandwidth, or 0.001 milliwatt per megahertz (-30 dBm/MHz).

Generally, in Hie case of wireless ∞mmurications, a multiplicity of UWB pulses may be transmitted at relatively low power density (milliwatts per megahertz). However, an alternative UWB communication system may transmit at ahigher power density. For example, UWB pulses maybe Iransmittedbefween 30 dBm to -50 dBm

Several differentmethoόi of ultø-widd^and (UWB) commum communications in the United States, all of these methods must meet tie constraints recently established by the Federal Communications Commission (FCQ in their Report and Order issued April 22, 2002 (ET Docket 98-153). Currently, the FCC is allowing limited UWB communications, but as UWB systems are deployed, and additional experience with Ibisnewtechnologyis gained, tie FC(J may expand the use ofUWB communicationtechnobgy. It willbe appreciated that thepresent inventionmaybe appliedto current forms ofUWB communications, as well as to future variations and/or varieties ofUWB αmmunication technology.

For example, the April 22 Report and Order requires that UWB pulses, or signals occupy greater than 20% factional bandwidth or 500 megahertz, whichever is smaller. Fractional bandwidth is defined as 2 times the difference between the high and low 10 dB cutoff frequencies divided by the sum of Hie high and low 10 dB cutoff frequencies. However, these requirements tor wireless UWB communications intheUnited States may change inthe future.

Communication standards committees associated with flie International Institute of Electrical and Electronics Engineers QEEE) aie considering a number of ultra-wideband (UWB) wireless αmmunication methods that meet Hie current constraints established by the FCC. One UWB comriiunic^onmeihodmaytransmitUWB signals that occupy 500 MHz bands within the 7.5 GHz FCC allocation (from 3.1 GHz to 10.6 GHz). In one embodiment of this communication method, UWB signals have about a 2-nanosecond duration, which corresponds to about a 500 MHz bandwidth Ηiecmterfiεquerκ&gt;y of 1he UWB signals c^ GHz allocation. In another embodiment of this communication method, an Inverse Fast Fourier Transform 0FFT) is performed on parallel data to produce 122 carriers, each ^proximately 4.125 MHz wide. In this embodiment, also known as Orthogonal Frequency Division Multiplexing (OFDM), the resultant UWB pulse, or signal is approximately 506 MHz wide, and has a 242 nanosecond duration It meets the FCC rules for UWB communications because it is an aggregation of many relativelynarrowband carriers rather thanbecause of the duration of eachpulse.

Another UWB communication method being evaluated by flie IKKK standards αmmittees comprises tiansmitting discrete UWB signals that occupy greater than 500 MHz of frequency spectrum. For example, in one embodiment of this communication method, UWB pulse durations may vary from 2 nanoseconds, which occupies about 500 MHz, to about 133 picoseconds, which occupies about 7.5 GHz ofbandwidth. That is, a single UWB signal rnaycrøipysubstantMyaUofihe entire aUocΛ GHzto 10.6GHz).

Yet another UWB communication method being evaluated by the IKKF, standards committees comprises transmitting a sequence of signals that maybe approxirnatelyθ.7 nanoseconds or less in duration, and atach^mg rate of approximately 1.4 gigapulses per second The signals are modulatedusing aDirect^equen∞ modulation technique, and iscaUedDS-UWB. Operationhtwobanώis∞ntemplated,wifa^ signal, while the second band is centered near 8 GHz, witha2.8 GHz wide UWB signal Operation may occur at either or bofti of the UWB bands. Data rates between about 28 Megabits/second to as much as 1320 Megabits/second are contemplated

Thus, described above are 1hree different methods of wireless ultra-wideband (UWB) αmmumcation. It will be appreciated that the present inventionmaybe employed using anyone of Hie above-describedmethods, variants ofthe above methods, or other UWB communicationmethods yet to be developed

Certain features of Hie present invention may be employed by an ultra-wideband communication system. For example, an ultra-wideband communication channel, which may occupy up to several gigahertz of radio frequency spectrum may be segmented into apluiali1yofnon-overlappingsub&lt;ihannek Abase sMonnmy be usedtoal plurality of sub-channels. Each of fhese sub-channels may occupy up to hundreds of megahertz of radio frequency spectrum A single serial message may then be divided into a plurality of parallel messages. Each of the plurality of parallel messages may then be encoded onto at least some of the plurality of sub-channels. Then, the plurality of parallel messages may be transmitted over the ultra-wideband communication channel. As discussed above, the transmission may comprise transmitting a multiplicity of discrete electromagnetic pulses, with each signal having a duration that may range from tens ofpicoseconds to hundreds ofnanoseconds, ormicroseconds.

Anoiher embodiment ultra-wideband communication system may operate similar to the one described above, however a plurality of serial messages maybe divided into a plurality of parallel messages, rather than dividing one serial message into apluralityofparallelmessages. Each offtiepluralilyofparaUelmessagesmaythenbeencxxledcaitoatleast some oflheplurality of sub-channels, and then transmitted as described above.

The UWB devices and base stations in the above-described embodiments cαrmunicate with each other by transmitting and receiving a plurality of discrete electromagnetic signals, as opposed to a substantially continuous carrier wave. Eachpulse, or signal may have a duration that can range between about 10 picoseconds to about 1 microsecond, andapowerthatmayrange between about +30 dBmto about -60 dBm, as measured at a single frequency.

The present inventionrnay be employed in any type of network, be it wireless, wire, or amix of wire media and wireless components. That is, a network may use both wire media, such as coaxial cable, and wireless devices, such as satellites, or cellular antennas. As defined herein, a network is a group of points or nodes connected by communication paths. The communication paths may use wires or they may be wireless. Anetworkasdefinedherebcaninterconnect with other networks and contain sub-networks. A network as defined herein can be characterized in terms of a spatial distance, for example, such as alocal area network (LAN), apersonal area network (PAN), ametropolitan area network (MAN), a wide area network (WAN), and a wireless personal area network (WPAN), among others. A network as defined herein can also be characterized by the type of data transmission technology used by the network, such as, for example, a Transinission Control Protocol/Internet Protocol (TCP/IP) network, a Systems Network Architecture network, among others. A network as defined herein can also be characterized by whether it carries voice, data, or botfi kinds of signals. A network as defined herein may also be characterized by users of the network, such as, for example, users of a public switched telephone network (PSTN) or other type of public network, and private networks (such as within a single room or home), among others. Anetworicasdefinedhereincanalsobecriarac^ ofits connections, for example, acM-up network, aswitchednetwork, adedicatednetwork, andanon-switchednetwork, among others. A network as defined herein can also be characterized by the types of physical links that it employs, for example, optical fiber, coaxial cable, amkofbofh, unshielded twistedpair, and shielded twisted pair, among others.

The present inventionmay be employed in any type of wireless network, such as a wireless PAN, LAN, MAN, or WAN. Ih addition, the present invention may be employed in wire media, as the present invention dramatically increases the bandwidth of conventional networks that employ wire media, such as hybrid fiber-coax cable networks, or CATV networks, yet it can be inexpensively deployed without extensive modification to tie existing wire media network.

Thus, it is seen that systems and methods of ultra-wideband communications are provided. One skilled in the art wffl arpedate tot Ihe present invention can be μ^ctic^ presented in this description for purposes of illustration and not of limitation. The specification and drawings are not intended to limit the exclusionary scope of this patent document It is noted that various equivalents for the particular embodiments discussed in this description may practice the invention as welL That is, while the present invention has been described in conjunction with specific embodiments, it is evident that many alternatives, modifications, permutations and variations will become apparent to tiose of ordinary skill in the art in light of the foregoing description. Accordingly, it is intended that the present invention embrace all such alternatives, modifications and variations as fill within the scope of the appended claims. The fact 1hat a product, process or method exhibits differences from one or more of tie above-described exemplary embodiments does not mean to tie product or process is outside the scope (literal scope and/or other legaUy-recognized scope) of the following claims.