Transmission system for OFDM-signals with optimized synchronization

0001More than one reissue application has been filed for the reissue of U.S. Pat. No. 6,731,594. The reissue applications are application Ser. Nos. 11/416,477 (this application is an application for reissue of U.S. Pat. No. 6,731,594), 12/621,544 (this application is a reissue divisional of 11/416,477 filed May 3, 2006 now U.S. Pat. No. Re. 43,305), 12/621,543 (this application is a reissue divisional of 11/416,477), and CONT, this present application, which is a continuation reissue of 12/621,543 filed Nov. 19, 2009 is now U.S. Pat. No. Re. 43,829).

0002The present invention relates to a transmission method according to the preamble of claim <b>1</b>, to a transmission apparatus according to the preamble of claim <b>5</b>, a receiving method according to claim <b>9</b>, a receiving apparatus according to claim <b>13</b> and a transmission system according to claim <b>17</b>.

0003In a conventional OFDM-system signals or information contained in signals are modulated onto subcarriers in the frequency domain. The spacing between the subcarriers is equal and the subcarriers are arranged orthogonally in the frequency domain. The respectively applied modulation scheme varies for example the magnitude and phase of the described subcarriers. A conventional transmission apparatus for transmitting OFDM-signals therefore comprises as basic elements modulation means for modulating said signal onto a plurality of subcarriers using a OFDM-modulation method, transformation means for transforming said modulated signals into the time domain, and transmission means for transmitting said signals. In a conventional OFDM-system, a transmission means for OFDM-signals extends a time domain signal after a transformation into the time domain (e. g. by an inverse discrete Fourier transformation) by some guard samples to overcome multipath effects during the transmission. Usually the extension of the time domain signal is done by a cyclic extension, wherein a part of the wave form is repeated. A corresponding OFDM-signal receiving apparatus can perform correlation utilizing the two identical wave form parts to obtain information on the timing of the OFDM-time bursts for further processing. Usually this timing information is used to optimally place the discrete Fourier transformation window in the receiving apparatus to be able to transform the modulated subcarriers into the frequency domain and to demodulate them there-after.

0004To provide an efficient transmission system, the guard time or cyclic extension has to be as short as possible, namely slightly larger than the longest expected transmission path duration, which can result in poor cyclic extension based correlation properties in a receiving apparatus if the cyclic extension is very short (e. g. only a few samples). In this case, in known OFDM-systems synchronization bursts are used, which contain only synchronization information. This reduces the transmission efficiency, since a special synchronization burst designed in the time domain does not contain information (in the frequency/subcarrier domain) to be transmitted.

0005The object of the present invention is therefore to provide a transmission method according to the preamble of claim <b>1</b>, a transmission apparatus according to the preamble of claim <b>5</b>, a receiving method according to claim <b>9</b>, and a receiving apparatus according to claim <b>13</b>, which provide optimized correlation possibilities.

0006This object is achieved by a transmission method according to claim <b>1</b>, a transmission apparatus according to claim <b>5</b>, a receiving method according to claim <b>9</b>, and a receiving apparatus according to claim <b>13</b>. Also, this object is achieved by a transmission system according to claim <b>17</b>.

0007The transmission method for transmitting OFDM-signals according to the present invention comprises the steps of modulating said signals onto a plurality of subcarriers using a OFDM-modulation method, transforming said modulated signals into the time domain, and transmitting said signals, characterized in that in said modulating step every M-th subcarrier is modulated with a signal, wherein M is an integer and M≧2.

0008The transmission apparatus for transmitting OFDM-signals according to the present invention comprises modulation means for modulating said signals onto a plurality of subcarriers using a OFDM-modulation method, transformation means for transforming said modulated signal into the time domain, and transmission means for transmitting said signals, characterized in that in said modulation means every M-th subcarrier is modulated, wherein M is an integer and M≧2.

0009The receiving method according to the present invention is adapted for receiving OFDM-signals comprising M identical or respectively mirrored wave forms within one OFDM-timeburst, wherein M is an integer and M≧2, and comprises the steps of receiving said OFDM-signals, correlating said wave forms to obtain time synchronization, transforming said signals into the frequency domain, and demodulating said signals.

0010The receiving apparatus according to the present invention is adapted for receiving OFDM-signal comprising M identical or respectively mirrored wave forms within one OFDM-timeburst, wherein M is an integer and M≧2, and comprises receiving means for receiving said OFDM-signals, correlation means for correlating said wave forms to obtain time synchronization, transformation means for transforming said signals into the frequency domain, and demodulation means for demodulating said signals.

0011Advantageous features of the present invention are defined in the respective subclaims.

0012The modulation of every M-th subcarrier according to the present invention, after the succeeding transformation of the signals into the time domain, e. g. by an inverse discrete Fourier transformation, results in a signal containing M identical or respectively mirrored wave forms, whereby the total duration of the OFDM-timeburst is still 1/f₀(f₀is the subcarrier spacing). With M identical wave forms within one OFDM-timeburst, the corresponding receiving apparatus can perform an optimized correlation in the time domain, e. g. to obtain time and frequency information and synchronization, respectively. Further on, information to be transmitted can be modulated onto every M-th subcarrier and the transmission of a special time-domain synchronization time burst usually not containing useful information in the frequency-subcarrier domain is not necessary.

0013The present invention can be applied to every transmission system based on a multicarrier OFDM-modulation method, e. g. wireless and wired transmission systems. Possible and advantageous applications of the present invention in a wireless transmission system are for example the RACH (Random Access Channel), the BCCH (Broadcast Control Channel), and the IACH (Initial Acquisition Channel). Generally, the present invention is especially effective in scenarios where conventional algorithms to improve correlation based time synchronization, e. g. averaging over multiple time bursts is not possible. The present invention can be applied to any OFDM-system, particularly, when a robust time synchronization for further signal processing, e. g. discrete Fourier transformation, is required.

0014Advantageously, in said modulation means the not modulated subcarriers are set to zero. Further advantageously, only subcarriers with even indices are modulated. If only subcarriers with even indices are modulated (e. g. M=2), a full (complex) time domain signal consisting of two equal wave forms is obtained after the transformation into the time domain (e. g. by an inverse discrete Fourier transformation). If, on the other hand, only subcarriers with odd indices are modulated (e. g. M=2), a full (complex) time domain signal after the transformation into the time domain is obtained, which contains two respectively mirrored wave forms. In this case, the two wave forms are mirrored so that the correlation result is negative and an additional absolute value unit (or inverter) is necessary in the receiving apparatus to achieve a positive correlation result and a correct frequency offset.

0015Advantageously, said modulation means comprises means for generating integer values from <b>0</b> to L−1, wherein L is the number of available subcarriers, whereby said modulation means modulates every M-th signal onto said subcarriers on the basis of said integer values.

0016Advantageously, in the correlation means of the receiving apparatus according to the present invention, the identical or respectively mirrored wave forms are correlated on the basis of a delay value L<b>1</b>=S/M and averaged over L<b>2</b>≦S/M samples, whereby S is the total number of samples in one OFDM-timeburst.

0017It is further advantageous in the receiving apparatus according to the present invention to provide a peak detection means after said correlation means for providing time synchronization for the transformation of said signals into the frequency domain. It is further advantageous to provide a frequency offset detection means after said correlation means for providing frequency synchronization for the transformation of the signals into the frequency domain.

0018The transmission system for transmitting OFDM-signals according to the present invention comprises a transmission apparatus according to the present invention and a receiving apparatus according to the present invention. This transmission system can be based on a wireless or wired transmission of signals.

0019The present invention is explained in detail by means of preferred embodiments relating to the enclosed drawings, in which

0020<figref idref="DRAWINGS">FIG. 1</figref> shows an-embodiment of the transmission apparatus according to the present invention,

0021<figref idref="DRAWINGS">FIG. 2</figref> shows the modulation unit of the transmission apparatus shown in <figref idref="DRAWINGS">FIG. 1</figref> in more detail,

0022<figref idref="DRAWINGS">FIG. 3</figref> shows an example of modulating every 4-th subcarrier with a signal in the frequency domain,

0023<figref idref="DRAWINGS">FIG. 4</figref> shows an example for a signal comprising four identical wave forms in the time domain,

0024<figref idref="DRAWINGS">FIG. 5</figref> shows an embodiment of a receiving apparatus according to the present invention,

0025<figref idref="DRAWINGS">FIG. 6</figref> shows the time/frequency synchronization means of the receiving apparatus shown in <figref idref="DRAWINGS">FIG. 5</figref> in more detail and in a general form,

0026<figref idref="DRAWINGS">FIG. 7</figref> shows the time/frequency synchronization means of the receiving apparatus shown in <figref idref="DRAWINGS">FIG. 5</figref> for a signal comprising two identical wave forms,

0027<figref idref="DRAWINGS">FIG. 8</figref> shows the time/frequency synchronization means of the receiving apparatus shown in <figref idref="DRAWINGS">FIG. 5</figref> for a signal comprising four identical wave forms,

0028<figref idref="DRAWINGS">FIG. 9</figref> shows the frequency offset detection means in more detail,

0029<figref idref="DRAWINGS">FIG. 10</figref> shows a frequency spectrum for a conventional correlation performed on the basis of a cyclic extension of a time burst, and

0030<figref idref="DRAWINGS">FIG. 11</figref> shows a frequency spectrum for a correlation according to the present invention for a random access channel.

0031<figref idref="DRAWINGS">FIG. 1</figref> shows an embodiment of a transmission apparatus according to the present invention. In the transmission apparatus shown in <figref idref="DRAWINGS">FIG. 1</figref>, data <b>1</b> are channel coded in a channel coding means <b>2</b> and interleaved in an interleaving means <b>3</b>. In a modulation unit <b>4</b>, the signals carrying the data to be transmitted are modulated with an OFDM-modulation method. An OFDM-system is a multicarrier system with a plurality of subcarriers. In the modulation unit <b>4</b>, the signals carrying the information to be transmitted are modulated on every M-th subcarrier, wherein M is an integer and M≧2. The modulated signals, for example APM-signals, amplitude-phase-modulated signals, are transformed into the time domain in an inverse discrete Fourier transformation means <b>5</b>. After the transformation into the time domain, the transformed signals are provided with a cyclic extension in a cyclic extension means <b>6</b>a and then shaped in a burst shaping means <b>6</b>b. In the cyclic extension means <b>6</b>a, the OFDM-time bursts are provided with a guard time (=cyclic extension of the signal) to mitigate multipath effects during transmission. This cyclic extension serves also to provide correlation (to achieve time and frequency synchronisation) in a corresponding receiving apparatus. The cyclic extension consists in a part of the signal being added to the end of the signal, so that the receiving apparatus can conduct calculations on the basis of the doubled signal parts to provide correlation. The burst shaping means <b>6</b>b does not have to be provided in the transmission apparatus according to the present invention, since the described correlation method (to achieve time and frequency synchronisation) is based on the cyclic extension only. The provision of the burst shaping means <b>6</b>b, however, improves the transmission spectrum (reduced out of band spurious emission).

0032After the burst shaping means <b>6</b>b, or, if the burst shaping means <b>6</b>b is not provided, after the cyclic extension means <b>6</b>a, the signals are digital/analog-converted in a D/A-converter <b>7</b> and then RF-upconverted in a RF-upconversion means <b>8</b> to be transmitted by an antenna <b>9</b>.

0033In <figref idref="DRAWINGS">FIG. 2</figref>, the modulation means <b>4</b> of the transmission apparatus shown in <figref idref="DRAWINGS">FIG. 1</figref> is shown in more detail. The transmission means <b>4</b> comprises a subcarrier number generator <b>10</b> for generating integer values 0,1 . . . L−1 corresponding to the available subcarrier number L in one frequency slot in the OFDM-system. The integer values generated by the subcarrier number generator <b>10</b> are fed to a modulation unit <b>17</b>. Also, the integer values generated by the subcarrier number generator <b>10</b> are fed to a modulo means <b>11</b>, which generates series of integer values depending on the chosen modulation step of the modulation means <b>4</b>. If, for example, every 4-th subcarrier is modulated with a signal, so that M=4, the modulo means <b>11</b> outputs integer values 0,1,2,3,0,1,2,3,0,1,2,3, . . . .

0034The output of the modulo means <b>11</b> is fed to a compare means <b>12</b>, which compares the integer values provided by the modulo means <b>11</b> with integer values generated by a compare value generator <b>13</b>. The compare means <b>12</b> gives an “active” signal to a switch means <b>14</b>, if the inputs from the modulo means <b>11</b> and the compare value generator <b>13</b> are equal. If, for example in the above example, the compare value generator <b>13</b> generates an integer value “1”, the compare means <b>12</b> outputs an “active” signal every 4-th time an integer value “1” is fed from the modulo means <b>11</b> (M=4). Otherwise, the output of the compare means <b>12</b> is a “not active” signal. If the switch means <b>14</b> obtains an “active” signal from the compare means <b>12</b>, it connects a line <b>16</b> providing signals with data to be modulated with the modulation unit <b>17</b>. If the switch means <b>14</b> obtains an “not active” signal from the compare means <b>12</b>, it connects a zero terminal <b>15</b> with the modulation unit <b>17</b>. In the above example (M=4), the switch means <b>14</b> therefore connects the data line <b>16</b> every 4-th time an integer value is generated by the subcarrier number generator <b>10</b> with the modulation unit <b>17</b>. Therefore, every 4-th subcarrier is modulated with signals carrying data in the modulation unit <b>17</b>. The other subcarriers are not modulated in the modulation unit <b>17</b>, since the switch means <b>14</b> selects the zero terminal <b>15</b> at the time these subcarriers are fed to the modulation unit <b>17</b>. At the zero terminal <b>15</b>, a “0” value is input (complex: 0=0+j×0) so that the other subcarriers are not modulated.

0035In <figref idref="DRAWINGS">FIG. 3</figref>, a frequency domain representation for the modulation of every 4-th subcarrier is shown. The horizontal axis shows the number S=32 of the inverse discrete Fourier transformation samples <b>0</b> . . . <b>31</b> and the vertical axis shows the magnitude of the subcarriers. Also, one frequency slot comprising L=24 (<b>0</b> . . . <b>23</b>) available subcarriers is shown wherein each subcarrier is sampled in the inverse discrete Fourier transformation means <b>5</b>. Each 4-th subcarrier <b>18</b> (subcarrier number 0, 4, 8, 12, 16 and 20) is modulated with a signal, wherein the spacing between adjacent subcarriers is f₀. The IDFT samples <b>0</b> . . . <b>3</b> and <b>28</b> . . . <b>31</b> are unmodulated guard subcarriers (to perform a power-of-2 DFT, here 32-point DFT), and the samples <b>4</b> . . . <b>27</b> are the used subcarrier samples (here we assumed one frequency slot consists of 24 subcarriers).

0036<figref idref="DRAWINGS">FIG. 4</figref> shows the corresponding time domain wave forms for the example shown in <figref idref="DRAWINGS">FIG. 3</figref>, wherein every 4-th subcarrier is modulated. The modulation of every 4-th subcarrier leads to time domain signals containing 4 identical wave forms, since only subcarriers with even indices (compare <figref idref="DRAWINGS">FIG. 3</figref>) have been modulated.

0037In <figref idref="DRAWINGS">FIG. 4A</figref>, the IN-part (in-phase part), and in <figref idref="DRAWINGS">FIG. 4B</figref>, the QUAD-part (quadrature part) of a wave form signal in the time domain, in which every 4-th subcarrier has been modulated in the frequency domain, is shown. <figref idref="DRAWINGS">FIG. 4C</figref> shows the envelope of the IN-part and the QUAD-part shown in <figref idref="DRAWINGS">FIG. 4A</figref> and <figref idref="DRAWINGS">FIG. 4B</figref>, respectively (envelope=SQRT {IN*IN+QUAD*QUAD}). As can be seen, the wave form signals contain 4 identical wave forms, since in the frequency domain only subcarriers with even indices have been modulated. The modulation of subcarriers with only odd indices leads to wave forms which are slightly different to the wave forms shown in <figref idref="DRAWINGS">FIG. 4</figref>. The modulation of subcarriers with only odd indices leads after the transformation in the time domain to wave form signals with respectively mirrored wave forms. In this case, every second wave form in the time domain signal is mirrored in respect to the correspondingly preceding wave form. If a sample in a first waveform is x₁=a+j×b, the corresponding sample in the second waveform is x₂=(−a−j×b)=(−<b>1</b>)*(a+j×b).

0038In <figref idref="DRAWINGS">FIG. 5</figref>, an embodiment of a receiving apparatus according to the present invention is shown. Data transmitted, e. g. from a transmission apparatus as shown in <figref idref="DRAWINGS">FIG. 1</figref>, are received in an antenna <b>19</b> and RF-downconverted in a RF-downconversion means <b>20</b>. Then, the signals are analog to digital converted in an A/D-converter <b>21</b> and fed to a time/frequency synchronization means <b>22</b>. In the time/frequency synchronization means <b>22</b>, the received signals are correlated and synchronized, so that a proper transformation to the frequency domain in a succeeding discrete Fourier transformation means <b>23</b> can be executed. The transformed signals are then demodulated in a demodulation means <b>24</b>. The demodulated signals are de-interleaved in de-interleaving means <b>25</b> and then channel-decoded in a channel-decoding means <b>26</b>. The channel-decoding means <b>26</b> outputs data signals <b>27</b> to be further processed.

0039In <figref idref="DRAWINGS">FIG. 6</figref>, the time/frequency synchronization means <b>22</b> of the receiving apparatus shown in <figref idref="DRAWINGS">FIG. 5</figref> is shown having a general structure. The time-/frequency synchronization means <b>22</b> consists generally of a correlation means with one or more correlator parts <b>28</b>, <b>29</b>, <b>30</b>, <b>31</b> and a moving average means <b>40</b>. After the moving average means <b>40</b>, an absolute value means <b>45</b> is provided. After the absolute value means <b>45</b>, a peak detection means <b>46</b> can be provided. The output of the peak detection means <b>46</b> and the output of the moving average means <b>40</b> can be fed to an also optionally provided frequency offset detection means <b>47</b>.

0040The time/frequency synchronization means <b>22</b> comprises (M−1) correlator parts. If, for example, every 4-th subcarrier is modulated, the time-/frequency synchronization means <b>22</b> comprises 3 correlator parts, as is shown in more detail in <figref idref="DRAWINGS">FIG. 8</figref>.

0041In <figref idref="DRAWINGS">FIG. 6</figref>, the output of the A/D-converter <b>21</b> is fed to a first correlator part <b>28</b> comprising a delay means <b>32</b> and a multiplier <b>35</b>. The output of the A/D-converter is fed to the delay means <b>32</b>, which delays the signal with a factor z^−L1. The output of the delay means and the output of the A/D-converter <b>21</b> are multiplied in the multiplier <b>35</b>. The output of the delay means <b>32</b> is further fed to a delay means <b>33</b> and a multiplier <b>36</b> of a second correlator part <b>29</b>. The delay means <b>33</b> delays the output of the delay means <b>32</b> with a factor z^−L1. The output of the delay means <b>33</b> is multiplied in the multiplier <b>36</b> with the output of the delay means <b>32</b>. The outputs of the multiplier <b>35</b> and the multiplier <b>36</b> are added in an adder <b>38</b>. Successive correlator parts and adders are symbolized by a block <b>30</b>. The (M−1)th correlator part <b>31</b> delays the output of the delay means of the preceding correlator part in a delay means <b>34</b> by a factor z^−L1and multiplies the output of the delay means <b>34</b>. The output of the delay means <b>34</b> is multiplied in a multiplier <b>37</b> with the output of the preceding delay means. The output of the multiplier <b>37</b> is added in an adder <b>39</b> to the output of a preceding adder.

0042Then, the output of the last adder <b>39</b> is fed to the moving average means <b>40</b>. In the moving average means <b>49</b>, the incoming signal is delayed in a delay means <b>41</b> by a factor z^−L2. In an adder <b>42</b>, the output of the delay means <b>41</b> is subtracted from the incoming signal. The output of the adder <b>42</b> is fed to an adder <b>43</b>, which is backfed with its own output delayed by factor z⁻¹in a delay means <b>44</b>. The moving average means thus performs the function

0043<maths id="MATH-US-00001" num="00001"><math overflow="scroll"><mrow><mrow><mrow><mrow><mo>(</mo><mrow><mn>1</mn><mo>-</mo><msup><mi>z</mi><mrow><mo>-</mo><mi>L2</mi></mrow></msup></mrow><mo>)</mo></mrow><mo>/</mo><mrow><mo>(</mo><mrow><mn>1</mn><mo>-</mo><msup><mi>z</mi><mrow><mo>-</mo><mn>1</mn></mrow></msup></mrow><mo>)</mo></mrow></mrow><mo>=</mo><mrow><munder><mover><mo>∑</mo><mi>L2</mi></mover><mrow><mi>m</mi><mo>=</mo><mn>0</mn></mrow></munder><mo>⁢</mo><msup><mi>z</mi><mrow><mo>-</mo><mi>m</mi></mrow></msup></mrow></mrow><mo>,</mo></mrow></math></maths><img file="USRE45293E_D0001.tif" /><br /> which means y(m)=x(m)+x(m−1)+. . . +x(m−L2) if the input signal of the MAV means <b>40</b> is defined as x(m) and its output signal is defined as y(m).

0044In the example of <figref idref="DRAWINGS">FIG. 6</figref> and also the examples of <figref idref="DRAWINGS">FIGS. 7 and 8</figref>, the correlation delay value L<b>1</b> is L<b>1</b>=S/M. The moving average value L<b>2</b> is L<b>2</b>≦S/M, so that a signal fed to the moving average means <b>40</b> is delayed over L<b>2</b>≦S/M samples. In both cases, S is the total number of samples in one OFDM-timeburst. In the example shown in <figref idref="DRAWINGS">FIGS. 3</figref>, S=32 and M=4, so that L<b>1</b>=8 and L<b>2</b>≦8. The best performance is achieved if L<b>2</b> is close to S/M, in the example of <figref idref="DRAWINGS">FIG. 3</figref> this means L<b>2</b> should be close to 8 samples (e. g. 6, 7 or 8 samples).

0045In the correlation means <b>28</b>, <b>29</b>, <b>30</b>, <b>31</b> and the moving average means <b>40</b>, correlation in the time domain to obtain time synchronization information for further processing of the incoming signals is performed. The output of the moving average means <b>40</b> is then fed to an absolute value means <b>45</b>. The output of the absolute value means <b>45</b> is fed to a peak detection means <b>46</b>, which identifies the best correlation result for an optimum estimate of the window position of the discrete Fourier transformation in the discrete Fourier transformation means <b>23</b>. In an ideal transmission case, the imaginary part of the correlated signal is zero. In the case of a frequency offset in the transmitted signal, the imaginary part of the correlated signal is not zero, so that a frequency offset detection has to be performed in a frequency offset detection means <b>47</b>. Conventionally, if all subcarriers are modulated, the frequency offset detection range is limited to −f₀/2 . . . +f₀/2, whereby f₀is the subcarrier spacing. According to the present invention, the frequency offset detection range in the frequency offset detection means <b>47</b> is extended to M×(−f₀/2) . . . M×(+f₀/2), wherein f₀is the subcarrier spacing. Therefore, the frequency offset detection range is advantageously extended according to the present invention. The output of the frequency offset detection means <b>47</b> and the peak detection means <b>46</b> are used for time-/frequency synchronization in the succeeding discrete Fourier transformation means <b>23</b>.

0046In a case, in which only subcarriers with odd indices are modulated, an additional absolute block means (or sign inverter) can be used in the receiving apparatus to achieve a positive correlation result. This additional absolute block means can, for example, be provided between the last correlation part and the moving average means <b>40</b>. In order to achieve time synchronisation only this block is not necessary, because the absolute value means <b>45</b> in <figref idref="DRAWINGS">FIG. 5</figref> already provides positive results. However, to achieve a correct frequency detection (synchronisation), this additional absolute block means is required.

0047In <figref idref="DRAWINGS">FIG. 7</figref>, a time-/frequency synchronization means <b>22</b> is shown for M=2. In this case, the correlations means consists only of one correlator part <b>28</b>. The correlation delay value L<b>2</b> is S/<b>2</b> and the moving average parameter L<b>2</b> is smaller or equal S/<b>2</b>, whereby the best performance is achieved if L<b>2</b> is close to S/<b>2</b>.

0048In <figref idref="DRAWINGS">FIG. 8</figref>, a time-/frequency synchronization means <b>22</b> is shown for M=4. In this case, L<b>1</b>=S/<b>4</b> and L<b>2</b>≦S/<b>4</b>.

0049In <figref idref="DRAWINGS">FIG. 9</figref>, the frequency offset detection means <b>47</b> shown in <figref idref="DRAWINGS">FIGS. 6</figref>, <b>7</b> and <b>8</b> is shown in more detail. As stated above, the frequency offset detection range is advantageously extended according to the present invention. The structure of the frequency offset detection means <b>47</b> shown in <figref idref="DRAWINGS">FIG. 9</figref> provides this extended frequency offset detection range.

0050The frequency offset is Δf=M×f₀×(½π)×arctan(q/i), wherein M is the number of the repeated wave forms in one OFDM time burst, f₀the subcarrier spacing, “i” the in-phase part and “q” the quadrature part of the complex output of the MAV means <b>40</b>. As shown in <figref idref="DRAWINGS">FIG. 9</figref>, the frequency offset detection means <b>47</b> comprises a split means <b>48</b>, a calculation means <b>49</b> and a multiplier <b>50</b>. In the split means <b>48</b>, the complex output of the MAV means <b>40</b> is separated in an “in” and a “quad” component, when the split means <b>48</b> receives a peak detection signal from the peak detection means <b>46</b>. The peak detection means produces a peak detection signal every time it detects a peak. The “in” and “quad” component from the split means <b>48</b> are then fed to the calculation means <b>49</b>. The calculation means <b>49</b> calculates the mathematical expression of (½π)×arctan(q/i), which can be done in a look-up table (hardware implementation) or calculated in a processor. The calculation result from the calculation means <b>49</b> is supplied to the multiplier <b>50</b>. The multiplier <b>50</b> multiplies the calculation result from the calculating means <b>49</b> with M(number of repeated wave forms in one OFDM time burst). The result of the multiplication in the multiplier <b>50</b> is the frequency offset Δf as a fraction of the subcarrier spacing f₀(result=Δf/f₀). The detected frequency offset is used in the synchronisation unit <b>22</b> of the receiving apparatus to obtain the frequency synchronisation.

0051In <figref idref="DRAWINGS">FIG. 10</figref>, a frequency spectrum of a conventionally correlated signal (cyclic extension) is shown and compared with a frequency spectrum shown in <figref idref="DRAWINGS">FIG. 11</figref> for a signal correlated according to the present invention. The parameter for the example shown in <figref idref="DRAWINGS">FIG. 10</figref> has been calculated for a RACH-burst. Its parameters are: signal to noise ratio: 6,0 dB, frequency offset: −0,30001×f₀, guard samples per burst: <b>16</b>, RACH-scheme: <b>4</b>, number of RACH-slots: <b>4</b>, discrete Fourier transformation size (=number of subcarriers or number of OFDM-burst samples): 128 and used subcarriers per slot: <b>96</b>.

0052As can be seen, the present invention provides for very good peak detection compared to the conventional correlation. The four bursts in the signal stream can be clearly identified. The detected frequency offset values are: 0,3004; 0,3081, 0,3117 and 0,3151 which is very accurate (error&lt;5%).