US7120217B2

Phase-locked loop circuit

Summary by NHIP

Dual-Channel PLL Circuit

The phase-locked loop circuit separates synchronization pulses into useful and disturbance signal components using two distinct channels. A first loop filter arrangement includes a capacitor dimensioned to produce a control voltage dependent only on the useful signal components from lagging phase deviations.

Claim Score by NHIP

Read claim 1, the broadest

Abstract

In a PLL circuit including a voltage-controlled oscillator, a phase detector and a final control element, the final control element contains two separate channels, between the phase detector and the voltage controlled oscillator, wherein one channel processes the useful signal components and the other channel processes the disturbance signal components of the synchronization pulses. Each channel has two tracks, for generation of a potential difference, wherein each track is connected to a capacitor plate.

US7120217B2, drawing sheet 1 — Sheet 1 of 5

Term

Term ended

Expired 31 July 2023, 3.2 years ago.

24 claims: 2 independent, 22 dependent

1
Broadest claimClaim Score 14, narrow(NHIP)A phase-locked loop circuit comprising:a voltage-controlled oscillator that has plural voltage inputs including at least one first voltage input and at least one second voltage input, and an oscillation output;a phase detector that has an actual frequency input and plural control outputs including first and second control outputs, and that is adapted to detect a phase deviation of an actual frequency received at said actual frequency input relative to a reference frequency, and to produce first synchronization pulses having useful signal components and disturbance signal components at said first control output when said phase deviation is a lagging phase deviation, and to produce second synchronization pulses having useful signal components and disturbance signal components at said second control output when said phase deviation is a leading phase deviation;a feedback path connecting said oscillation output of said voltage-controlled oscillator to said actual frequency input of said phase detector;and a control arrangement interposed and connected between said control outputs of said phase detector and said voltage inputs of said voltage-controlled oscillator;wherein: said control arrangement includes a first channel and a second channel that connect said control outputs to said voltage inputs, said first channel includes respective control inputs respectively connected to said first control output and said second control output, a first charge pump arrangement, and a first loop filter arrangement including at least one first capacitor dimensioned and connected so as to produce a first control voltage signal dependent only on said useful signal components of respective ones of said first and second synchronization pulses, said first channel is connected to said at least one first voltage input of said voltage-controlled oscillator so as to provide said first control voltage signal to said at least one first voltage input, said second channel includes respective control inputs respectively connected to said first control output and said second control output, a second charge pump arrangement, and a second loop filter arrangement differing from said first loop filter arrangement and including at least one passive filter element dimensioned and connected so as to produce a second control voltage signal dependent only on said disturbance signal components of respective ones of said first and second synchronization pulses, and said second channel is connected to said at least one second voltage input of said voltage-controlled oscillator so as to provide said second control voltage signal to said at least one second voltage input.
16
A phase-looked loop circuit comprising:a voltage-controlled oscillator that has plural voltage inputs including at least one first voltage input and at least one second voltage input, and an oscillation output;a phase detector that has an actual frequency input and plural control outputs including first and second control outputs, and that is adapted to detect a phase deviation of an actual frequency received at said actual frequency input relative to a reference frequency, and to produce first synchronization pulses having useful signal components and disturbance signal components at said first control output when said phase deviation is a lagging phase deviation, and to produce second synchronization pulses having useful signal components and disturbance signal components at said second control output when said phase deviation is a leading phase deviation;a feedback path connecting said oscillation output of said voltage-controlled oscillator to said actual frequency input of said phase detector;and a control arrangement interposed and connected between said control outputs of said phase detector and said voltage inputs of said voltage-controlled oscillator;wherein: said control arrangement includes a first channel and a second channel;said first channel includes a first charge pump arrangement, first channel control inputs respectively connected to said first control output and said second control output of said phase detector, and a first loop filter arrangement including at least one first capacitor connected between said first charge pump arrangement and said at least one first voltage input of said voltage-controlled oscillator;said second channel includes a second charge pump arrangement, second channel control inputs respectively connected to said first control output and said second control output of said phase detector, and a second loop filter arrangement including at least one passive filter element connected between said second charge pump arrangement and said at least one second voltage input of said voltage-controlled oscillator;said first channel has first parameters including a first pump current that characterizes said first charge pump arrangement, a first filter characteristic that characterizes said first loop filter arrangement, and a first frequency-voltage gain characteristic that characterizes said at least one first voltage input of said voltage-controlled oscillator;said second channel has second parameters including a second pump current that characterizes said second charge pump arrangement, a second filter characteristic that characterizes said second loop filter arrangement, and a second frequency-voltage gain characteristic that characterizes said at least one second voltage input of said voltage-controlled oscillator;and at least one of said second parameters differs from a counterpart one of said first parameters so that a first control voltage signal provided to said at least one first voltage input of said voltage-controlled oscillator from said first channel is dependent predominantly on said useful signal components of said first and second synchronization pulses, and a second control voltage signal provided to said at least one second voltage input of said voltage-controlled oscillator from said second channel is dependent predominantly on said disturbance signal components of said first and second synchronization pulses.

Independent claims2

35 paragraphs in 5 sections, as filed

FIELD OF THE INVENTION

0001The invention relates to a PLL circuit comprising a voltage controlled oscillator, a phase detector and a final control element.

BACKGROUND OF THE INVENTION

0002<figref idref="DRAWINGS">FIG. 1</figref> shows a conventional phase-locked loop (PLL) circuit. The conventional PLL circuit comprises a phase detector (PD) 3, a charge pump (CP) 4, an external loop filter (LF) 5, a voltage-controlled oscillator (VCO) 1 and a frequency divider (DIV) 2. The phase detector 3 detects the phase difference between the reference frequency fRef and the actual frequency first dependent on the VCO. If the phase of the actual frequency fist leads the phase of the reference frequency fRef, a falling pulse DW, the duration of which corresponds to the phase difference between the reference frequency fRef and the actual frequency first, is generated and directed to a DW output 15 of the phase detector 3. If the phase of the actual frequency fist lags the phase of the reference frequency fRef, a rising pulse UP, the duration of which corresponds to the phase difference between the reference frequency fRef and the actual frequency fist, is generated and directed to a UP output 14 of the phase detector 3. If the actual frequency fist synchronizes with the reference frequency fRef, a UP pulse as well as a DW pulse are directed simultaneously to the UP and DW outputs 14, 15 respectively.

0003Upon receipt of the UP pulse from the phase detector 3, the charge pump 4 charges the external loop filter 5. Upon receipt of the DW pulse from the phase detector 3, the charge pump 4 discharges the external loop filter 5. If the applied voltage V1 of the loop filter 5 increases upon receipt of a high voltage from the charge pump 4, the frequency fvco in the VCO 1 also rises. If, on the other hand, the applied voltage V1 in the external loop filter 5 decreases as a result of receipt of the lower voltage from the charge pump 4, the oscillation frequency fvco the VCO 1 drops. The loop filter 5 and the charge pump 4 form components of the final control element 36 for the VCO 1. The final control element 36 converts the measurement result from the chase detector into a voltage, with which the oscillation frequency of the VCO can be influenced.

0004If the actual frequency fist dependent on the oscillation frequency fvco of the VCO 1 is lower than the reference frequency fRef and the phase of the actual frequency lags the phase of the reference frequency fRef, UP pulses are generated in the phase detector 3, as a result of which the charge pump 4 increases the voltage V1 at the external loop filer 5. As a result of the increase in the voltage V1 the oscillation frequency fvco of the VCO 1 rises. The increase in the oscillation frequency fvco causes the increase in the actual frequency fist at the phase detector 3 and thus the approximation to the reference frequency fRef.

0005If, however, the actual frequency fist dependent on the oscillation frequency fvco of the VCO 1 is higher than the reference frequency fRef, i.e. if the phase of the actual frequency fist leads the phase of the reference frequency fRef, a DW pulse DW is generated in the phase detector 3, as a result of which the charge pump 4 decreases the voltage V1 at the external loop filter 5. As a result of the decrease in the voltage V1 the oscillation frequency fvco the VCO 1 also decreases. The decrease in the oscillation frequency fvco causes the drop in the actual frequency fist at the phase detector 3 and thus the approximation to the reference frequency fRef.

0006To smooth the current pulses of the charge pump 4, which are initiated because of the UP and DW pulses in the phase detector 3, a filter capacitor CG 8 is located between the charge pump 4 and the loop filter 5.

0007By virtue of the fact that the sensitivity and thus the transconductance Kvco=f(V1) of the VCO 1 is not constant over the whole voltage range, but can vary by a factor 4 over the whole voltage range, for example, an additional switch 17 is located between the input of the VCO 1 and the charge pump 4, which should at least partially compensate this effect.

0008The external loop filter 5 comprises a first capacitor CS1 9, with which the voltage V1 for the VCO 1 is generated, and an array connected in series comprising a resistor RS 28 with a second capacitor CS2 10 connected in parallel.

0009The dimensioning of such a PLL circuit for applications in the field of telecommunications is configured such that circuit-related disturbances are as small as possible. Firstly, disturbances are caused by the currents I1 and I2. However, this disturbance influence decreases in relation to the useful signal V1, the greater the current I from the charge pump 4. This, in turn, necessitates an increase in the capacitance of the capacitor CS1 9 in order to come to the identical voltage value V1.

0010Secondly, a further disturbance is caused by the modulation of the output frequency with the thermal noise of the resistor RS 28 in the VCO 1. This disturbance increases, the greater the resistance RS 28 and the greater the transconductance Kvco of the VCO. Typical values for a loop filter 5 in the field of telecommunications are I=0.1 mA−2 mA; CS1+CSs2=200 pF−5 nF; RS=100 Ω−2 kΩ.

0011However, a disadvantage here is that, because of the large capacitances required as a result of the reduction of disturbance influences, the loop filter 5 cannot be integrated into the PLL circuit because, for example, even in the case of a typically used capacitance of CS1 of 1 nF, for example, a surface area of approximately 1 mm2 is required. If the capacitances CS1 and CS2 were of smaller dimensions in the loop filter 5, then the current I would also have to be decreased and the resistance RS increased. As a result, the disturbances arising in the PLL circuit would be too great and therefore the requirements above all for the field of telecommunications could not be met.

SUMMARY OF THE INVENTION

0012The present invention seeks to provide a PLL circuit, in which the functions of a loop filter are also integrated in the PLL circuit without causing an increase in the disturbance effects.

0013According to the present invention, there is provided a PLL circuit comprising a voltage-controlled oscillator for generation of an oscillation frequency (fvco), a phase detector for measurement of the phase deviation, wherein synchronization pulses, which can have useful signal and disturbance signal components, are generated dependent on the phase deviation, and a final control element containing at least one capacitor, which, dependent on the synchronization pulses, generates a voltage which serves as a correcting variable for the voltage-controlled oscillator, wherein the final control element contains at least two separate channels between the phase detector and the voltage-controlled oscillator, and wherein at least one channel processes the useful signal components and at least one other channel processes the disturbance signal components of the synchronization pulses.

0014These different signal components can be separated from one another and processed on the basis of their different duration of effect.

0015The advantages of the invention lie in the fact that the previous opposing demands on the dimensioning of the components are dispensed with as a result of the division into a separate useful signal channel and a separate disturbance signal channel.

0016Each component of the final control element is subject to only one channel and thus to only one requirement, and therefore it can be dimensioned in an optimum manner for the respective purpose.

0017A further advantageous development of the invention is that at least one channel in the final control element has at least two tracks, with which potential differences, and no longer absolute voltage as previously usual, are generated in the final control element. In this case, each track is connected to a capacitor plate which is charged or discharged by the current sources so that potential differences are set between two tracks of a feed line. Each current source is connected to a switch, with which the current source is connected to the track or disconnected, the switches being operated in dependence on the synchronisation pulses generated in the phase detector. Currents acting in opposite directions flow because of the generation of potential differences. These currents, in turn, generate opposed disturbances, which, although identical with respect to quantity, compensate one another. As a result of this disturbance compensation, the capacitance, and thus the surface area requirement of the capacitor or the capacitors, can be reduced to such an extent that they can be integrated into a PLL circuit. As a result of such a construction, with which potential differences are generated instead of absolute voltages, disturbances in supply voltage or supply current may be balanced. The transconductance of the VCO may also be split into two independent sub-ranges by feeding several control voltages into the VCO. One sub-range processes the useful signal. This area is insensitive to disturbance influences. The transconductance K1vco of this area has no influence on the properties of the circuit. The other sub-range processes the disturbances. The transconductance K2vco of this area has great influence on the properties of the circuit. An advantageous K2vco value can be selected here for this reason. This K2vco value and the resistance value RS, which determines the signal/noise ratio, are adjusted to a desired value by advantageous scaling by changing the current of the current sources in spite of the small capacitances.

0018Another advantage of the invention is that the entire PLL circuit, with all the components thereof as disclosed herein, can be monolithically integrated, i.e. arranged in a monolithic configuration.

BRIEF DESCRIPTION OF THE DRAWINGS

Preferred embodiments of the present invention will now be described, by way of example only, with reference to the accompanying drawings, of which:

<figref idref="DRAWINGS">FIG. 1</figref> shows a PLL circuit with external loop filter according to the prior art;

<figref idref="DRAWINGS">FIG. 2</figref>a shows a PLL circuit according to a first embodiment of the present invention with fully integrated final control element;

<figref idref="DRAWINGS">FIG. 2</figref>b shows a PLL circuit according to a second embodiment of the present invention with fully integrated final control element and a common mode loop;

<figref idref="DRAWINGS">FIG. 3</figref> shows an alternative PLL circuit according to a third embodiment of the present invention with fully integrated final control element.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

0024<figref idref="DRAWINGS">FIG. 2</figref>a shows a PLL circuit with fully integrated final control element. The VCO 1 makes available the oscillation frequency fvco of the PLL circuit at its output 11. The oscillation frequency fvco is divided by the factor N in a frequency divider 2 in a feedback path from the output 11 to a frequency detector 3. This results in an oscillation with an actual frequency, which lies in the region of the reference frequency fRef and the phase difference of which can be determined by the phase detector 3. For this, the constant reference frequency fRef is fed in at one input 13 of the phase detector 3 and the actual frequency fist dependent on the VCO is fed in at the other input 12. The phase differences of the two oscillations are determined in the phase detector 3. In this case, the phase detector 3 can also be configured as a phase frequency detector, which not only compares the phase, but also the frequency of the two oscillations fRef and fist. Depending on the result of this comparison, as already described in connection with <figref idref="DRAWINGS">FIG. 1</figref>, two output signals, a UP pulse and/or a DW pulse, are formed at the two outputs 14, 15. These pulses are used to control the final control element 36, which converts the measurement result, which is present in the form of pulses, into voltages with which the VCO 1 can be controlled. With each of these UP and DW pulses several (four in each case in this illustration) switches SU1, SU2, SU3, SU4 and SD1, SD2, SD3, SD4 are operated, and thereby the current sources 6, 7, 18, 19, 20, 21, 22 and 23 are connected or disconnected. This part of the arrangement forms the charge pump 4 of the final control element 36. In this example embodiment, two double-track channels 37, 38 are formed, wherein each track 371, 372 of one channel 37 and each track 381, 382 of the other channel 38 is connected respectively to a respective capacitor plate, as a result of which a potential difference is respectively built-up between the tracks 371 and 372 or 381 and 382 in a channel 37 or 38.

0025In the case of a UP pulse the current sources 6 and 19 generate a current I by operation of the switches SU1 and SU2, in the case of a DW pulse current sources 7 and 18 generate the same current flowing in the opposite direction through CS1 by operation of the switches SD1 and SD2. In the case of a UP pulse the current sources 20 and 23 generate a current x·y·I by operation of the switches SU3 and SU4, in the case of a DW pulse current sources 21 and 22 generate the same current flowing in the opposite direction through CS2 by operation of the switches SD3 and SD4. The current sources are connected to capacitors CS1, CS2, i.e. 9, 10 via the switches in such a manner that current source 6 charges one plate of the capacitor CS1 via the UP pulse-dependent switch SU1 and current source 7 charges this same plate of the capacitor CS1 via a DW pulse-dependent switch SD1. The other plate of CS1, is connected, on the one hand, to current source 19 via switch SU2 and, on the other hand, to current source 18 via switch SD2. Each plate of CS1 9 is connected to a respective input 24, 25 of the VCO.

0026As a result of this arrangement of the current sources, a potential difference V3A−V3B is generated in the CS1 and forwarded to the VCO 1. The currents I1 and I3, which flow between the CS1 and the VCO inputs, nonetheless still cause disturbances which are equally large in quantity and counterbalance one another because of their symmetric arrangement relative to one another, and therefore the potential difference V3A−V3B of the CS1 can be fed into the VCO free from disturbance. The plates of the second capacitor CS2 10 are likewise operated with four current sources. One plate of the CS2 10 is connected to current source 20 via the UP pulse-dependent switch SU3 and to current source 21 via the DW pulse-dependent switch SD3. The other plate of the CS2 is supplied by current source 23 via switch SU4 and by current source 22 via switch SD4. A resistor RS 28 is arranged parallel to the CS2 10. The two capacitors 9, 10 and the resistor 28 form the loop filter 5 of the final control element 36. The RS 28 determines the signal/noise ratio of the array together with the transconductance K2vco. Each plate of the CS2 10 is connected to a further input 26, 27 of the VCO 1. As a result of this arrangement of the current sources 20, 21, 22, 23, a potential difference V2A−V2B is generated in the CS2 and forwarded to the VCO. The currents I2 and I4, which flow between the CS2 and the VCO inputs, nonetheless still cause disturbances which are equally large in quantity and counterbalance one another because of their symmetric arrangement relative to one another, and therefore the potential difference V2A−V2B of the CS2 can likewise be fed into the VCO free from disturbance. In the example of application, the VCO 1 is fed with two voltage values V3A−V3B, V2A−V2B and from the sum of these voltage values and the base frequency (K1vco(V3A−V3B))+(K2vco(V2A−V2B))+fvco0 generates an oscillation with the frequency fvco at its output 11. The two potential differences V3A−V3B, V2A−V2B contribute to the adjustment of the oscillation frequency to a different degree. The potential difference V3A−V3B generated in the first channel 37 of CS1 is only dependent on the useful signal which results from the phase difference between the actual and the reference frequency. Dynamic disturbances have little influence on this potential difference. The potential difference V2A−V2B generated in the second channel 38 of CS2 only differs from zero in the case of a disturbance. The second channel 38 determines the dynamic properties of the arrangement. In the example of application, different transconductances K1vco and K2vco are used at the VCO 1. The transconductance K1vco is assigned to the potential difference V3A−V3B dependent on the useful signal and the transconductance K2vco is assigned to the potential difference V2A−V2B dependent on the disturbance signal. The two transconductances will generally differ greatly from one another K1vco>>K2vco. However, they are sensitive to changes in the respective potential difference, and therefore K1vco=f(V3A−V3B) and K2vco=f(V2A−V2B) applies. However, since V2A−V2B≈0, K2vco=constant, which in turn causes constant dynamic properties because the dynamic properties are only dependent on K2vco. The factor between K1vco and K2vco is balanced by the dimensioning of the current sources 20, 21, 22 and 23 of the resistor 28 and the capacitor 10. In the example of application, the following interrelationship exists between the sensitivities K1vco, K2vco, the dimensioning of the resistor RS and the dimensioning of the current sources 20, 21, 22 and 23: K2vco=K1vco/y and RS/x=constant. Since the signal/noise ratio is proportional to K2vco and √RS, a larger resistance value RS can be compensated by reducing the transconductance value K2vco. If the potential differences amount to (V3A−V3B)=(V2A−V2B)=0V, then a base oscillation with a frequency of fvco0≠0 Hz should be available at the output of the VCO 1.

0027This PLL circuit can operate with low noise with small capacitances of CS1 and CS2 and a resistor with a large resistance value RS. As a result of the symmetric arrangement of the disturbance influences, caused by the generation of potential differences, these disturbance influences counterbalance one another, as a result of the favourable scaling x·y of the current sources 20, 21, 22 and 23 the noise can be suppressed by a low K2vco in spite of the high resistance. As a result of the feed of potential differences, this PLL circuit is also insensitive to fluctuations of supply systems. Because of the smaller components, in particular the capacitances of CS1 and CS2, such a PLL circuit can be fully housed in a single integrated circuit (IC), especially if the IC is manufactured using bipolar, CMOS, BICMOS or other IC technology. It is an advantage, particularly in the communications field, if ICs operate independently of fluctuations in supply voltages or supply currents, are small and operate free from disturbance. External loop filters are no longer necessary. The additional current sources and switches can be accommodated in a very small space.

0028<figref idref="DRAWINGS">FIG. 2</figref>b shows a PLL circuit with fully integrated final control element and a common mode loop. The VCO 1 makes available the oscillation frequency fvco, where fvco=960 MHz, of the PLL circuit at its output 11. The oscillation frequency fvco is divided in a frequency divider 2 by the factor N=64. This results in an oscillation with an actual frequency fist=15 MHz, which lies in the range of the reference frequency fRef, where fRef=14.2969 MHz, and the phase difference of which can be determined by the phase detector 3. For this, the constant reference frequency fRef, where fRef=14.2969 MHz, is fed in at one input 13 of the phase detector 3 and the actual frequency fist dependent on the VCO, where fist=15 MHz, is fed in at the other input 12. The phase differences of the two oscillations are determined in the phase detector 3. In this case, the phase detector 3 can also be configured as a phase frequency detector, which compares not only the phase, but also the frequency of the two oscillations fRef and fist. Depending on this comparison result, as already described in connection with <figref idref="DRAWINGS">FIG. 1</figref>, two output signals, a UP pulse and/or a DW pulse, having pulse durations depending on the magnitude of the phase difference, occur at the two outputs 14, 15. These pulses are used to control the final control element 36, which converts the measurement result, which is present in the form of pulses, into voltages with which the VCO 1 can be controlled. With each of these UP and DW pulses several (four in each case in this illustration) switches SU1, SU2, SU3, SU4 and SD1, SD2, SD3, SD4 are operated, and thereby the current sources 6, 7, 18, 19, 20, 21, 22 and 23 are connected or disconnected. This part of the arrangement forms the charge pump 4 of the final control element 36. In this example embodiment, two double-track channels 37, 38 are formed, wherein each track 371, 372 of one channel 37 and each track 381, 382 of the other channel 38 is connected respectively to a respective capacitor plate, as a result of which a potential difference is respectively built-up between the tracks 371 and 372 or 381 and 382 in a channel 37 or 38.

0029In the case of a UP pulse the current sources 6 and 19 generate a current I, where I=2.5 μA, by operation of the switches SU1 and SU2, in the case of a DW pulse current sources 7 and 18 generate the same current flowing in the opposite direction through CS1, by operation of the switches SD1 and SD2. In the case of a UP pulse the current sources 20 and 23 generate a current x·y·I, where x·y·I=12.5 μA, by operation of the switches SU3 and SU4, in the case of a DW pulse current sources 21 and 22 generate the same current flowing in the opposite direction by operation of the switches SD3 and SD4. The current sources are connected to capacitors 9, 29, 30, 31, 32 via the switches in such a manner that current source 6 charges respective first plates of the capacitors 9 and 31 via the UP pulse-dependent switch Su1, and current source 7 charges these same first plates of the capacitors 9 and 31 via a DW pulse-dependent switch SD1. The capacitance of capacitor 9 CS1 amounts to CS1=44 pF in this example embodiment, while the capacitance of the other capacitor 31 amounts to only 10 pF. The other plate of the capacitor 31 is connected to earth, i.e. ground. The other plate of capacitor 9 CS1, together with one plate of a fourth capacitor 32, is connected, on the one hand to the current source 19 via the switch SU2 and, on the other hand, to the current source 18 via the switch SD2. The other plate of the fourth capacitor 32 is connected to ground. One plate of the capacitor CS1 9 is connected to the input 24 of the VCO 1 together with one plate of the capacitor 31. The other plate of capacitor CS1 9 is connected to the input 25 of the VCO 1 together with one plate of the capacitor 32. The capacitance of the capacitor 32 amounts to 10 pF. A common mode loop (CML) 33 is arranged parallel to the capacitor 9 CS1. The purpose of this CML is to set the working point of the first channel 37 at (V3A+V3B)/2. In addition, a second common mode loop (CML) 33 is arranged in the second channel 38 parallel to capacitors 29, 30 and to resistor RS. This CML has the task of setting the working point (V2A+V2B)/2 of the second channel 38.

0030As a result of the arrangement of the current sources, a potential difference V3A−V3B is forwarded to the VCO. The currents I1 and I3, which flow at the VCO inputs, nonetheless still cause disturbances which are equally large in quantity and counterbalance one another because of their symmetric arrangement relative to one another, and therefore the potential difference V3A−V3B is fed into the VCO free from disturbance. One plate of the second capacitor 30 is connected to currant source 20 via the UP pulse-dependent switch SU3 and to current source 21 via the DW pulse-dependent switch SD3. The capacitance or capacitor 30 amounts to 10 pF. The other plate of the second capacitor is connected to earth. In the case of a fifth capacitor 29, one plate is supplied by current source 23 via the switch SU4 and by current source 22 via the switch SD4. The capacitance of the capacitor 29 amounts to 10 pF. The other plate of the capacitor 29 is connected to earth. A connection to a resistor RS 28 is located between one plate of the second capacitor 30 and one plate of the capacitor 29. The capacitors 9, 29, 30, 31, 32 and the resistor 28 form the loop filter 5 of the final control element 36. The value of the resistance 28 amounts to RS=60 kΩ in this example of application. The RS value determines the signal/noise ratio of the arrangement. One plate of the second capacitor 30 is connected to the third input 26 of the VCO 1. One plate of the fifth capacitor 29 is connected to the fourth input 27 of the VCO 1. As a result of the arrangement of the current sources 20, 21, 22, 23, a potential difference V2A−V2B is generated in the VCO 1 by the capacitors 30 and 29. The currents I2 and I4, which flow between the one plates of the capacitors 30 and 29 and the VCO inputs, nonetheless still cause disturbances which are equally large in quantity and counterbalance one another because of their symmetric arrangement relative to one another, and therefore the potential difference V2A−V2B can likewise be fed into the VCO free from disturbance. In the example of application, the VCO 1 is supplied with two potential difference values V3A−V3B, V2A−V2B and from the sum of these voltage values and the base frequency (K1vco(V3A−V3B))+(K2vco(V2A−V2B))+fvco0 generates an oscillation with the frequency fvco at its output 11. The two potential differences V3A−V3B, V2A−V2B contribute to the adjustment of the oscillation frequency to a different degree.

0031The potential difference V3A−V3B generated in the first channel 37 of capacitor 9 CS1 is only dependent on the useful signal which results from the phase difference between the actual and the reference frequency. Dynamic disturbances do not influence this potential difference. The potential difference V2A−V2B generated in the second channel 38 of capacitors 29 and 30 only differs from zero in the case of a disturbance. The second channel 38 determines the dynamic properties of the arrangement. In this example embodiment, different transconductances (or frequency-voltage grain characteristics) K1vco and K2vco are used at the VCO 1. The transconductance K1vco=1000MHz/v is assigned to the potential difference V3A−V3B dependent on the useful signal and the transconductance K2vco=100MHz/v is assigned to the potential difference V2A−V2B dependent on the disturbance signal. The two transconductances will generally differ greatly from one another K1vco>>K2vco. The factor between K1vco and K2vco is balanced by the dimensioning of the current sources 20, 21, 22 and 23, the resistor 28 and the capacitor 10. In the present example embodiment, the following interrelationship exists between the sensitivities K1vco, K2vco, the dimensioning of the resistor RS and the dimensioning of the current sources 20, 21, 22 and 23: K2vco=K1vco/y, wherein in the present example embodiment y=10, and RS/x=constant, wherein in the present example embodiment x=0.5. Since the signal/noise ratio is proportional to K2vco and √RS, a larger resistance value RS can be compensated by reducing the transconductance value K2vco. If the sum of the potential differences amounts to (V3A−V3B)=(V2A−V2B)=0V, then a base oscillation with a frequency of fvco0=915 Hz should be available at the output of the VCO 1. This circuit can operate with low noise with small capacitances of capacitors 9, 29, 30, 31, 32 and a resistor with a large resistance value RS.

0032As a result of the symmetric arrangement of the disturbance influences caused by the generation of potential differences, these disturbance influences counterbalance one another. As a result of the favourable scaling x·y of the current sources 20, 21, 22 and 23, the noise can be suppressed in spite of the high resistance on the basis of the low transconductance value K2vco. As a result of the feed of potential differences, this PLL circuit is also insensitive to fluctuations of supply systems. Because of the smaller components, in particular the capacitances, such a PLL circuit can be fully housed in a single integrated circuit (IC), especially if the IC is manufactured using bipolar, CMOS, BICMOS or other IC technology. It is an advantage, particularly in the communications field, if ICs operate independently of fluctuations in supply voltages or supply currents, are small and operate free from disturbance. External loop filters are no longer necessary. The additional current sources and switches can be accommodated in a very small space.

0033<figref idref="DRAWINGS">FIG. 3</figref> shows an alternative PLL circuit with operational amplifiers and the integrated functions of the loop filter. The VCO 1 makes available the oscillation frequency fvco of the PLL circuit at its output 11. The oscillation frequency fvco is divided in a frequency divider 2 by the factor N. This results in an oscillation with an actual frequency, which lies in the range of the reference frequency fRef and the phase difference of which can be determined by the phase detector. For this, the constant reference frequency fRef is fed in at one input 13 of the phase detector 3 and the actual frequency fist dependent on the VCO is fed in to the other input 12. The phase differences of the two oscillations are determined in the phase detector 3. In this case, the phase detector 3 can also be configured as a phase frequency detector, which compares not only the phase, but also the frequency of the two oscillations fRef and fist. Depending on this comparison result, as already described in connection with <figref idref="DRAWINGS">FIG. 1</figref>, two output signals, a UP pulse and/or a DW pulse, are formed at the two outputs 14, 15. These pulses are used to control the final control element 36, which converts the measurement result which is present in the form of pulses, into voltages with which the VCO 1 can be controlled. With each of these UP and DW pulses several (four in each case in this illustration) switches SU1, SU2, SU3, SU4 and SD1, SD2, SD3, SD4 are operated, and thereby the current sources 6, 7, 18, 19, 20, 21, 22 and 23 are connected or disconnected. This part of the arrangement forms the charge pump 4 of the final control element 36. In the example embodiment, two double-track channels 37, 38 are formed, wherein each track 371, 372 of one channel 37 and each track 381, 382 of the other channel 38 is connected respectively to a respective capacitor plate, as a result of which a potential difference is respectively built-up between the tracks 371 and 372 or 381 and 382 in a channel 37 or 38.

0034In the case of a UP pulse the current sources 6 and 19 generate a current I by operation of the switches SU1 and SU2, in the case of a DW pulse current sources 7 and 18 generate the same current acting in the opposite direction by operation of the switches SD1 and SD2. In the case of a UP pulse the current sources 20 and 23 generate a current x·y·I by operation of the switches SU3 and SU4, in the case of a DW pulse current sources 21 and 22 generate the same current acting in the opposite direction by operation of the switches SD3 and SD4. The current sources are connected to capacitors 9, 10 via the switches in such a manner that current source 6 charges one plate of the capacitor CS1, via the UP pulse-dependent switch SU1 and current source 7 charges this same plate of the capacitor CS1 via a DW pulse-dependent switch SD1. The other plate of CS1 is connected, on the one hand, to current source 19 via switch SU2 and, on the other hand, to current source 18 via switch SD2. Each plate of CS1 is connected to a respective input of a first operational amplifier OP134. The potential difference V3A−V3B is determined in the OP1 and the result V3 is connected to a first input of the VCO. The currents, which flow between the CS1 and the OP1, nonetheless still cause disturbances which are equally large in quantity and counterbalance one another because of their symmetric arrangement relative to one another, and therefore the potential difference V3A−V3B of the CS1 can be fed into the OP1 free from disturbance. The plates of the second capacitor CS2 10 are likewise operated with four current sources. One plate of the CS2 10 is connected to current source 20 via the UP pulse-dependent switch SU3 and to current source 21 via the DW pulse-dependent switch SD3. The other plate of the CS2 is supplied by current source 23 via switch SU4 and by current source 22 via switch SD4. A resistor RS 28 is arranged parallel to the CS2 10. The two capacitors 9, 10 and the resistor 28 form the RS loop filter 5 of the final control element 36. The RS 28 and the transconductance K2vco determine the signal/noise ratio of the arrangement. Each plate of the CS2 10 is connected to a respective input 26, 27 of a second operational amplifier OP235. As a result of this arrangement of the current sources 20, 21, 22, 23, a potential difference V2A−V2B is applied in the OP2 and the result is forwarded from the output of OP2 to the VCO. The currents, which flow between the CS2 and the OP2 inputs, nonetheless still cause disturbances which are equally large in quantity and counterbalance one another because of their symmetric arrangement relative to one another, and therefore the result V2 can likewise be fed into the VCO 1 free from disturbance. In this example embodiment, the VCO 1 is fed with two voltage values V3, V2 and from the sum of these voltage values and the base frequency generates an oscillation with the frequency fvco at its output 11. The two supplied voltages V3, V2 contribute to the adjustment of the oscillation frequency to a different degree. The potential difference V3A−V3B generated in the first channel 37 of CS1 is only dependent on the useful signal which results from the phase difference between the actual and the reference frequency. Dynamic disturbances do not influence this potential difference. The potential difference V2A−V2B generated in the second channel 38 of CS2 only differs from zero in the case of a disturbance. The second channel 38 determines the dynamic properties of the arrangement. In this example embodiment, different transconductances K1vco and K2vco are used at the VCO 1. The transconductance K1vco is assigned to the potential difference V3A−V3B dependent on the useful signal and the transconductance K2vco is assigned to the potential difference V2A−V2B dependent on the disturbance signal. The two transconductances will generally differ greatly from one another K1vco>>K2vco. However, they are sensitive to changes in the respective potential difference, and therefore K1vco=f(V3A−V3B) and K2vco=f(V2A−V2B) applies. However, since in the adjusted state V2A−V2B≈0, K2vco=constant, which in turn causes the constant dynamic properties because the dynamic properties are only dependent on K2vco. The factor between K1vco and K2vco is balanced by the dimensioning of the current sources 20, 21, 22 and 23, the resistor 28 and the capacitor 10. In this example embodiment, the following interrelationship exists between the sensitivities K1vco, K2vco, the dimensioning of the resistor RS and the dimensioning of the current sources 20, 21, 22 and 23: K2vco=K1vco/y and RS/x=constant. Since the signal/noise ratio is proportional to K2vco and √RS, a larger resistance value RS can be compensated by reducing the transconductance value K2vco. If the sum of the potential differences amounts to (V3A−V3B)=(V2A−V2B)=0V, then a base oscillation with a frequency of fvco0≠0 Hz should be available at the output of the VCO 1. This PLL circuit can operate with low noise with small capacitances of CS1 and CS2 and a resistor with a large resistance value RS. As a result of the symmetric arrangement of the disturbance influences, caused by the generation of potential differences, these disturbance influences counterbalance one another, as a result of the favorable scaling x·y of the current sources 20, 21, 22 and 23 the noise can be suppressed in spite of the high resistance by a low K2vco. As a result of the feed of potential differences, this PLL circuit is also insensitive to fluctuations of supply systems. Because of the smaller components, in particular the capacitances of CS1 and CS2, such a PLL circuit can be fully housed in a single integrated circuit (IC), especially if the IC is manufactured using bipolar or BICMOS technology. It is an advantage, particularly in the communications field, if ICs operate independently of fluctuations in supply voltages or supply currents, are small and operate free from disturbance. External loop filters are no longer necessary. The additional current sources, operational amplifiers and switches can be accommodated in a very small space.

0035It will be understood that the above description of the present invention is susceptible to various modifications, changes and adaptations.

Contents5

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Citations
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10050294	Germany	–	2000-10-10
10050294	Germany	A	2000-10-10
10050294	Germany	A	2000-10-10
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FR2815198A1	France	A1	2002-04-12
GB2367961A	United Kingdom	A	2002-04-17
DE10050294A1	Germany	A1	2002-07-11
ITMI20011928A1	Italy	A1	2003-03-14
GB2367961B	United Kingdom	B	2005-01-12
FR2815198B1	France	B1	2005-10-14
DE10050294B4	Germany	B4	2006-08-24
US7120217B2This record	United States of America	B2	2006-10-10

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