Apparatus and method for controlling brushless DC motor

CROSS-REFERENCE TO RELATED APPLICATION

0001This application claims the benefit of Korean Patent Application No. 2003-977, filed Jan. 8, 2003, in the Korean Intellectual Property Office, the disclosure of which is incorporated herein by reference.

BACKGROUND OF THE INVENTION

00021. Field of the Invention

0003The present invention relates, in general, to an apparatus and method for controlling motors and, more particularly, to an apparatus and method for controlling a brushless direct current motor, which minimizes torque ripple.

00042. Description of the Related Art

0005As is well known to those skilled in the art, a brushless Direct Current (DC) motor employs a rectifying circuit including switching devices instead of mechanical elements, such as a brush and a commutator. The brushless DC motor is characterized in that the replacement of the brush due to abrasion is not necessary, and little electromagnetic interference and drive noise exist. A control apparatus driving the brushless DC motor is described in detail with reference to <figref idref="DRAWINGS">FIG. 1</figref>.

0006<figref idref="DRAWINGS">FIG. 1</figref> is a block diagram of a conventional apparatus to control a brushless DC motor. As shown in <figref idref="DRAWINGS">FIG. 1</figref>, a power converting device including a converter <b>104</b>, a capacitor <b>108</b> and an inverter <b>106</b> converts Alternating Current (AC) power supplied from an AC power supply unit <b>102</b> to pulse-shaped 3-phase AC power, and supplies the 3-phase AC power to a brushless DC motor (BLDC motor) <b>110</b>. Of U, V and W phase currents of the 3-phase AC power supplied to the brushless DC motor <b>110</b> from the inverter <b>106</b>, U and V phase currents are detected by first and second current sensors <b>112</b><i>a </i>and <b>112</b><i>b</i>. The U and V phase current information detected by the first and second current sensors <b>112</b><i>a </i>and <b>112</b><i>b</i>, respectively, is provided to a control unit <b>114</b> and inverter control signals provided to the inverter <b>106</b> are based on the detected U and V phase current information. A position and a speed of a rotator of the brushless DC motor <b>110</b> are detected by a position/speed detecting unit <b>116</b>. The detected position/speed information of the rotator is further provided to the control unit <b>114</b> and inverter control signals provided to the inverter <b>106</b> are based on the detected position/speed information. That is, the control unit <b>114</b> controls the speed of rotation of the brushless DC motor <b>110</b> with reference to the phase current information inputted from the current sensors <b>112</b><i>a </i>and <b>112</b><i>b </i>and the rotator position/speed information inputted from the position/speed detecting unit <b>116</b>. The control unit <b>114</b> outputs inverter control signals to control phase commutation times of the 3-phase AC power outputted from the inverter <b>106</b> and the intensities of phase currents thereof, thus allowing the rotation speed of the brushless DC motor <b>110</b> to comply with a speed command.

0007Torque of the brushless DC motor may be expressed by the multiplication of an induced voltage by a current. In phase commutation periods of the 3-phase AC power, phase currents are temporarily decreased, thus generating torque ripple. Since the torque ripple is a cause of noise generation and vibration, a plan to minimize the torque ripple is required.

0008To minimize the torque ripple of the brushless DC motor, phase currents temporarily reduced during the phase commutation periods must be provided with compensation. The compensation must be performed only during the phase commutation periods. If the phase commutation periods are not accurately detected, undercompensation or overcompensation of the phase currents occurs. Accordingly, to minimize the torque ripple, accurately detecting the phase commutation periods is important. In the related art, current sensors, such as hall effect sensors, are used to detect the phase commutation periods of the brushless DC motor. However, using the current sensors to detect respective phase currents, as shown in <figref idref="DRAWINGS">FIG. 1</figref>, increases a manufacturing cost of a brushless DC motor control apparatus, thereby imposing a heavy burden to both a manufacturer and a user.

SUMMARY OF THE INVENTION

0009Accordingly, it is an aspect of the present invention to provide an apparatus and method for controlling a brushless direct current (DC) motor, which accurately detects phase commutation times and periods of 3-phase AC power and compensates for a reduction of currents occurring during the detected phase commutation periods so as to minimize torque ripple due to a temporary reduction of phase currents occurring during the phase commutation periods of 3-phase currents supplied to the brushless DC motor.

0010Additional aspects and/or advantages of the invention will be set forth in part in the description which follows and, in part, will be obvious from the description, or may be learned by practice of the invention.

0011The above and/or other aspects are achieved by providing an apparatus for controlling a brushless direct current DC motor, including a power converting unit to convert commercial alternating current (AC) power to polyphase AC power and supply the polyphase AC power to the brushless DC motor. A rotator position/speed detecting unit detects status information of a rotator of the brushless DC motor. A terminal voltage detecting unit detects variations of terminal voltages of the polyphase AC power supplied to the brushless DC motor. A control unit detects phase commutation periods of the polyphase AC power using the status information of the rotator provided from the rotator position/speed detecting unit and the terminal voltage variation information of the polyphase AC power provided from the terminal voltage detecting unit. Further, the control unit controls the power converting unit to supply to the brushless DC motor a compensation voltage used to constantly maintain a mean voltage of a non-commutation phase of the polyphase AC power during the detected phase commutation periods.

0012The above and/or other aspects are achieved by a method of controlling a brushless DC motor supplied with power from a power converting unit to convert commercial AC power to polyphase AC power. In the control method, the brushless DC motor is driven through a supply of a non-commutation phase current thereto. A starting time of a phase commutation is monitored using position information of a rotator. A compensation voltage is supplied to the brushless DC motor to minimize torque ripple due to the phase commutation when the phase commutation starts while the phase commutation starting time is monitored. When the phase commutation is complete, the supply of the compensation voltage is stopped.

BRIEF DESCRIPTION OF THE DRAWINGS

0013These and/or other aspects and advantages of the invention will become apparent and more readily appreciated from the following description of the preferred embodiments, taken in conjunction with the accompanying drawings of which:

0014<figref idref="DRAWINGS">FIG. 1</figref> is a block diagram of a Prior Art apparatus for controlling a brushless DC motor;

0015<figref idref="DRAWINGS">FIG. 2</figref> is a block diagram of an apparatus for controlling a brushless DC motor, according to an embodiment of the present invention;

0016<figref idref="DRAWINGS">FIG. 3</figref> is a circuit diagram of a 3-phase full-bridge inverter in the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>;

0017<figref idref="DRAWINGS">FIGS. 4A to 4F</figref> are graphs showing inverter control signals to drive the inverter of the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>;

0018<figref idref="DRAWINGS">FIG. 5</figref> is a block diagram of a control unit in the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>;

0019<figref idref="DRAWINGS">FIG. 6</figref> is a circuit diagram of a terminal voltage detecting unit in the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>;

0020<figref idref="DRAWINGS">FIGS. 7A to 7C</figref> are views showing directions of currents flowing through the inverter of <figref idref="DRAWINGS">FIG. 3</figref> during a phase commutation period;

0021<figref idref="DRAWINGS">FIGS. 8A to 8D</figref> are waveform diagrams showing input/output signals of the terminal voltage detecting unit of <figref idref="DRAWINGS">FIG. 6</figref>;

0022<figref idref="DRAWINGS">FIG. 9</figref> is a block diagram of a control signal converting unit in the control unit of the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 5</figref>;

0023<figref idref="DRAWINGS">FIGS. 10A to 10D</figref> are waveform diagrams showing inverter control signals and a PWM carrier signal of the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>; and

0024<figref idref="DRAWINGS">FIG. 11</figref> is a flowchart of a method of controlling the brushless DC motor, according to the embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

0025Reference will now be made in detail to the present preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings, wherein like reference numerals refer to like elements throughout.

0026<figref idref="DRAWINGS">FIG. 2</figref> is a block diagram of an apparatus for controlling a brushless DC motor according to an embodiment of the present invention. As shown in <figref idref="DRAWINGS">FIG. 2</figref>, a power converting device includes a converter <b>204</b>, a DC-link capacitor <b>208</b> and an inverter <b>206</b>. The power converting device converts AC power supplied from an AC power supply unit <b>202</b> to DC power, converts the DC power to 3-phase AC power, and provides the 3-phase AC power to a brushless DC motor <b>210</b>. The converter <b>204</b> converts the AC power to the DC power, and the inverter <b>206</b> converts the DC power to pulse-shaped 3-phase AC power.

0027A terminal voltage detecting unit <b>216</b> detects phase terminal voltages U, V and W of the 3-phase AC power supplied to the brushless DC motor <b>210</b> from the inverter <b>206</b>, and provides terminal voltage information U_C, V_Cand W_Cof the phases to a control unit <b>214</b>. A position/speed detecting unit <b>246</b> obtains position/speed information F/G of a rotator of the brushless DC motor <b>210</b>, and provides the position/speed information F/G to the control unit <b>214</b>. The position/speed detecting unit <b>246</b> detects a position of the rotator using a position sensor, or phase commutation information of the phase terminal voltages U, V and W supplied to the brushless DC motor <b>210</b>. The control unit <b>214</b> generates inverter control signals P<b>1</b> to P<b>6</b> to control a rotation speed of the brushless DC motor <b>210</b> with reference to the terminal voltage information U_C, V_Cand W_Cof the 3-phase AC power, provided from the terminal voltage detecting unit <b>216</b>, and rotator position/speed information F/G of the brushless DC motor <b>210</b>, provided from the position/speed detecting unit <b>246</b>. The inverter control signals P<b>1</b> to P<b>6</b> generated by the control unit <b>214</b> are used to control commutation times and current amounts of respective phases U, V and W of the 3-phase AC power outputted to the brushless DC motor <b>210</b> from the inverter <b>206</b>. The inverter control signals P<b>1</b> to P<b>6</b> allow the rotation speed of the brushless DC motor <b>210</b> to comply with a speed command H inputted from an outside.

0028<figref idref="DRAWINGS">FIG. 3</figref> is a circuit diagram of the 3-phase full-bridge inverter <b>206</b> in the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>. Referring to <figref idref="DRAWINGS">FIG. 3</figref>, respective switching devices, which are transistors Q<b>1</b> to Q<b>6</b>, are switched on/off in response to the 3-phase AC signals P<b>1</b> to P<b>6</b> each having a 2-phase excitation pattern, which will be described later in <figref idref="DRAWINGS">FIG. 4</figref>, thereby commuting phase currents.

0029<figref idref="DRAWINGS">FIGS. 4A to 4F</figref> are graphs showing the inverter control signals to drive the inverter <b>206</b> in the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>. In this case, <figref idref="DRAWINGS">FIGS. 4A to 4F</figref> show the inverter control signals P<b>1</b> to P<b>6</b> which are implemented in a form of the 3-phase AC signals each having the 2-phase excitation pattern and used in a rear end unipolar pulse width modulation. In the rear end unipolar pulse width modulation, a pulse width is modulated by a turn-on/off operation of a single switching device in a latter <b>600</b> interval (out-going phase) of a 120° conduction period.

0030<figref idref="DRAWINGS">FIG. 5</figref> is a block diagram of the control unit <b>214</b> in the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 2</figref>. As shown in <figref idref="DRAWINGS">FIG. 5</figref>, an adder <b>502</b> arranged at an input end of the control unit <b>214</b> calculates a difference between the speed command H provided from the outside and speed information G provided from the position/speed detecting unit <b>246</b>, and provides the difference to a speed control unit <b>504</b>. The speed control unit <b>504</b> generates a first current control signal A to allow the rotation speed of the brushless DC motor <b>210</b> to comply with the speed command H in consideration with the difference between the speed command H and the speed information G. The first current control signal A outputted from the speed control unit <b>504</b> is provided to a control signal converting unit <b>510</b>. The control signal converting unit <b>510</b> converts the first current control signal A outputted from the speed control unit <b>504</b> to internally generate a second current control signal B (not shown in <figref idref="DRAWINGS">FIG. 5</figref>, refer to <figref idref="DRAWINGS">FIGS. 8A to 8D</figref>). The control signal converting unit <b>510</b> selects one of the first current control signal A and the internally generated second current control signal B (not shown), and outputs the selected signal as a third current control signal C. In this case, a selection is performed in response to a commutation period detection signal D provided from a commutation period detecting unit <b>516</b>.

0031A commutation time detecting unit <b>518</b> detects a time when each phase commutation starts, and provides the detected phase commutation starting time information t_Cto the commutation period detecting unit <b>516</b>. The commutation period detecting unit <b>516</b> detects accurate commutation periods using the phase commutation starting time information t_C, provided from the commutation time detecting unit <b>518</b>, and the terminal voltage information U_C, V_Cand W_Cof respective phases, provided from the terminal voltage detecting unit <b>216</b>, and generates the commutation period detection signal D using the detected commutation period information. That is, the commutation period detection signal D is activated only during the phase commutation periods to allow the control signal converting unit <b>510</b> to output the second current control signal B as the third current control signal C. During a 2-phase conduction period, the commutation period detecting signal D is inactivated, so the first current control signal A is outputted as the third current control signal C during the 2-phase conduction period. The third current control signal C, outputted to an inverter control unit <b>514</b> from the control signal converting unit <b>510</b>, determines a time ratio of each of the inverter control signals P<b>1</b> to P<b>6</b> having the 2-phase excitation pattern, outputted from the inverter control unit <b>514</b>.

0032<figref idref="DRAWINGS">FIG. 6</figref> is a circuit diagram of the terminal voltage detecting unit <b>216</b> of <figref idref="DRAWINGS">FIG. 2</figref>. As shown in <figref idref="DRAWINGS">FIG. 6</figref>, the phase terminal voltages U, V and W supplied to the brushless DC motor <b>210</b> and a DC-link voltage V_DCacross ends of the DC-link capacitor <b>208</b> are inputted to the terminal voltage detecting unit <b>216</b>. The phase terminal voltages U, V and W, and the DC-link voltage V_DCare each divided in ratios of the resistors R<b>1</b>, R<b>2</b>, R<b>3</b> and R<b>4</b> constituting the terminal voltage detecting unit <b>216</b>, and each pair of corresponding voltages determined as corresponding ratios of the resistors R<b>1</b>, R<b>2</b>, R<b>3</b> and R<b>4</b> are compared with each other by respective comparators <b>602</b>, <b>604</b> and <b>606</b>. The compared results of the pair of corresponding voltages are outputted as the terminal voltage information U_C, V_Cand W_C. That is, each of the phase terminal voltages is compared with, for example, ½ of the DC-link voltage V_DCacross the ends of the DC-link capacitor <b>208</b>. If each of the phase terminal voltages is greater than V_DC/2, each terminal voltage information U_C, V_C, and W_Cis outputted as a “1” state (a high logic level state), while if each of the phase terminal voltages is less than V_DC/2, each terminal voltage information is outputted as a “0” state (a low logic level state). A principle of obtaining the terminal voltage information U_C, V_Cand W_Cof the respective phases through the comparison of the phase terminal voltages U, V and W with V_DC/2, and detecting the phase commutation periods using the phase terminal voltage information U_C, V_Cand W_Cis described below.

0033<figref idref="DRAWINGS">FIGS. 7A to 7C</figref> are views showing directions of currents flowing through the inverter <b>206</b> during a phase commutation period. <figref idref="DRAWINGS">FIGS. 7A to 7C</figref> illustrate by stages current flows generated during a period in which a U-V phase (0 to 60° interval) is commuted into a U-W phase (60 to 120° interval). In <figref idref="DRAWINGS">FIG. 7A</figref>, a current flow of the inverter <b>206</b> in the U-V phase (0 to 60° interval) is depicted. During the 0 to 600 interval, transistors Q<b>1</b> and Q<b>6</b> are turned on to allow a U-V phase current to flow through the transistors Q<b>1</b> and Q<b>6</b>.

0034In this state, the transistor Q<b>6</b> is turned off and the transistor Q<b>2</b> is turned on, so that the phase commutation from the U-V phase to the U-W phase begins, as shown in <figref idref="DRAWINGS">FIGS. 7B and 7C</figref>. <figref idref="DRAWINGS">FIGS. 7B and 7C</figref> illustrate current flows of an extinguished V phase current <b>702</b> and an ignited W phase current <b>704</b> during a phase commutation period. In <figref idref="DRAWINGS">FIG. 7B</figref>, transistors Q<b>1</b> and Q<b>2</b> are turned on, such that the newly ignited W phase current <b>704</b> appears on a negative (−) terminal of the DC-link capacitor <b>208</b>. In <figref idref="DRAWINGS">FIG. 7C</figref>, the transistor Q<b>1</b> is turned off and the transistor Q<b>2</b> continues to be turned on. In this case, the extinguished V phase current <b>106</b> flows through a diode D<b>3</b>. Therefore, the V phase terminal voltage V has a same intensity (i.e., magnitude or instantaneous value) as that of the DC-link voltage V_DCuntil the extinguished V phase current decreases to “0”. After the phase commutation from the V phase to the W phase is complete, the terminal voltage V of the opened V phase terminal is expressed by the following Equations (1),

0035<maths id="MATH-US-00001" num="00001"><math overflow="scroll"><mtable><mtr><mtd><mtable><mtr><mtd><mrow><mi>V</mi><mo>=</mo><mrow><msub><mi>E</mi><mi>v</mi></msub><mo>+</mo><mrow><mfrac><msub><mi>V</mi><mrow><mi>D</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><mi>C</mi></mrow></msub><mn>2</mn></mfrac><mo>⁢</mo><mstyle><mspace width="1.1em" height="1.1ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mrow><mi>Q1</mi><mo>=</mo><mi>ON</mi></mrow><mo>)</mo></mrow></mrow></mrow></mrow></mtd></mtr><mtr><mtd><mrow><mi>V</mi><mo>=</mo><mrow><mn>0</mn><mo>⁢</mo><mstyle><mspace width="6.4em" height="6.4ex" /></mstyle><mo>⁢</mo><mrow><mo>(</mo><mrow><mi>Q1</mi><mo>=</mo><mi>OFF</mi></mrow><mo>)</mo></mrow></mrow></mrow></mtd></mtr></mtable></mtd><mtd><mrow><mo>(</mo><mn>1</mn><mo>)</mo></mrow></mtd></mtr></mtable></math></maths><br /> where E_Vis a back electromotive force (EMF) of the V phase. Since an intensity of the back EMF E_Vis less than “0” immediately after the phase commutation is complete, the V phase terminal voltage is less than V_DC/2. That is, the terminal voltage V of the opened V phase terminal is greater than V_DC/2 during the phase commutation period, and is less than V_DC/2 after the phase commutation is complete. Therefore, the phase commutation period may be accurately detected by comparing each of the phase terminal voltages with V_DC/2 using a comparison circuit shown in <figref idref="DRAWINGS">FIG. 6</figref>.

0036<figref idref="DRAWINGS">FIGS. 8A to 8D</figref> are waveform diagrams showing input/output signals of the terminal voltage detecting unit <b>216</b> of <figref idref="DRAWINGS">FIG. 6</figref> such that respective phase terminal voltages U, V and W are depicted together with the phase terminal voltage information U_C, V_Cand W_Cthereof. As shown in <figref idref="DRAWINGS">FIGS. 8A to 8D</figref>, each phase terminal voltage information U_C, V_Cand W_Cis in a “1” state during intervals in which each of the phase terminal voltages U, V and W is greater than V_DC/2. Further, each phase terminal voltage information U_C, V_Cand W_Cis in a “0” state during intervals in which each of the phase terminal voltages U, V and W is less than V_DC/2.

0037In <figref idref="DRAWINGS">FIGS. 8A to 8D</figref>, three phase commutation times #1, #2 and #3, which represent commutation starting times of the W, V and U phases, respectively, are depicted. As shown in <figref idref="DRAWINGS">FIGS. 7A to 7C</figref>, the terminal voltage of an extinguished phase has the intensity of “0” or V_DCuntil a current of the extinguished phase decreases to “0” after each phase commutation begins. Therefore, a terminal voltage of the phase extinguished immediately after phase commutation takes place is compared with V_DC/2, and a period in which a compared result is maintained at “1” or “0” is a phase commutation period.

0038In <figref idref="DRAWINGS">FIGS. 8A to 8D</figref>, after the commutation time #1, the W phase terminal voltage is maintained at V_DC, whereby an output of the comparator <b>606</b> of <figref idref="DRAWINGS">FIG. 6</figref> is maintained at “1” for a time period t_W. The time period t_Wis the phase commutation period of the W phase. Further, after the commutation time #2, the V phase terminal voltage is maintained at “0”, whereby an output of the comparator <b>604</b> is maintained at “0” for a time period t_V. The time period t_Vis a phase commutation period of the V phase. Similar to this, after the commutation time #3, the U phase terminal voltage is maintained at V_DC, whereby an output of the comparator <b>602</b> is maintained at “1” for a time period t_U. The time period t_Uis a phase commutation period of the U phase. Therefore, the commutation period detecting unit <b>516</b> of <figref idref="DRAWINGS">FIG. 5</figref> recognizes each phase commutation starting time using the phase commutation starting time information t_Cprovided from the commutation time detecting unit <b>518</b>, and allows the control signal converting unit <b>510</b> to output the third current control signal C capable of preventing voltage variation due to the phase commutation for time periods t_U, t_Vand t_W, each representing a period from the phase commutation starting time to a completion of the phase commutation, thus minimizing torque ripple generated during the phase commutation periods.

0039<figref idref="DRAWINGS">FIG. 9</figref> is a block diagram of the control signal converting unit <b>510</b> of the brushless DC motor control apparatus of <figref idref="DRAWINGS">FIG. 5</figref>. As shown in <figref idref="DRAWINGS">FIG. 9</figref>, the control signal converting unit <b>510</b> outputs one of the first current control signal A provided from the speed control unit <b>504</b> and the internally generated second current control signal B as the third current control signal C through a switch <b>910</b>. The second current control signal B is generated through the following process. A first amplifier <b>902</b> with a gain of ¾ amplifies the first current control signal A, and provides an amplified result to an adder <b>908</b>. A second amplifier <b>904</b> with a gain of ½ amplifies a signal A_MAXwhich is a maximum intensity of the first current control signal A and provides an amplified result to the adder <b>908</b>. A signal generator <b>906</b> generates a signal K_eω/2V_DC, and provides the signal K_eω/2V_DCto the adder <b>908</b>. In this case, K_eis a back electromotive force constant, ω is a rotation speed of a rotator of the brushless DC motor <b>210</b>, and V_DCis a voltage across the DC-link capacitor <b>208</b>. The signal added by the adder <b>908</b> is variably amplified by a Variable Gain Control (AGC) amplifier <b>914</b> to generate the second current control signal B. During a conduction period, a contact point of the switch <b>910</b> is connected to a terminal S<b>1</b>, so that the first current control signal A is outputted as the third current control signal C. Further, during the phase commutation period, the contact point of the switch <b>910</b> is connected to a terminal S<b>2</b> by an activated commutation period detection signal D, so that the second current control signal B to compensate for a mean voltage variation generated during the phase commutation period is outputted as the third current control signal C.

0040A gain K of the AGC amplifier <b>914</b> that outputs the second current control signal B is controlled in response to a gain control signal E provided from the commutation period detecting unit <b>516</b>. The gain K is used to prevent a current ripple due to overcompensation generated when the phase commutation is completed within one cycle of a PWM carrier signal. The gain K has a value between 0 to 1, and the value thereof is determined depending on a ratio of a width of the phase commutation period to a cycle of the PWM carrier signal inputted to the inverter control unit <b>514</b>. For example, if the cycle of the PWM carrier signal is 250 μs and the width of the phase commutation period is 125 μs, the gain K is 0.5. At this time, the second control current signal B is defined by a value obtained by multiplying a time ratio of each of the inverter control signals P<b>1</b> to P<b>6</b> used to control the transistors Q<b>1</b> to Q<b>6</b> of the inverter <b>206</b> by the gain K. If the cycle of the PWM carrier signal is 250 μs and the width of the phase commutation period is 300 μs, that is, if the phase commutation period is longer than one cycle of the PWM carrier signal, the gain K is 1 during one cycle of the PWM carrier signal immediately after the phase commutation, and the gain K is 0.2 during the next cycle of the PWM carrier signal. In this way, if the phase commutation period is longer than the one cycle of the PWM carrier signal, the commutation period detection signal D controls the switch <b>910</b> to output the second current control signal B as the third current control signal C during two cycles of the PWM carrier signal. Generally, the time ratio of each of the inverter control signals P<b>1</b> to P<b>6</b> has a value between 0 to 1, so a limiter <b>912</b> limits the time ratio of the third current control signal C to a range from 0 to 1.

0041As described above, to control the time ratio of each of the inverter control signals P<b>1</b> to P<b>6</b> is important so as to compensate for the mean voltage variation during the phase commutation period and is described below in detail. First, if the U-V phase is conducting, a mean phase voltage V_M1applied to each of conducting U and V phases is defined by the following Equation (2),

0042<maths id="MATH-US-00002" num="00002"><math overflow="scroll"><mtable><mtr><mtd><mrow><msub><mi>V</mi><mi>M1</mi></msub><mo>=</mo><mrow><mfrac><mrow><msub><mi>V</mi><mrow><mi>D</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><mi>C</mi></mrow></msub><mo>·</mo><msub><mi>T</mi><mn>1</mn></msub></mrow><mn>2</mn></mfrac><mo>⁢</mo><mrow><mo>(</mo><mrow><mn>0</mn><mo>&lt;</mo><msub><mi>T</mi><mn>1</mn></msub><mo>&lt;</mo><mn>1</mn></mrow><mo>)</mo></mrow></mrow></mrow></mtd><mtd><mrow><mo>(</mo><mn>2</mn><mo>)</mo></mrow></mtd></mtr></mtable></math></maths><br /> where V_DCis a voltage across the ends of the DC-link capacitor <b>208</b> in parallel with the inverter <b>206</b>, and T₁is a time ratio of the inverter control signal P<b>6</b> applied to the transistor Q<b>6</b> of the inverter <b>206</b>. A mean voltage V_M2applied to a non-commutation U phase during a phase commutation period from the U-V phase conduction to a U-W phase conduction is expressed by the following Equation (3),

0043<maths id="MATH-US-00003" num="00003"><math overflow="scroll"><mtable><mtr><mtd><mrow><msub><mi>V</mi><mi>M2</mi></msub><mo>=</mo><mrow><mrow><mfrac><msub><mi>V</mi><mrow><mi>D</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><mi>C</mi></mrow></msub><mn>3</mn></mfrac><mo>⁢</mo><mrow><mo>(</mo><mrow><mrow><mn>2</mn><mo>⁢</mo><msub><mi>T</mi><mn>2</mn></msub></mrow><mo>-</mo><mn>1</mn></mrow><mo>)</mo></mrow></mrow><mo>-</mo><mrow><mfrac><mrow><msub><mi>E</mi><mi>U</mi></msub><mo>+</mo><msub><mi>E</mi><mi>V</mi></msub><mo>+</mo><msub><mi>E</mi><mi>W</mi></msub></mrow><mn>3</mn></mfrac><mo>⁢</mo><mrow><mo>(</mo><mrow><mn>0</mn><mo>&lt;</mo><msub><mi>T</mi><mn>2</mn></msub><mo>&lt;</mo><mn>1</mn></mrow><mo>)</mo></mrow></mrow></mrow></mrow></mtd><mtd><mrow><mo>(</mo><mn>3</mn><mo>)</mo></mrow></mtd></mtr></mtable></math></maths><br /> where T₂is a time ratio of the inverter control signal P<b>1</b> applied to the transistor Q<b>1</b> of the inverter <b>206</b> during the phase commutation period from the U-V phase to the U-W phase, and E_U, E_Vand E_Ware back electromotive forces of U, V and W phases, respectively.

0044As indicated in Equations (2) and (3), if the time ratios T₁and T₂of the inverter control signals P<b>6</b> and P<b>1</b> during the 2-phase conduction period and the phase commutation period, respectively, are the same, the mean voltage V_M1applied to the non-commutation U phase is temporarily varied to V_M2due to the phase commutation. The temporary variation of the mean voltage causes ripple in the non-commutation phase current. Torque of the brushless DC motor <b>210</b> is proportional to the non-commutation phase current during the phase commutation interval, so the ripple of the non-commutation phase current due to the phase commutation causes a torque ripple. Therefore, to reduce the torque ripple due to the phase commutation, the time ratio T₂of the inverter control signal P<b>1</b> provided to the inverter <b>206</b> during the phase commutation period is adjusted, as shown in Equation (4), so as to allow a constant current without a current ripple to flow through the non-commutation phase during the phase commutation period, that is, V_M1to be V_M2.

0045<maths id="MATH-US-00004" num="00004"><math overflow="scroll"><mtable><mtr><mtd><mrow><msub><mi>T</mi><mn>2</mn></msub><mo>=</mo><mrow><mfrac><mn>1</mn><mn>2</mn></mfrac><mo>+</mo><mrow><mfrac><mn>3</mn><mn>4</mn></mfrac><mo>⁢</mo><msub><mi>T</mi><mn>1</mn></msub></mrow><mo>+</mo><mfrac><mrow><msub><mi>E</mi><mi>U</mi></msub><mo>+</mo><msub><mi>E</mi><mi>V</mi></msub><mo>+</mo><msub><mi>E</mi><mi>W</mi></msub></mrow><mrow><mn>2</mn><mo>⁢</mo><msub><mi>V</mi><mrow><mi>D</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><mi>C</mi></mrow></msub></mrow></mfrac></mrow></mrow></mtd><mtd><mrow><mo>(</mo><mn>4</mn><mo>)</mo></mrow></mtd></mtr></mtable></math></maths>

0046Provided that an intensity of each of the back electromotive forces E_U, E_Vand E_Wof U, V and W phases is constant and is proportional to a rotation speed of the brushless DC motor <b>210</b> during the phase commutation period, Equation (4) may be simplified as the following Equation (5).

0047<maths id="MATH-US-00005" num="00005"><math overflow="scroll"><mtable><mtr><mtd><mrow><msub><mi>T</mi><mn>2</mn></msub><mo>=</mo><mrow><mfrac><mn>1</mn><mn>2</mn></mfrac><mo>+</mo><mrow><mfrac><mn>3</mn><mn>4</mn></mfrac><mo>⁢</mo><msub><mi>T</mi><mn>1</mn></msub></mrow><mo>+</mo><mfrac><mrow><msub><mi>K</mi><mi>e</mi></msub><mo>·</mo><mi>ω</mi></mrow><mrow><mn>2</mn><mo>⁢</mo><msub><mi>V</mi><mrow><mi>D</mi><mo>⁢</mo><mstyle><mspace width="0.3em" height="0.3ex" /></mstyle><mo>⁢</mo><mi>C</mi></mrow></msub></mrow></mfrac></mrow></mrow></mtd><mtd><mrow><mo>(</mo><mn>5</mn><mo>)</mo></mrow></mtd></mtr></mtable></math></maths>

0048Consequently, when a 2-phase excitation type brushless DC motor <b>210</b> is controlled, the variation of the mean voltage applied to the non-commutation phase during the phase commutation period must be prevented so as to reduce the current ripple due to the phase commutation. For these operations, the time ratio of each of the inverter control signals P<b>1</b> to P<b>6</b> must be controlled in accordance with the above Equation (5) simultaneously with the phase commutation. If the time ratio of each of the inverter control signals P<b>1</b> to P<b>6</b> is T₁during a 2-phase conduction period, the mean voltage variation of the non-commutation phase may be prevented only when the time ratio thereof is changed to a time ratio T₂simultaneously with a start of the phase commutation and is again changed to a time ratio T₁at a time when the phase commutation is completed (that is, when the phase current of an extinguished phase becomes “0”).

0049If the time ratio thereof is varied to the time ratio T₁from the time ratio T₂, in advance, before the phase commutation is complete, a reduction of the mean voltage due to the phase commutation may not be of a sufficient compensation (undercompensation), thus causing the current ripple. If the time ratio thereof is continuously maintained at the time ratio T₂even after the phase commutation is complete, the current ripple is caused due to an overcompensation. Therefore, to minimize the torque ripple due to the phase commutation, the phase commutation period must be accurately detected and the time ratio of each of the inverter control signals P<b>1</b> to P<b>6</b> must be maintained at the time ratio T₂only during the phase commutation period.

0050<figref idref="DRAWINGS">FIGS. 10A to 10D</figref> are waveform diagrams showing the inverter control signals to control transistors Q₁, Q₂and Q₆and the PWM carrier signal of the brushless DC motor control apparatus of the embodiment of the present invention. In the case where the control unit <b>214</b> of <figref idref="DRAWINGS">FIG. 5</figref> is used to reduce the torque ripple due to phase commutation, the PWM carrier signal must be synchronized with a turn-on time of a newly ignited switching device, as shown in FIGS. <b>10</b>A to <b>10</b>D, otherwise, the mean voltage applied to the non-commutation phase is varied at the turn-on time of the newly ignited switching device, thus causing the current ripple. As shown in <figref idref="DRAWINGS">FIGS. 10A to 10D</figref>, when the frequency of the PWM carrier signal is changed so as to synchronize the PWM carrier signal with the turn-on time of the newly ignited switching device, the time ratio of the PWM carrier signal must be maintained at a constant value by suitably adjusting a turn-on time of a PWM pulse. If only the frequency of the PWM carrier signal is changed while the turn-on time of the PWM pulse is constant, an excessive or insufficient input signal is applied to cause the current ripple. In order to synchronize the PWM carrier signal with an ignition time of a new phase, the carrier frequency of the PWM pulse applied to the extinguished phase may be changed immediately before the phase commutation based on the commutation time, as shown in <figref idref="DRAWINGS">FIGS. 10A to 10D</figref>. Further, a method of changing the carrier frequency of the PWM carrier signal by an angle of 60° may also be used.

0051<figref idref="DRAWINGS">FIG. 11</figref> is a flowchart of a method of controlling the brushless DC motor according to the embodiment of the present invention. As shown in <figref idref="DRAWINGS">FIG. 11</figref>, a non-commutation phase current is supplied to drive the brushless DC motor <b>210</b> at operation <b>1102</b>. A starting time of a phase commutation is monitored using position information of a rotator at operation <b>1104</b>. If the phase commutation starts in operation <b>1106</b>, a compensation voltage is supplied to the brushless DC motor <b>210</b> to minimize a torque ripple in operation <b>1108</b>. If the phase commutation is complete in operation <b>1110</b>, the supply of the compensation voltage is stopped.

0052As is apparent from the above description, an apparatus and method for controlling a brushless DC motor is provided, which can accurately detect phase commutation times and phase commutation periods of 3-phase AC power and compensate for a reduction of phase currents occurring during the detected phase commutation periods so as to minimize a torque ripple due to the temporary reduction of phase currents occurring during the phase commutation periods of 3-phase currents supplied to the brushless DC motor.

0053Although an embodiment of the present invention has been shown and described, it would be appreciated by those skilled in the art that changes may be made in the embodiment without departing from the principles and spirit of the invention, the scope of which is defined in the claims and their equivalents.