Distance to fault measurements in cable TV networks

PRIORITY

0001This application is a Continuation of commonly assigned and co-pending U.S. patent application Ser. No. 15/802,232, filed Nov. 2, 2017, which is a Continuation of U.S. patent application Ser. No. 14/811,630, filed Jul. 28, 2015, now U.S. Pat. No. 9,838,679, issued Dec. 5, 2017.

TECHNICAL FIELD

0002The present disclosure generally relates to testing and monitoring of cable TV networks, and more particularly relates to devices and methods for locating faults in a cable TV network.

BACKGROUND

0003Providers of digital television (DTV) services typically use two-way hybrid fiber-coaxial (HFC) networks, which are shared bi-directional networks with point-to-multipoint transmission in the downstream direction using digital signals or a mix of analog and digital signals, and multipoint-to-point transmission in the upstream direction. Signals are distributed via a fiber optic connection from a head-end to a node that converts the optical signal to an electrical signal and then distributes the signals to residences via a tree and branch coaxial cable distribution network termed ‘cable plant’. At the subscriber side, terminal equipment supports the delivery of cable services, which may include video, data and voice services, to subscribers via cable modems.

0004Delivery of data services over cable networks, and in particular cable television (CATV) networks, is typically compliant with a Data Over Cable Service Interface Specifications (DOCSIS®) standard. The term ‘DOCSIS’ generally refers to a group of specifications published by CableLabs that define industry standards for cable headend equipment, such as Cable Modem Termination System (CMTS), and cable modem (CM) equipment. The physical layer specification of DOCSIS provides for the use of frequency multiplexing and several specific forms of quadrature amplitude modulation (QAM) for both upstream (CM to headend) and downstream (headend to CM) communications. Upstream and downstream signals occupy separate frequency bands called upstream and downstream frequency bands. Downstream information channel signals co-propagate in the downstream frequency band, and upstream signals co-propagate in the upstream frequency band. The frequency separation of the upstream and the downstream signals allows bidirectional amplification of these signals, which propagate in a common cable in opposite directions. In the United States, most of the cable equipment installed at the time of the writing complies with the DOCSIS 3.0 version of the DOCSIS standard, which provides for the upstream spectral band from 5 MHz to 42 MHz typically, with the downstream channels using 64-QAM or 256-QAM modulation format and 6 MHz spacing within the downstream spectral band spanning from 50 MHz to 860 MHz. The upstream channel widths are configurable and may take a set of define values between 200 kHz and 6.4 MHz, each corresponding to a specific symbol rate, with the upstream data modulated with either QPSK, 16-QAM, 32-QAM, 64-QAM or 128-QAM.

0005The upstream and downstream signals are prone to impairments that may originate at a plurality of network locations in the network. As the result of the “tree” structure of the cable plant, there may be numerous devices, cable segments and connectors located between the fiber optic node and the end user. This provides for a plurality of locations were a defect can occur, resulting in either no service or a reduced service to the end user. In order to ensure adequate performance, the cable plant needs to be monitored and tested and the source of impairments identified and located.

0006Tracing the source of impairment typically requires that a technician travels to different network locations and compares measurements to locate the impairment. Portable network testing devices currently used in the industry may help to identify certain types of defects in the cable plant by performing specific spectral and noise measurements in the upstream and/or downstream directions using specialized testing methods at different network locations. A number of tests can also be performed to evaluate quality of digital TV signal transmission on higher logical levels of data transmission, for example by measuring such parameters as carrier level or amplitude, modulation error ratio (MER), bit error rate (BER), ingress under carrier (IUC), and other parameters. The measurements may be performed on channel-by-channel basis, each channel diagnostic data being summarized on a separate screen or data page viewed by the technician on the tester's visual display.

0007One type of defects that may be particularly hard to locate are defects that lead to changes in impedance along the signal path in the cable plant. These defects may be caused by cable corrosion, which may result from the cable getting scratched and the outer shield rusting away due to exposure to water, “rodent chews”, crushed, pinched or kinked cables, opens, shorts or partials in the cable. Impedance changes may also be caused by set screws inside housings, i.e. the screws that connect the center conductor of the cable to internal circuits of the amplifiers, splitters, taps, and fiber nodes, becoming loose if they are under-torqued or causing oxidation through the galvanic process if they are over torqued. Defects of these types may be located using time domain reflectometry (TDR), which may include launching a short pulse into the cable and detecting reflections from the location of the impedance change, with the time delay between the transmission and the reflection indicating the distance to the fault. This may, however, require that the service to the customers be disconnected during the measurements so that the strong TDR pulses do not interfere with the downstream TV signals at the end user locations, and the weak reflected TDR pulses are not obscured by the upstream DTV signals from the end users. As the service to many customers may be impacted due to the tree structure of the cable plant, cable operators are understandably reluctant to perform such measurements due to potential customer complaints. Another approach could be to replace all possibly suspicious connections, cables and/or devices hoping that the defected part is among them. Drawbacks of this approach includes increased costs and that the root cause of the problem remains un-identified.

0008Accordingly, it may be understood that there may be significant problems and shortcomings associated with current solutions and technologies for locating impedance-changing faults in a cable TV network.

SUMMARY

0009Accordingly, the present disclosure relates to a method and device for locating a fault in a cable TV network using chirped TDR pulses to spread TDR pulse energy over a relatively long time interval, and further using pulse-matched filtering to process signals returned from the cable network and to magnify echoes of the TDR pulses that may be comprised therein.

0010According to one aspect of the present disclosure, the method includes using a cable network tester comprising a signal transmitter and a signal receive, the method further comprising: a) using the signal transmitter of the network tester to generate a pulsed probe signal and to launch said pulsed probe signal into the cable TV network, wherein the pulse probe signal comprises a probe pulse that is characterized by a probe signal frequency that sweeps across a pre-defined probe frequency band over the duration of the probe pulse; b) using the signal receiver of the network tester to receive a return signal from the cable TV network; c) filtering the return signal with a matched filter that is matched to the probe pulse; and d) analyzing the return signal to identify one or more peaks therein corresponding to one or more echoes of the probe pulse reflected at a fault location in the network.

0011Another aspect of the present disclosure relates to a cable network tester for locating a fault in a cable TV network, comprising: a signal transmitter configured to generate a pulsed probe signal and to launch said pulsed probe signal into the cable TV network, wherein the pulse probe signal comprises a probe pulse that is characterized by a probe signal frequency that sweeps across a pre-defined probe frequency band over the duration of the probe pulse; a signal receiver configured to receive a return signal from the cable TV network and comprising a matched filter that is matched to the probe pulse for filtering the return signal therewith and for amplifying in magnitude one or more echoes of the probe pulse in the return signal; and, a processor for analyzing the return signal to identify one or more peaks therein corresponding to the one or more echoes of the probe pulse reflected at a fault location in the network.

0012According to a feature of one or more embodiments disclosed herein, the duration of each probe pulse and/or a repetition rate of the probe pulse generation may be selected so that symbol errors that may be caused by collisions of probe pulses with downstream signals at the subscriber end are correctable by an error correction scheme used by the cable network in the downstream transmission.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments disclosed herein will be described in greater detail with reference to the accompanying drawings which represent preferred embodiments thereof, in which like elements are indicated with like reference numerals, and wherein:

<figref idref="DRAWINGS">FIG. 1</figref> is a schematic block diagram of a portion of a cable network;

<figref idref="DRAWINGS">FIG. 2</figref> is a schematic representation of a chirped probe signal that may be used for locating faults in a cable network;

<figref idref="DRAWINGS">FIG. 3</figref> is a schematic block diagram of a cable network tester for locating faults in a cable network;

<figref idref="DRAWINGS">FIG. 4</figref> is a flowchart of an embodiment of a method for locating faults in a cable network using chirped probe pulses;

<figref idref="DRAWINGS">FIG. 5</figref> is a graph representing a matched filter waveform;

<figref idref="DRAWINGS">FIG. 6A</figref> is a schematic diagram illustrating signal connections through a Y-type connector at the output of the cable network tester of <figref idref="DRAWINGS">FIG. 3</figref>;

<figref idref="DRAWINGS">FIG. 6B</figref> is a schematic diagram illustrating an example three-leg summing resistor network implementing the Y-type connector of <figref idref="DRAWINGS">FIG. 6A</figref>;

<figref idref="DRAWINGS">FIG. 7</figref> is a flowchart illustrating processing of a return signal in an embodiment of the method of <figref idref="DRAWINGS">FIG. 4</figref> for locating faults in a cable network;

<figref idref="DRAWINGS">FIG. 8(A)</figref> is a graph illustrating a sequence of chirped probe pulses launched into the cable network;

<figref idref="DRAWINGS">FIG. 8(B)</figref> is a graph illustrating the slicing of return signal corresponding to the sequence of chirped probe pulses of <figref idref="DRAWINGS">FIG. 8(A)</figref>, including directly received copies of the chirped probe pulses;

<figref idref="DRAWINGS">FIG. 9</figref> is a schematic block diagram of the downstream signal transmission in a DTV network.

DETAILED DESCRIPTION

0025In the following description, for purposes of explanation and not limitation, specific details are set forth, such as particular circuits, circuit components, techniques, etc. in order to provide a thorough understanding of the present disclosure. However, it will be apparent to one skilled in the art that aspects of the present disclosure may be practiced in other embodiments that depart from these specific details. In other instances, detailed descriptions of well-known methods, devices, and circuits are omitted so as not to obscure the description. The appearances of the phrase “in one embodiment” in various places in the specification are not necessarily all referring to the same embodiment, nor are separate or alternative embodiments mutually exclusive of other embodiments. Note that as used herein, the terms “first”, “second” and so forth are not intended to imply sequential ordering, but rather are intended to distinguish one element from another unless explicitly stated. Furthermore, the following abbreviations may be used:

0026HFC Hybrid Fiber-Coaxial

0027LAN Local Area Network

0028ASIC Application Specific Integrated Circuit

0029FPGA Field Programmable Gate Array

0030DSP Digital Signal Processor

0031QAM Quadrature Amplitude Modulation

0032CPE Customer Premises Equipment

0033CMTS Cable Modem Termination System

0034RE Radio Frequency

0035RAM Random Access Memory

0036ADC Analog to Digital Converter

0037DAC Digital to Analog Converter

0038DOCSIS Data Over Cable Service Interface Specification

0039Exemplary embodiments described hereinbelow relate to a method and a related apparatus for in-service location of faults in a cable TV network using time domain reflectometry (TDR) with chirped TDR pulses. Advantageously, the method may be implemented in a multi-function portable cable TV tester that can also perform other cable test, functions known in the art, such as, for example, channel scan for active upstream and downstream channels, upstream channel demodulation and error analysis, etc.

0040With reference to <figref idref="DRAWINGS">FIG. 1</figref>, there is shown a schematic diagram of an exemplary cable TV network <b>1</b> wherein embodiments of the present disclosure may be practiced. The exemplary cable TV network <b>1</b>, which is also referred to herein simply as the cable network <b>1</b>, may be a local portion of an HFC network that delivers Cable Television (CATV) signals, including digital TV signals and data and control signals, to end users. It will be appreciated how ever that methods and techniques described herein can also be used in, other types of wired communication networks, including but not limited to those using QAM formats, such as for example DVB-T based digital TV systems, OFDM-based networks, DSL networks, and other types of networks, possibly with modifications which would be evident to those skilled in the art on the basis of the present disclosure.

0041As illustrated in <figref idref="DRAWINGS">FIG. 1</figref>, a fiber-optic node <b>10</b> of the cable network <b>1</b> includes a downstream (DS) optoelectronic converter <b>10</b>A that converts downstream (DS) optical signals generated by a remote CMTS (not shown) into downstream electrical RF signals <b>11</b>, and an upstream (US) electro-optic converter <b>10</b>B that converts upstream (US) electrical RF signals <b>13</b> into US optical signals for upstream transmission to the remote CMTS. The fiber-optic node <b>10</b> is coupled via a coaxial cable <b>12</b> to a bidirectional amplifier <b>15</b>, which amplifies the downstream signals <b>11</b> for distribution to first and second groups of homes <b>50</b>A and <b>50</b>B. The downstream RF signals <b>11</b> generated by the downstream optoelectronic converter <b>10</b>A of the fiber node <b>10</b> are distributed to a plurality of end-of-the-line subscribers, or end users, customer premises or homes <b>53</b>, via one or more trunk coaxial cables <b>44</b> and subscriber taps <b>51</b>. At the customer premises <b>53</b>, the DS signals are demodulated using cable modems (not shown). One or more two-way trunk RF amplifiers <b>40</b> may further be provided in each trunk cable <b>44</b> to suitably amplify the upstream and downstream CATV signals on their way to and from the subscriber premises <b>53</b>. The first and second groups of homes <b>50</b>A, <b>50</b>B may send upstream signals <b>31</b>A and <b>31</b>B, respectively, which may be combined by the signal transmitter <b>110</b> into the upstream RF signal <b>13</b> propagating towards the fiber node <b>106</b> for delivering to the remote CMTS at the headend (not shown).

0042A cable network tester <b>100</b> may be connected to the cable network <b>1</b> at a desired test point where the tester can inject a probe signal <b>101</b> into the network in the downstream or upstream direction and receive back from the cable network return signals <b>103</b>. The cable network tester <b>100</b>, which is also referred to herein simply as tester <b>100</b>, may be configured for testing a signal path in the cable network <b>1</b>, including determining the location of a fault in the cable network <b>1</b>.

0043The cable network <b>1</b> may serve a large number of homes <b>53</b>, which may be connected by taps <b>51</b> to a plurality of different cable trunks <b>44</b> at a plurality of different locations. The trunk cables <b>44</b> may be buried into the ground or they may be elevated above the ground on utility poles, or a combination of both. Various faults and defects can occur anywhere in the network <b>1</b>, resulting in a lost or reduced service to the end users <b>53</b>. Many types of defects typically cause an abrupt change in the impedance along a signal propagation path in the network; such defects or faults include kinks, breaks or rust in the coaxial cables, rusty or otherwise imperfect cable connections and connections within various cable equipment along the signal path, etc. These and other types of cable plant faults tend to reflect back RF signals incident upon them and may be discovered using principles of time domain reflectometry (TDR). The TDR typically involves sending a short impulse of a duration τ along a signal transmission path and waiting for any echoes of the pulse that may return back as the result of the impulse reflection from the locations of the abrupt impedance change in the transmission line caused by a fault. The distance to the fault (DTF) l=v·T/2 may then be estimated by measuring the time T elapsed between the sending of the original pulse and the reception of its echo based on an expected value v of the speed of propagation of the TDR pulse along the transmission path. The range resolution δR˜vτ2 of this technique is defined by the pulse duration τ, so that shorter pulses generally yield better resolution.

0044One potential issue that may be associated with using the short-pulse TDR technique in a cable TV network such as the cable network <b>1</b> is that the TDR pulses of a sufficiently high energy may be needed due to relatively high cable losses and to detect low-reflectivity defects, and such pulses may interfere with the cable TV signals at the subscriber premises. Another possible issue associated with using the short-pulse TDR technique in the cable TV network <b>1</b> is that the echoes of the TDR pulse may be difficult to detect in the presence of the US signals <b>31</b>A, <b>31</b>B generated by the end user equipment at <b>53</b>. These issues may be resolved by taking the network <b>1</b> out of operation during the TDR measurement, which is undesirable.

0045In order to address one or more of these issues, tester <b>100</b> may implement an improved TDR technique as described hereinbelow with reference to <figref idref="DRAWINGS">FIGS. 2-9</figref>. The technique includes launching a pulsed probe signal <b>101</b> that is comprised of one or more frequency-chirped probe pulses <b>111</b> into a cable network, and monitoring a return signal <b>103</b> from the network for echoes of these frequency-chirped probe pulses originating a network fault. The one or more frequency-chirped probe pulses <b>111</b> are electrical pulses that are formed of an oscillatory electrical signal which oscillation frequency f is “swept”, i.e. continuously increased or decreased, in time during, each pulse across a pre-defined frequency sweep band f_max, f_min) between a minimum frequency f_minand a maximum frequency f_max. The frequency sweep band (f_max, f_min) is also referred to herein as the probe frequency band. By way of example, tester <b>100</b> may be connected to cable network <b>1</b> in a manner allowing it to inject the frequency-chirped pulsed probe signal <b>101</b> into the downstream direction, and to receive the return signal <b>103</b> propagating in the upstream direction. The return signal <b>103</b> may generally include echoes of the probe signal <b>101</b> reflected from various locations in the network downstream from tester <b>100</b>, and upstream signals generated by the end users. One advantage of using the frequency-chirped probe pulses <b>111</b> for locating faults in an operational cable network is that the pulse energy may be spread over a comparatively longer time period, i.e. the pulse duration τ, which may enable either reducing the power of the probe signal in order to lessen the interference with the downstream signals for the end users, or increasing the signal to noise ratio (SNR) when detecting the reflected echoes, or both. The frequency-chirped probe pulses <b>111</b> may also be referred to herein as the frequency-swept (probe) pulses <b>111</b> or simply as probe pulses <b>111</b>. In one embodiment the frequency f, which may also be referred to herein as the probe signal frequency, may be changing linearly in time from f_minto f_maxduring the duration of the pulse, in which case the probe pulses <b>111</b> may be referred to as linearly frequency modulated (LFM) pulses or as the LFM TDR pulses. In other embodiments, the dependence of the probe signal frequency on time f(t) within each probe pulse <b>111</b> may deviate from linear. An example of the pulsed probe signal <b>101</b> that is composed of a sequence of the frequency-swept probe pulses <b>111</b> of a pulse duration τ and pulse period P is schematically illustrated in <figref idref="DRAWINGS">FIG. 2</figref>.

0046The return signal <b>103</b> may be processed using a pulse compression technique based on the known waveform of the probe pulse <b>111</b>, which enables achieving a higher time-bandwidth product than for an equivalent CW pulsed signal. This results in an impulse signal that combines the higher signal to noise ratio (SNR) of a long duration pulse with the higher range resolution of the short pulse. The range, or distance-to-fault (DTF), resolution δR of this TDR technique may be estimated from the width B=(f_max−f_min) of the probe frequency band: <br /><i>δR=v</i>/(2<i>B</i>)  (1)

0047By way of example, using chirped probe pulses <b>111</b> with B=80 MHz may yield the range (DTF) resolution of about 1.5 m, assuming v˜0.85·c, where c is the speed of light in vacuum.

0048Referring now to <figref idref="DRAWINGS">FIG. 3</figref>, there is schematically illustrated a an exemplary block diagram of the cable network tester <b>100</b> that is configured for implementing an embodiment of the chirped-pulse TDR technique for determining fault locations in the cable network <b>1</b>. The exemplary cable network tester <b>100</b>, hereinafter also referred to as tester <b>100</b>, may include a signal transmitter <b>110</b> coupled to an output cable port <b>118</b>, and a signal receiver <b>120</b> coupled to an input cable port <b>128</b>. The signal transmitter <b>110</b> includes a probe pulse generator <b>112</b> for generating the probe signal <b>101</b>, which may be followed by a DAC <b>114</b> that may in turn be followed by an optional output filter <b>116</b>, which in one embodiment may be a low-pass filter having a bandwidth B. The signal receiver <b>120</b> includes an optional input filter <b>126</b>, which may be for example a low pass filter having the same or different bandwidth as the output filter <b>116</b>, and which may be followed by an ADC <b>124</b>, which output is passed to a matched filter <b>122</b>. One or more memory devices <b>130</b>, hereinafter referred to as memory <b>130</b>, may further be provided and may be operatively coupled to the pulse generator <b>112</b> and the matched filter <b>122</b>. A return signal processor (RSP) <b>150</b> may be coupled to the memory <b>130</b> and configured for processing the return signal <b>103</b> after it passed the matched filter <b>122</b> as described hereinbelow, which may include processing the return signal for presenting to a viewer in a form suitable for extracting ranging information therefrom, identifying locations of possible faults in the cable plant, and estimating the DTF. An output device <b>170</b>, such as a suitable display, that is coupled to the return signal processor <b>150</b> may further be provided for outputting, e.g. displaying, results of the processing.

0049The return signal processor <b>150</b>, pulse generator <b>112</b>, and matched filter <b>122</b> may be embodied using a single dedicated or shared hardware processor or using multiple hardware processors, and/or a combination of software and digital hardware. Examples of hardware processors that may be used to implement blocks <b>112</b>, <b>122</b>, <b>150</b> include digital signal processor (DSP), Application Specific Integrated Circuit (ASIC), Field programmable Gate Array (FPGA), network processor, system on a chip such as an FPGA with integrated ARM or micro processor, complex Programmable Logic Device (CPLD), Erasable programmable logic device (EPLD), Simple programmable logic device (SPLD), or macrocell array. In one exemplary embodiment, a hardware processor implementing the return signal processor <b>150</b>, such as a DSP, a suitable microcontroller, or a general purpose processor, runs a software or firmware program or programs including computer instructions for performing one or more operations described hereinbelow with reference to blocks <b>250</b>, <b>330</b>, <b>340</b>, <b>350</b> in <figref idref="DRAWINGS">FIGS. 4 and 7</figref>; functionality of this software program will become apparent from the description hereinbelow and may include one or more of the following; dividing a saved sample or duration of the return signal in tine-domain slices synchronized to consecutive probe pulses <b>111</b>, averaging the signal slices to obtain a cable response characteristics, and calibrating the cable response characteristics. In one embodiment, this software program is executable by a hardware processor implementing RSP <b>150</b> and is stored in a non-volatile memory (not shown) that is coupled to the hardware processor. In one embodiment, the pulse generator <b>112</b> and the matched filter <b>122</b> may be embodied as hardware logic, for example using an FPGA. The memory <b>130</b> may be embodied, for example, as RAM or as a combination of RAM and a non-volatile memory device.

0050One or more modes of operation of tester <b>100</b> will now be described with reference to <figref idref="DRAWINGS">FIG. 3</figref>, and further with reference to flowcharts shown in <figref idref="DRAWINGS">FIGS. 4 and 7</figref>, which illustrate embodiments of the method for locating faults in a cable network using tester <b>100</b> according to the present disclosure.

0051Turning first to the flowchart of <figref idref="DRAWINGS">FIG. 4</figref> while continuing to refer to <figref idref="DRAWINGS">FIG. 3</figref>, one embodiment of the method may start with an optional step or operation <b>210</b> wherein a technician connects tester <b>100</b> to the cable network <b>1</b> at a suitable location for launching the probe signal <b>101</b> into a section of the network to be tested. This operation may be omitted if tester <b>100</b> is already connected to the network, for example if tester <b>100</b> is installed in the network on a permanent or semi-permanent basis. When present, this step may include connecting tester <b>100</b> to a cable connection point in the network using a suitable Y-type three-port connector <b>160</b>, if tester <b>100</b> has separate output and input ports <b>118</b>, <b>128</b> as illustrated in <figref idref="DRAWINGS">FIG. 3</figref>, and these two ports are separately utilized for the transmission of the pulsed TDR signal <b>101</b> and for receiving the return signal <b>103</b>. The three-port connector <b>160</b> includes a first port <b>161</b>, a second port <b>162</b> and a third port <b>163</b>, and is configured so that a signal received in the first port <b>161</b> is transmitted out of the third port <b>163</b>, and a signal received in the third port <b>163</b> is transmitted out of the second port <b>162</b>. Step <b>210</b> may include connecting the first and second ports <b>161</b>, <b>162</b> of the Y-type connector <b>160</b> to the output and input ports <b>118</b>, <b>128</b> of tester <b>100</b>, respectively, and connecting the third port <b>163</b> of the Y-type connector <b>160</b> to a cable connection point in the cable network <b>1</b>. In one embodiment, the pulsed probe signals received in the first port <b>161</b> may also be coupled into the second port <b>162</b>, which may be utilized in some embodiments for probe pulse—echo synchronization as described hereinbelow. The Y-type connector <b>160</b> may be embodied, for example, as a three-leg summing resistor network of the type schematically illustrated in <figref idref="DRAWINGS">FIG. 6B</figref>.

0052Once the tester is connected to the cable network <b>1</b>, the method proceeds to step or operation <b>220</b> wherein the signal transmitter <b>110</b> of the tester <b>100</b> generates the pulsed probe signal <b>101</b> and launches it into the cable TV network through the output port <b>118</b> of the tester <b>100</b>. The pulsed probe signal <b>101</b> includes at least one probe pulse <b>111</b> during which the probe signal frequency f sweeps across the pre-defined probe frequency band (f_min, f_max), as described hereinabove. Once launched into the cable plant, the probe pulse <b>111</b> propagates through the coaxial cables and installed equipment of the cable network <b>1</b>, experiencing attenuation and reflections from impedance discontinuities in its path, such as those that may occur due to faults or imperfect cable connections. By way of example, <figref idref="DRAWINGS">FIG. 1</figref> indicates the location of a cable fault <b>33</b> where the probe pulses <b>111</b> are partially reflected back towards the tester <b>100</b>. The reflected echo or echoes of the probe pulse <b>111</b> will propagate in the reverse direction, reaching the tester <b>100</b> as the return signal <b>103</b>, with a time delay T relative to the time of the transmission of the probe pulse <b>111</b> that is indicative of the distance to the fault from the tester. At step <b>230</b> the return signal <b>103</b> is received by the signal receiver <b>120</b> of the tester <b>100</b>, where it may be optionally passed through the input filter <b>126</b>. The received return signal <b>103</b> is passed to the matched filter <b>122</b> at step <b>240</b>. The matched filter <b>122</b> is matched to the probe pulse <b>111</b> for compressing the received echoes of the probe pulses <b>111</b> in time, for example as described hereinbelow. At step <b>250</b>, the return signal passed through the matched filter <b>122</b> is analyzed by the signal processor <b>150</b> to identify one or more peaks therein corresponding to the echoes of the probe pulse or pulses <b>111</b> reflected at the fault location <b>33</b> in the cable network.

0053The generation of the probe pulses <b>111</b> and the matched filtering of the return pulses may each be performed either digitally or using suitable analog circuitry. In the embodiment illustrated in <figref idref="DRAWINGS">FIG. 3</figref>, both the initial generation of the probe pulses <b>111</b> and the matched filtering of the echoes is performed in the digital domain using one or more digital processors embodying the probe pulse generator <b>112</b> and the matched filter <b>122</b>, such as for example an FPGA or an ASIC. In the shown embodiment, the pulse generator component <b>112</b> may be in the form of a logic circuit, for example defined in an FPGA, that loads a digital, waveform of the pulse from a binary file saved in memory <b>130</b>, and sends it to the DAC <b>114</b> for convening it into the analogue probe signal <b>103</b>. The return signal <b>103</b> received by the tester <b>100</b> is digitized by the ADC <b>124</b> prior to the matched filtering operation, which in the shown embodiment is performed in the digital domain by a digital logic embodying the matched filter <b>122</b>, which by way of example may be defined in the same FPGA or ASIC or in a different hardware processor.

0054In one embodiment, the digital waveform of the probe pulse <b>111</b> may be generated, for example using a computer code, as a burst of an oscillatory LFM signal, such as for example a sampled sine wave of a desired length, which frequency is changing linearly in time, and then saved in a memory region <b>131</b> of memory <b>130</b> that is readable by the pulse generation logic <b>112</b>. In operation the pulse generation logic <b>122</b> may load the saved burst waveform, scale it as required and output to the DAC <b>114</b> for converting into the chirped probe pulse <b>111</b>. The length and the rate of frequency change of the saved digital waveform of the pulse may be selected for a given sampling rate of the DAC <b>114</b> so as to provide the pre-determined duration τ of the probe pulse and the desired minimum and maximum values f_min, f_maxof the probe pulse frequency. By way of example, f_minmay be equal to zero, and f_maxmay be in tens of MHz or greater, and the digital waveform may be converted to a desired bit depth prior to saving, for example 12 bit full scale value. The maximum frequency f_max, and therefore the bandwidth B=(f_max−f_min) of the probe pulses <b>111</b>, may be limited by the sampling rate of DAC <b>114</b> and the output filter <b>116</b>. By way of example, the sampling rate of DAC <b>114</b> may be 204.8 MHz, which limits f_maxand the pulse bandwidth B to 102.4 MHz. The bandwidth B of the pulsed signal <b>101</b> at the output of tester <b>100</b> may be further limited by the output filter <b>116</b>. In some of the exemplary embodiments described herein f_maxis taken to be equal to about 80 MHz, although one skilled in the art will appreciate that this is not a requirement, and other values of the maximum frequency f_maxof the chirped pulses and of their bandwidth B may also be realized, as supported by the used DAC and the output filter bandwidth. Further by way of example, the duration τ of the probe pulse or burst may be in the range of 2 to 5 microseconds (μs), but may also be outside of this range. Some of possible considerations that may guide the selection of the probe pulse duration τ are described hereinbelow. In one embodiment, the pulse generator <b>112</b> may generate a digital probe signal wherein the digital probe pulse is periodically or aperiodically repeated, with a buffer of zeros of sufficient length placed between the pulses. The length of the buffer spacing between the probe pulses may be selected so as to allow time for the echoes of the probe pulse <b>111</b> to return into the tester unimpeded by the next probe pulse <b>111</b> being transmitted.

0055In one embodiment, the matched filter <b>122</b> is a digital filter that performs, substantially, a cross-correlation of the received signal <b>103</b> with a waveform that substantially reproduces the waveform of the probe pulse <b>111</b>; this cross-correlation may be computed in the digital domain by convolving the matched filter waveform with the digitized return signal, producing peaks at the time points of maximum correlation of the received signal <b>103</b> with the matched filter. As the result, echoes of the probe pulse <b>111</b> are compressed in time and increased in amplitude relative to all other signals that may be present in the return signal <b>103</b>, including the upstream signals from the subscribers. Thus, the use of the chirped probe pulses <b>111</b> in combination with the matched filtering of the returned signals <b>103</b> makes it easier to detect echoes of the probe signal in the presence of upstream TV signals from the subscribers by selectively amplifying the echoes of the probe pulses. The matched filter waveform may be created, for example, by terminating to the ground the third Y-connector port <b>163</b>, which in operation would be connected to the cable under test, and capturing in memory <b>130</b> a copy of a chirped probe pulse <b>111</b> that is generated by the signal transmitter <b>110</b> and received in the input port <b>128</b> from the output port <b>118</b> through the bridge <b>160</b>, after it is digitized by the ADC <b>124</b> without passing through the matched filter <b>122</b>. The captured pulse is substantially the probe pulse <b>111</b> generated by the signal transmitter <b>110</b> that is shaped by the input filter <b>126</b> and possibly transmission responses of other components of the tester frontend. Optionally a Harming or other suitable window may be applied to the captured pulse for sidelobe attenuation as known in the art, and the resulting filter waveform saved into a binary file. An example of the matched filter waveform, with the optional Hanning window applied to it, is illustrated in <figref idref="DRAWINGS">FIG. 5</figref>.

0056In order to determine the time delay T between the launching of the probe pulse <b>111</b> into the cable plant and the return of an echo pulse corresponding thereto, the return signal <b>103</b> received from the cable plant should be synchronized, to transmission of the probe pulses <b>111</b>. In one embodiment, tester <b>100</b> may include an internal clock <b>140</b> that provides a receiver—transmitter time synchronization, and may further include logic for marking the time of the probe pulse transmission in the digitized received signal that is provided to the return signal processor <b>150</b>. In the absence of such internal synchronization mechanism, the desired synchronization may be facilitated by sending a copy of each probe pulse <b>111</b> from the output port <b>118</b> of the signal transmitter <b>110</b> directly into the input port <b>128</b> of the signal receiver <b>120</b> to provide the reference markers from which the delay T may be measured, for example as described hereinbelow.

0057Referring to <figref idref="DRAWINGS">FIG. 6A</figref>, in one embodiment the time synchronization between the pulse transmission and echo reception events may be accomplished using an embodiment of the three-port coupler <b>160</b> that splits the probe signal <b>101</b><i>a </i>received form the output port <b>118</b> of the signal transmitter <b>110</b> in two, sending a copy <b>111</b><i>b </i>of each probe pulse <b>111</b> launched into the cable plant directly into the input port <b>128</b> of the signal receiver <b>120</b>. In this embodiment, the signal <b>103</b><i>a </i>received by the signal receiver <b>120</b> is a sum of the return signal <b>103</b> received from the cable plant and the direct signal <b>101</b><i>b </i>received directly from the signal transmitter <b>110</b>, with the coupler <b>160</b> providing a shortcut between the output and input ports <b>118</b>, <b>128</b> of the tester <b>100</b> circumventing the cable plant. Thus, in this embodiment the signal <b>103</b><i>a </i>received by the signal receiver <b>120</b> of tester <b>100</b> includes both the probe pulse copy <b>111</b><i>b</i>, which is received directly from the output port of the signal transmitter <b>110</b> and may be referred to herein as the direct pulse, and a return pulse <b>113</b>, which is an echo of the probe pulse <b>111</b> resulting from the reflection of the latter at a fault in the cable plant, and which is typically much weaker than the direct pulse <b>111</b><i>b</i>. The return pulse <b>113</b> is also referred to herein as the echo pulse, or as the probe echo, or simply as the echo. The process of synchronization of the pulse transmission and reception events in the tester <b>100</b> may then include i) identifying the position or timing of the direct pulse <b>111</b><i>b </i>in the received signal <b>103</b><i>a</i>, ii) identifying the position or timing of a corresponding echo pulse <b>113</b> in the received signal <b>103</b><i>a</i>, and iii) determining the time delay T therebetween.

0058In one embodiment, tester <b>100</b> may generate a sequence of the chirped probe pulses <b>111</b> as illustrated in <figref idref="DRAWINGS">FIG. 8(<i>a</i>)</figref>, and perform an averaging of their echoes <b>113</b> to further suppress noise and increase the signal to noise ratio (SNR). Note that upstream signals <b>31</b> from the subscribers typically have characteristics of a random signal and therefore may also be suppressed by the averaging. The wait time P between the transmission of consecutive probe pulses <b>111</b> in the probe signal <b>101</b> should exceed a maximum expected value T_maxof the time delay T between the transmission of the probe pulse <b>111</b> and the receptions of its echo <b>113</b> for a desired maximum distance Dmax in the cable plant that tester <b>100</b> may probe. By way of example, for Dmax=1000 feet, the wait time P between consecutive pulses should exceed Tmax=2Dmax/v˜2.4 μs.

0059Referring now to <figref idref="DRAWINGS">FIG. 7</figref>, in one embodiment of the method tester <b>100</b> may implement the following steps or operations when processing the received signal <b>103</b> or <b>103</b><i>a </i>that is received at the input port <b>128</b> of the signal receiver <b>120</b>. First, the received signal is digitized in step <b>310</b> using the ADC <b>124</b>, which converts it into the digitized received signal <b>141</b>. At step <b>320</b>, this digitized received signal <b>141</b> is passed through the matched filter <b>122</b>, which outputs a filtered received signal <b>143</b> wherein any signals that correlate with the probe pulse waveform are compressed in time and amplified in magnitude. <figref idref="DRAWINGS">FIG. 8(<i>b</i>)</figref> illustrates this filtered digital received signal for the embodiment wherein the received signal <b>103</b><i>a </i>includes the direct probe pulses <b>111</b><i>b </i>and their echoes <b>113</b> received from the cable plant. The matched filter operation converts these direct pulses <b>111</b><i>b </i>and echo pulses <b>113</b> into narrow peaks <b>211</b> and <b>213</b>, with the bigger peaks <b>211</b> typically corresponding to the direct pulses <b>111</b><i>b</i>, and the smaller peaks <b>213</b> in-between them corresponding to their echo pulses <b>113</b> received from the cable plant.

0060From the output of the matched filter <b>122</b>, a duration or sample of the filtered received signal <b>143</b> that includes a desired number N≥1 of the direct pulses <b>211</b> may be saved in a memory region <b>135</b> of memory <b>130</b> that is readable by the signal processor <b>150</b>. In one embodiment, the signal processor <b>150</b> may be configured, for example programmed, to process the saved sample of the filtered received signal <b>143</b> as follows. At step <b>330</b>, the saved sample of the filtered signal <b>143</b> is divided in time domain into a plurality of N>1 time-domain signal slices <b>222</b>₁to <b>222</b>_Nthat are synchronized to the sequence of probe pulses <b>111</b>, i.e, each starting at a same position relative to a corresponding direct pulse peak <b>211</b>. In one embodiment each of these signal slices, which are generally referred to herein as slices <b>222</b>, includes one big peak <b>211</b> corresponding to a direct signal <b>101</b><i>b </i>and a certain duration of the received signal <b>103</b><i>a </i>that includes one or more smaller peaks <b>213</b> corresponding to the one or more echoes of the probe signal <b>101</b>. By way of example, each signal slice <b>222</b> may start with one of the direct-pulse peaks <b>211</b> and have the same length or duration, i.e. have the same number of data points. The length, or duration, of each slice <b>222</b> may be selected so as to include echoes reflected anywhere within a desired length Dmax of the cable plant being probed.

0061The RSP <b>150</b> may include logic embodying a suitable algorithm for identifying the big peaks <b>211</b> in the saved sample of the received signal; this identification may be facilitated by ensuring that the direct pulses <b>111</b><i>b </i>received at the input port <b>118</b> exceed in amplitude or height any signal received from the cable plant. In one embodiment, the process of identifying the direct-pulse peaks <b>211</b> in the filtered signal <b>143</b> may first determine a maximum value in the saved sample of the filtered signal, and then find all peaks therein that exceed a certain percentage of the maximum signal value, for example are at least 90% thereof. The positions of the big peaks <b>211</b> in the saved signal sample <b>143</b> provide reference time points for obtaining the desired ranging information, i.e. determining the time delay T between the transmission of the probe pulses <b>111</b> and receiving back their echoes <b>113</b>, from which the DTF may be computed.

0062In some embodiments tester <b>100</b> may implement an internal time synchronization between the transmission and reception events without requiring the sending of a copy of the probe pulses <b>111</b><i>a </i>from the output to the input port of the device. For example, a specific digital timing signal may be added to the output of the matched filter <b>122</b> every time a probe pulse is generated by the pulse generator <b>112</b>, to provide a pulse transmission marker. The starting points of each slice <b>222</b> may then be identified in the saved received signal by the processor <b>150</b> in the absence of the big peaks <b>211</b> therein, with each slice <b>222</b> starting at a same position relative to a marker identifying the instance of transmission of one of the probe pulses <b>111</b>.

0063At step <b>340</b>, the processor <b>150</b> may perform an averaging operation on the time-domain slices <b>222</b> of the return signal to obtain an average signal slice <b>223</b> wherein noise is suppressed and the echo peak or peaks <b>213</b> may be easier to identify. The number of signal slices N used in the averaging may depend on the application, measurements conditions, strength of the back reflections in the cable plant, etc. Generally, the greater is the number of slices N, the better is the noise suppression and the longer is the measurement time and the stronger requirements on the device memory and processing speed. By way of example, the number of signal slices N in the 50-100 range should result in an additional 6-7 bits of DTF resolution, or about 36-42 dB SNR improvement.

0064In one embodiment, the method may include an optional calibration step <b>350</b> which produces a cable plant response characteristics, or an echo response, <b>225</b> wherein the direct-pulse peak <b>211</b> is removed so the presence of the strong direct pulse <b>111</b><i>b </i>wouldn't obscure the much weaker reflected signals <b>103</b> received from the cable plant, and the smaller echo peak or peaks <b>213</b> would be easier to identify. This step may include subtracting from the averaged signal slice <b>223</b> a calibration slice that includes the direct-pulse peak but not the return signal from the cable. This calibration slice may be obtained using a process that is similar to the creation of the matched filter waveform. It may include terminating to the ground the third port <b>163</b> of the three-port bridge <b>160</b>, transmitting a suitably large number of the probe pulses <b>111</b><i>a</i>, and performing the operations <b>310</b>-<b>340</b> outlined hereinabove with reference to <figref idref="DRAWINGS">FIG. 7</figref>. The resulting direct response slice is saved in a memory region <b>133</b> of memory <b>130</b> and used as the calibration slice during normal operation as described hereinabove.

0065Once the calibration slice is subtracted out from the average slice <b>223</b>, the resulting echo signal <b>225</b> may be processed, for example scaled, for presenting to the user. In one embodiment, the echo signal <b>225</b> may be logarithmically scaled, such as using a 20 log 10 (x) operation to obtain a magnitude ranging response in dB. This raging response may be plotted for the user as a function of time or distance along the cable plant, and/or passed for further analysis to a suitable peak detection algorithm that the RSP <b>150</b> may implement.

0066One advantage of using the chirped probe pulses for fault location in a cable network is that, by suitably selecting parameters of the probe pulse signal <b>101</b> such as the pulse width τ, the pulse spacing P and the pulse bandwidth B, the TDR measurements described hereinabove may be performed while the network <b>1</b> is in operation without significantly affecting subscriber services. Specific approaches to selecting at least some of these parameters will be described hereinbelow by way of example with reference to a cable TV network, such as that illustrated in <figref idref="DRAWINGS">FIG. 1</figref>, that implements data services according to Data-Over-Cable Service Interface Specification (DOCSIS), and in particular using upstream and downstream signals specified in DOCSIS 3.0, Annex B. It will be appreciated however that the approaches described hereinbelow can also be used in cable or other wired networks using alternative communication formats, possibly with modifications that would be evident to those skilled in the art based on the current disclosure.

0067In one embodiment, the cable network <b>1</b> may be a digital cable TV network wherein video and other information is transmitted to the end users over a plurality of downstream channels of a pre-defined width B_cheach, with information-carrying signals in each channel modulated with sequences of symbols of a specific modulation format and duration T_s. One or more of these downstream channels may overlap in spectrum with the probe pulses <b>111</b>, which may therefore interfere with the downstream signals in those channels at a subscriber modem when received at the same time. In accordance with one aspect of the present disclosure, the probe signal frequency is swept during each probe pulse over the probe bandwidth B that is greater than the channel bandwidth B_chof the downstream signals <b>11</b>, which reduces the interference time t_iby a factor B_ch/B: <br /><i>t</i>_i<i>=τB</i>_ch<i>/B.</i>  (2)

0068Here the interference time t_i, which may also be referred to herein as the collision time, is the time that the energy of a probe pulse <b>111</b> spends within the downstream channel bandwidth B_ch. By selecting B to be significantly greater than B_ch, for example by a factor of 10 or more, the interference time t_imade quite small, thereby reducing negative effects of the interference on, the subscriber. By way of example, B_ch=6 MHz, B=80 MHz, and the time t_iany particular probe pulse may interfere with a downstream signal in any one downstream channel is shorter than the pulse duration τ by a factor of 0.075. Further by way of example, a probe pulse <b>111</b> of duration τ=2.48 μs may only affect any given downstream channel for a duration t_i=τ·B_ch/B˜0.186 μs, which compares to 0.18655 as symbol duration T, specified for 256 QAM transmission in DOCSIS 3.0.

0069In some embodiments, tester <b>100</b> may be configured, to generate the probe signal <b>101</b> that takes advantage of error correction coding that is typically implemented in digital communications, including digital TV (DTV) signal transmission. The error correction coding used in the downstream signal transmission in DTV networks is typically designed for correcting bursts of errors up to a specific length, i.e. a maximum correctable burst duration T_bsrt, which may depend on a particular error correction coding scheme chosen for a given channel. Accordingly, in one embodiment the probe pulse duration τ and/or the probe pulse bandwidth B is/are selected so that the probe signal frequency f is swept over the channel bandwidth B_chfaster than the maximum correctable burst duration T_brst, i.e. <br /><i>t</i>_i<i>&lt;T</i>_brst,  (3)<br /> which corresponds to choosing the probe pulse duration that satisfices the following condition: <br />τ&lt;<i>T</i>_brst<i>·B/B</i>_ch.  (4)

0070Referring to <figref idref="DRAWINGS">FIG. 9</figref>, the error correction schemes used in DTV networks typically include using a forward error correction (FEC) encoder <b>410</b> followed by an interleaver <b>420</b> at the DTV signal transmitter, and a de-interleaver <b>450</b> followed by a FEC decoder <b>440</b> in the receiver of the subscriber's modem. By way of example, the downstream, signal transmission in DOCSIS 3.0 networks is specified in the International Telecommunication Union (ITU) Recommendation J.83B, “Digital Transmission of Television Signals, Annex B”, which is incorporated herein by reference. This, document specifies the use of either 64-QAM or 256-QAM modulation formats, and a Reed-Solomon encoder/decoder in combination with a variable-depth convolutional (I, J) interleaver in the downstream transmission. Table 1 lists some of the possible values of the number of taps I and the interleaving increment J, along with the corresponding values of the burst protection interval T_brst(third column) and the interleaver latency L=((I−1)·I·J·T_s) (fourth column) for the 64-QAM and 128-QAM modulation formats defined in the J.83B Recommendation.

0071<tables id="TABLE-US-00001" num="00001"><table frame="none" colsep="0" rowsep="0"><tgroup align="left" colsep="0" rowsep="0" cols="4"><colspec colname="1" colwidth="35pt" align="center" /><colspec colname="2" colwidth="42pt" align="center" /><colspec colname="3" colwidth="77pt" align="center" /><colspec colname="4" colwidth="63pt" align="center" /><thead><row><entry namest="1" nameend="4" rowsep="1">TABLE 1</entry></row><row><entry namest="1" nameend="4" align="center" rowsep="1" /></row><row><entry>I (# </entry><entry>J </entry><entry>Burst protection</entry><entry>Latency</entry></row><row><entry>of Taps)</entry><entry>(Increment)</entry><entry>64 QAM/256 QAM</entry><entry>64 QAM/256 QAM</entry></row><row><entry namest="1" nameend="4" align="center" rowsep="1" /></row></thead><tbody valign="top"><row><entry /></row></tbody></tgroup><tgroup align="left" colsep="0" rowsep="0" cols="6"><colspec colname="1" colwidth="35pt" align="char" char="." /><colspec colname="2" colwidth="42pt" align="char" char="." /><colspec colname="3" colwidth="28pt" align="right" /><colspec colname="4" colwidth="49pt" align="left" /><colspec colname="5" colwidth="21pt" align="right" /><colspec colname="6" colwidth="42pt" align="left" /><tbody valign="top"><row><entry>18</entry><entry>16</entry><entry>5.9</entry><entry>μs/4.l μs</entry><entry>0.22</entry><entry>ms/0.15 ms</entry></row><row><entry>16</entry><entry>8</entry><entry>12 </entry><entry>μs/82 μs</entry><entry>0.48 </entry><entry>ms/0.33 ms</entry></row><row><entry>32</entry><entry>4</entry><entry>24 </entry><entry>μs/16 μs</entry><entry>0.98</entry><entry>ms/0.68 ms</entry></row><row><entry>64</entry><entry>1</entry><entry>47 </entry><entry>μs/33 μs</entry><entry>2.0</entry><entry>ms/1.4 ms</entry></row><row><entry>128</entry><entry>1</entry><entry>95 </entry><entry>μs/66 μs</entry><entry>4.0 </entry><entry>ms/2.8 ms</entry></row><row><entry>128</entry><entry>2</entry><entry>190</entry><entry>μs/132 μs</entry><entry>8.0</entry><entry>ms/5.6 ms</entry></row><row><entry>128</entry><entry>8</entry><entry>759 </entry><entry>μs/528 μs</entry><entry>32</entry><entry>ms/22 ms</entry></row><row><entry namest="1" nameend="6" align="center" rowsep="1" /></row></tbody></tgroup></table></tables>

0072By way of example, a DOCSIS 3.0 network may correct any potential errors that may be caused by a collision with a solitary probe pulse <b>111</b> up to about 40-50 μs duration, for the probe pulse bandwidth B=80 MHz and assuming the worst-case burst correction case of 256 QAM and (8,16) interleaver.

0073The burst correction capabilities specified in Table 1 are generally applicable when the error bursts of the specified maximum size happen infrequently. For example, they may not hold with respect to symbol errors that may be caused by a periodic sequence of the probe pulses with the period P that is smaller than the latency L of the interleaver <b>420</b> used in the downstream channel transmission. By way of example, the interleaver latency of the deepest (128,8) interleaver specified by DOCSIS 3.0 is about 32 ms for 6 QAM DOSCIS signals, and about 22 ms for 256 QAM DOSCIS signals having a shorter symbol duration. In order to estimate potential effects of the more frequent probe signal transmission on the subscribers of the cable services, the duration of potential interference t_i=τ·B_ch/B, i.e. the time interval during which any given probe pulse <b>111</b> may overlap in frequency with a downstream channel, may be compared to the duration T_sof one symbol of the downstream DOCSIS signals. The probability of the probe pulses leading to symbol errors at reception may be reduced, for example, by selecting the probe signal parameters τ and/or B so that t_iis significantly smaller than T_s.

0074Furthermore, we found that the risk of unrecoverable symbol errors due to a longer probe pulse may be completely avoided, or at least significantly reduced, by a proper timing of the probe pulses transmission, for example by synchronizing them with the FEC codeword frequency in the downstream channel. Indeed, it can be seen that by selecting t_i≤n·T_s, where n=1, 2, . . . is an integer, no more than (n+1) consecutive downstream symbols may potentially experience an interference with any given probe pulse <b>111</b> at the reception. This interference may not lead to symbol errors after the de-modulation by the user modem if the downstream signal received by the modem is strong enough so that it can be decoded in the presence of the interference from the probe pulse. Furthermore, potential symbol errors caused by this interference may be corrected by the FEC decoder at the subscriber's modem if the number of symbols in each FEC codeword that may be affected by the probe signal <b>101</b> does not exceed the number of symbols m that the FEC decoder is capable of correcting. By way of example, the ITU Recommendation J.83B specifies the use of Reed-Solomon (RS) block (128, 122) code for FEC encoding/decoding, with a 128 symbol long codeword, or coding block, that is capable of correcting up to in m=3 symbol errors per codeword. The same RS code is used for both 64-QAM and 256-QAM signals; the actual duration T_CWof each codeword in the downstream signal is however different for the 64 QAM and 256 QAM signals due to the differing duration of individual symbols T_sfor these two modulation formats.

0075Accordingly, in one embodiment the probe pulses <b>111</b> are generated no more frequently than one probe pulse per one FEC codeword of the downstream channel signal, and so that, during each of the probe pulses <b>111</b>, the probe signal frequency f is swept across the channel bandwidth B_chover a time interval t_ithat is no greater than (m−1) symbol intervals T_sof the downstream digital signal. This correspond to selecting the bandwidth B and/or the probe pulse duration τ so that <br /><i>t</i>_i<i>=τ·B</i>_ch<i>/B</i>≤(<i>m−</i>1)·<i>Ts,</i>  (5)<br />or<br />τ≤(<i>m−</i>1)·<i>T</i>_s<i>·B/B</i>_ch  (6)

0076By way of example for m=3, B_ch=6 MHz, B=80 MHz, and T_s=0.187 μs (256 QAM signals), condition (6) yields that the duration of each probe signal <b>103</b> should not exceed about 4.97 μs. Further by way of example, the probe pulses are generated with the probe pulse period P not exceeding the RS codeword length or duration T_CW, which is about 47.756 μs for the 256 QAM DOCSIS signal.

0077An analysis of the operation of the DOCSIS error correction schemes revealed that the convolutional de-interleaver specified by DOCSIS 3.0 may group together into a single codeword downstream symbols that experience collisions with different probe pulses even when the period of probe pulse generation P somewhat exceeds the codeword duration T_cd, but is smaller than the interleaver latency. However, this grouping does not occur if P=T_cd, i.e. if the generation of the probe pulses is synchronized with the codeword frequency F_cd=1/T_cdin the downstream channel. Accordingly, in one embodiment tester <b>100</b> generates the sequence of the probe signals <b>103</b> that is synchronized to the codeword frequency F_cdin the downstream digital signal.

0078A complication may arise when two different downstream channels that lie within the frequency band swept by the probe signal <b>101</b> utilize two different transmission formats that have differing symbol rates and therefore differing symbol and codeword durations. Accordingly, in one embodiment tester <b>100</b> generates the sequence of probe pulses that is synchronized to the codeword frequency for one of the two transmission formats that has a lower noise tolerance. By way of example, DOCSIS 3.0 specifies symbol rate of 5.056941 MHZ for 64 QAM signals, and a slightly higher symbol rate of 5360537 MHz for 256 QAM signals. The codeword rates for these downstream signals are approximately 19.7 KHz and 20.9 KHz, respectively. However, the noise tolerance of the 256 QAM signals are about 6 dB lower than that of the 64 QAM signals. Accordingly, in this exemplary embodiment the probe pulses may be generated at the 2.5 QAM codeword rate of about 20.9 KHz, or at any higher-order harmonics thereof.

0079The aforedescribed method for locating faults in the cable network may be implemented, for example, in a multi-purpose portable cable network tester that may be used by a technician in the field, and that may also implement other network test functions. For example, in embodiments wherein the probe pulse generator <b>112</b> and the matched filter <b>122</b> are implemented using a programmable hardware logic device such as an FPGA and the RSP <b>150</b> is implemented using a hardware processor such as a DSP executing software instructions, the same FPGA and the hardware processor may be shared to implement other network tests, such as for example those disclosed in U.S. Patent Documents No. 2015/0009795, 2015/0020129, and 2015/0009340, which are incorporated, herein by reference. Implementing the aforedescribed TDR measurement technique in a portable cable network tester that can also perform measurements on upstream and downstream DTV signals may have an additional advantage of enabling to adjust the TDR parameters, such as the pulse probe power, duration and periodicity, in dependence on the DTV signals that are present in the network, so as to facilitate non-interfering TDR. For example, in one embodiment step <b>220</b> of the TDR measurement illustrated in <figref idref="DRAWINGS">FIG. 3</figref> may be preceded by using the tester <b>100</b> to measure the power of the DS network signals <b>11</b> at the tester location in the DS channels that lie within, or overlap with, the frequency sweep (f_min, f_max) of the probe signal <b>101</b>, and adjusting one of the probe pulse power, the probe pulse duration τ, and the probe pulse period P in dependence on the measurements. For example, the probe pulse duration τ may be increased to gain SNR at the receiver <b>120</b>. In one embodiment, tester <b>100</b> may perform symbol timing measurements in the DS transmission to ensure that the TDR probe pulse transmission is synchronized with the interleaver codewords.

0080The above-described exemplary embodiments are intended to be illustrative in all respects, rather than restrictive, of the present disclosure. Thus the present disclosure is capable of many variations in detailed implementation that can be derived from the description contained herein by a person skilled in the art. All such variations and modifications are considered to be within the scope and spirit of the present disclosure as defined by the following claims. All statements herein reciting principles, aspects, and embodiments of the disclosure, as well as specific examples thereof, are intended to encompass both structural and functional equivalents thereof. Additionally, it is intended that such equivalents include both currently known equivalents as well as equivalents developed in the future, i.e., any elements developed that perform the same function, regardless of structure. Thus, for example, it will be appreciated that block diagrams herein can represent conceptual views of illustrative circuitry embodying the principles of the technology. Similarly, it will be appreciated that any flow charts, state transition diagrams, pseudocode, and the like represent various processes which may be substantially represented in computer readable medium and so executed by a computer or processor, whether or not such computer or processor is explicitly shown.

0081The functions of the various elements including functional blocks labeled or described as “processors” or “controllers” may be provided through the use of dedicated hardware as well as hardware capable of executing software in association with appropriate software. When provided by a processor, the functions may be provided by a single dedicated processor, by a single shared processor, or by a plurality of individual processors, some of which may be shared or distributed. Moreover, explicit use of the term “processor” or “controller” should not be construed to refer exclusively to hardware capable of executing software, and may include, without limitation, digital signal processor (DSP) hardware, read only memory (ROM) for storing software, random access memory (RAM), and non-volatile storage.

0082Thus, the present disclosure is not to be limited in scope by the specific embodiments described herein. Indeed, other various embodiments of and modifications to the present disclosure, in addition to those described herein, will be apparent to those of ordinary skill in the art from the foregoing description and accompanying drawings, and such other embodiments and modifications are intended to fall within the scope of the present disclosure. Further, although the present disclosure has been described herein in the context of a particular implementation in a particular environment for a particular purpose, those of ordinary skill in the art will recognize that its usefulness is not limited thereto and that the present disclosure may be beneficially implemented in any number of environments for any number of purposes.