Measuring baseline vectors using signals from gps satellites

(54) NAME Method and systems for obtaining position-related data using satellites (56) QUOTE PUBLICATIONS: US A 4 368 469 (57) SUMMARY: A method for deriving position-related data from signals modulated with mutually orthogonal codes and sent simultaneously with the same frequency of each of several satellites. This is done independently of externally obtained knowledge of the information content of the corresponding modulation codes. The method comprises: obtaining from the satellites (GPS-L, GPS-2) a first composition of overlapping scattering spectrum signals with an upwardly acting, radiating antenna located at a first mark (SM-1); reconstructing the first composition of spread spectrum signals to form a second composition, which at the same time includes a plurality of reconstructed components which correspond to the single code modulated signals in the signals received from the satellites; and providing projections on the frequencies of the signals from the satellites to the second composition to obtain data therefrom.

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Background of the invention. 460 685

The present invention relates generally to a method and a system for determining position by radio, and more particularly to a method and a system for measuring the baseline vector between a couple of points, e.g. measuring points, on the ground by radio interferometry using radio signals, transmitted from orbiting satellites.

Some systems for determining position by radio utilize the alignment of the radiation pattern of a transmitting and receiving antenna. Other systems, including the system of the invention, are not based on the alignment of any antenna. The present invention belongs to the general class of systems, wherein the position of a receiving antenna is determined by measuring the difference between the phases or group delays or by both of signals arriving from two or more different transmitting antennas, the positions of which are already known. If two t 'transmission sources are synchronized or if the deviation from synchronization of two transmitters is known separately, then a measurement at the receiver site determines the difference between the group delays of the signals arriving from the two sources that the receiver is located in three dimensions, on a special orbital hyperboloid, the focal points of which are the positions of the transmitters. If similar measurements at the same receiving point for signals are combined from several different, appropriately positioned transmitters, then the reception position can be determined only from the intersection of the corresponding hyperboloids.

Methods for determining relative locations of different locations, one with respect to the other, from measurements of phase or group delay differences between radio signals received simultaneously at these sites are also known in the art and are collectively referred to as methods of geodesy by radio interferometry. The antennas at the individual locations are considered to form an interferometer and the relative position vector which extends from one antenna to the other referred to as the baseline vector of the interferometer. The baseline or relative position vector between two antennas can usually be determined with less uncertainty than the position of each individual antenna, since many potential sources of error tend to affect the measurements at the two antennas almost equally and therefore tend to eliminate the same when differences between the two are calculated.

2 * - * · * «·. 460 685 antennas. The microwave radio interferom geometry technique is known to provide an unparalleled combination of accuracy, speed and width for the determination of relative position vectors or baseline vectors interferometers. Such a determination can be based on measurements of either the group delay difference or phase difference, or of both differences between the signals received at both ends of the baseline vector. Phase measurements are in themselves more accurate than group delay measurements, but the interpretation of phase measurements is more complicated due to their real overall cycle ambiguity. A general discussion of interferometric measurement procedures and associated interpretation problems is given in an article entitled Radio Astrometry, found in Annual Reviews of Astronomy and Astrophysics, Volume 14 (1976), pages 197-214 by Charles C. Counselman III. A large collection of relevant technical documents can be found in Conference Publication 2115 of the National Aeronautics and Space Administration entitled Radio Interferometry Techniques for Geodesy. Radio interferometry geodesy has been practicing radio signals emitted by various sources, including natural sources, such as quasars, and artificial sources, such as satellites under the Navstar Global Positioning System (GPS).

As is known, there are currently about six GPS satellites orbiting the Earth. The orbits of the satellites can be determined with an accuracy of approximately 2 meters. These satellites transmit radio signals with wavelengths close to 19.0 cm and also 24.4 cm. Provided that the overall cycle ambiguities of interferometric phase observations of these signals can be properly analyzed, the baseline vector extending from one antenna to the other can be determined interferometrically with much less uncertainty than the wavelengths of the GPS transmissions. Determinations of three baselines, each with a length of the order of 100 m, by means of interferometric phase measurements of GPS signals were found to have been accurate within approximately 1 cm according to a report published in Eos (Transactions of the American Geopysical Union), volume 62, page 260, April 28, 1981, by Charles C. Conselman III,

SA Gourevitch, RW King, TA Herring, II Schapiro, RL Greenspan, AEE Rogers, AR Whitney and RJ Cappallo. The method used in these interferometric baseline determinations was based on the known technique of direct cross-correlation at a central location of the signals received separately at the two ends of each baseline.

U.S. Patent No. 4,170,776 describes a system for measuring changes in a baseline vector between a pair of locations on the ground using signals transmitted from the GPS satellites, the radio signals received at each location being accurately timed and then transmitted over telephone lines to a central location, where a phase comparison in almost real time is done by cross-correlation of the two groups of signals. The system illustrated in the patent comprises dish reflector-type receiver antennas. Since the radio flow density of a GPS signal is small relative to the background noise level and because the bandwidth of a GPS signal far exceeds the bandwidth of a telephone line, the signal to noise ratio of the power transmitted through the telephone line from each location is small. It is largely for the purpose of increasing this signal-to-noise ratio to a useful level that dish-type antennas with large collection areas are used in this system. Another important reason for using such antennas is that they are directional, so that signals transmitted to the antenna other than directly from the desired source are rejected.

Systems for measuring baseline vectors using other kinds of signals from orbiting satellites are also known.

An article titled Miniature Interferometer Terminals for Earth Surveying (MITES), which appears in Bulletin Geodesique, Volume 53 (1979), pages 139-193, by Charles C. Conselman III and Irwin I. Shapiro, describes a proposed measurement system of baseline vectors using multifrequency radio signals, which could be transmitted from orbiting satellites, and in this system the phases of the received signals are determined separately at each end of the baseline. Thus, the signal received at one location is not cross-correlated with the signal received at another location for determining the phase difference between the two signals. To solve the fixed ambiguity, the MITES system relies on the combination of measurements at a group of up to 10 frequencies, which are suitably spread between one and two GHz. Unfortunately, there are

460 685 there are not currently, as far as is known, some orbiting satellites, which emit similar signals.

Relative position measurement systems using signals transmitted from sources other than artificial satellites are also known. An example of such a system using a lunar transmission is also disclosed in US Patent No. 4-170,776.

Systems for measuring either a single position or a relative location using signals from sources other than orbiting satellites are also known. For example, an article by WO Henry, entitled Some Developments in Loran, which appears in the Journal of Geophysical Research, volume 65, pages 506-513, February 1960, describes a system for determining a location (e.g., for a vessel at sea) using signals from ground-based (stationary) transmitters. The system, known as the Loran-C Navigation System, uses several thousands of kilometers of synchronized transmitters stationed on the Earth's surface, all transmitters using the same carrier frequency, 100 kHz, and each transmitter being amplitude modulated with a unique, periodic pattern. of pulses. This pattern, which includes character reversals for the amplitude, allows the receiver to distinguish between signals from different transmitters. A suitable combination of observations on more than one pair of transmitters can provide a determination of the receiver's location on the earth's surface.

Another example of a system of this type is the Omega system described in an article by Pierce entitled Omega, which appears in IEEE Ttansactions on Aerospace and Electronic Systems, volume AES-1, no. 3, pages 206-215, December 1965. In the Omega system, the phase differences of the received signals are measured instead of the group delay in principle as in the Loran-C system. While the frequencies used in both the Loran-C system and the Omega system are very low, the accuracy of positioning with these systems is very poor compared to the mentioned satellite systems.

The prior art also includes other methods for determining position and relative position by Global Positioning

System. The standard method, for example described in an article in Navigation, Volume 25, no. 2, (1978) pages 121-146 by JJ Spilker Jr. and further described in several other λαπ 685 articles, appearing in the same edition of this publication, is based on measurements of the differences between the group delays or times for receiving the coded modulation of the GPS signals. In principle, the method is a hyperbolic alignment method and it is essentially similar to the Hasom method, according to Loran. The bandwidth of the GPS modulation of approximately 10 MHz limits the accuracy of group delay measurement and consequently location determination by the standard method to several tens of centimeters. An accuracy of the order of magnitude

1 cm is potentially available through the use of carrier phase measurements, as described, for example, in an article by JD Bossier, CM Goad and PL Bender entitled Using the Global Positioning System for Geodetic Positioning, which appears in Bulletin Geodesique, volume 54, no. 4, page 553 (1980).

However, any published method for using the GPS carrier phase determination for position determination has the disadvantage that it requires knowledge and use of encryptable code modulation, or that it requires cross-correlation of signals received at different locations, or that it requires the use of large antennas to increase the signal-to-noise ratio of the received signal and suppress interference from reflected signals, or otherwise have more than one of these disadvantages. The invention has none of these drawbacks.

Specifically, the present invention does not require any recognition of the codes modulating the GPS carriers, it does not require cross-correlation of a signal received at one location with a signal received at another location, and it does not require the use of a large or 'precisely targeted receiver antenna.

Summary of the Invention.

An object of this invention is to provide a method and a system for determining position by radio. Another object of the invention is to provide a method and a system for measuring baseline vector between a couple of points using radio interferometry.

Yet another object of the invention is to provide a method and a system for determining the baseline vector between a few points on the earth, e.g. measuring marks, using dual sided radio signals and sub 40 printed carrier and of the type transmitted from the earth

460 685 orbiting satellites according to the Global Positioning System.

A further object of the invention is to provide a method and a system for determining the baseline vector between a pair of measurement marks using radio signals from orbiting satellites according to the Global Positioning System, the determination comprising measuring the phases of the carrier implicitly received at each measurement mark.

Yet another object of the invention is to provide a technique for processing phase information obtained at two locations on earth from radio signals received from different directions to determine relative position.

Another object of the invention is to provide a method and a system for measuring the effects and carrier phases of the radio signals received from satellites according to the Global Positioning System without knowledge of the coded signals, which in these transmitters modulate the carrier waves.

Yet another object of the invention is to provide a method and a system for determining the baseline vector between two points by measuring also the phases of radio signals received at each point, without cross-correlating the signal received at one point with the signal , received at the second point, without registering the signal received at any of the points and without otherwise transmitting a signal from one point to the other or from both points to a common location. '

Yet another object of the invention is to provide a method and system for determining radio position without requiring the use of a directional antenna.

The method of measuring a baseline vector between a pair of points on the earth by radio interferometry utilizing radio signals transmitted by GPS satellites according to the principles of the present invention comprises measuring the

ς. The implicit carrier phases of the signals received from the satellites at each end of the baseline and then joint processing of the phase information from the two locations for determining the baseline vector. The system for measuring a baseline vector between a few points on the earth by radio interferometry utilizing radio signals transmitted so that GPS satellites according to the principles of the present invention comprise a pair of interferometer field terminals, wherein a

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460 The 685 interferometer field terminal is intended to be located at each point, each interferometer field terminal comprising an antenna, a separator for upper and lower sidebands, a plurality of correlators and numeric oscillators, and a field terminal computer.

Brief description of the drawing.

In the drawing, like reference numerals refer to like parts.

Fig. 1 shows a system for determining a baseline vector by radio interferometry with GPS satellites according to the principles of the invention.

Fig. 1 is a view15

Fig. 2 shows a block diagram of one of the interferometer field terminals located.

the

Fig. 3 shows the antenna unit ..

Fig. 4 shows the receiver unit.

Fig. 5 shows a one in block diagram block diagram block diagram digital electronics unit.

Fig. 6 shows a block diagram of the signal conditioning device.

FIG. 7 is a block diagram showing a feed of the feed of the feed of that of FIG.

Fig. Fig. Fig. 3 illustrated illustrate illustrate illustrate the correlator modules one of the correlator unit illustrated in Fig. 5.

Fig. 8 is a block diagram of the oscillator modules of the oscillator unit of Fig. 5.

Fig. 9 shows a block diagram of the field terminal computer.

Detailed description of preferred embodiments.

The present invention is directed to a technique for measuring the baseline vector between a couple of points, e.g. measurement marks, on the ground by radio interferometry using dual-band radio signals and suppressed carrier, transmitted by orbiting satellites according to NAVSTAR Global Positioning Systems (GPS). The technique involves measuring the phases of the carrier implicitly in the signals received at each location, and then processing the obtained phase information at both locations for determining the baseline vector. An advantage of this technique is that it measures the carrier phases without reference to knowledge of the coded signals, which for one of the numerical illustrated numeric for the illustrated in FIG.

460 685. · Used in the satellites to modulate the carrier waves. Another advantage is that the technology does not require transmission of the received. the signals, neither in real time nor through the transfer of registrations, from two places to, c'tt common place ·. Another advantage is that the technology does not require the use of large or well-directed antennas. Yttcrlig.arc_k It is also advantageous that the technology is relatively immune to errors caused by, scattering and reflections of radio waves occurring near the receiving antennas.

While the invention below will be described in particular for use with GPS satellites, it will be appreciated that certain pages thereof are not only limited to use with such satellites but may be used in conjunction with signals received from other sources:

As is known, satellites pass through the NAVSTAR Global Positioning System (GPS) around the Earth at an altitude of approximately 20,000 kilometers and transmit signals in a frequency band centered at 1575.42 MHz, known as, such as the L1 band. , and signals in a secondary band, centered at 1227.60 MHz, which is known as the L2 band. The signals are modulated so that almost symmetrical upper and lower sidebands are generated with the carrier completely suppressed.

For each band, the signal from a given satellite at a given location as a function of time can be considered to be in the form:

s (t) = m (t) cos (2πίθί + φ) + n (t) sinfΣπίθΐ + φ) where m (t) and n (t) are modulation functions, each real-time function; wherein f is the nominal carrier frequency equal to 1,557.42 MHz for the L1 band and 1227.60 MHz for the L2 band; and wherein φ is the received waveguide phase, in radians, which is unknown and to be determined.

Each of the modulation functions, m- (t) and n (t), is a zero-mean pseudo-random time function. The two functions are mutually orthogonal. Each of the functions used for the modulation of the L1 carrier for any satellite is also orthogonal to the opposite function used for each other satellite, although the same m (t) - or The n (t) function or both functions can be used for a given satellite to modulate both the L1 and L2 carriers. 'The bandwidths of the two functions, m (t) and n (t), differ from one another

460 685 factor of exactly 10, where m (t) has the narrower and n (t) the wider bandwidth. Usually, at the L1 band, both the m (t) and n (t) signal components and the L2 band, only the n (t) component occurs, with the m (t) function set to zero or disconnected. The power spectral density of m (t), which corresponds to the inodulation signal known in the GPS literature as the blank / acquisition code, is proportional to its function.²(7rF / 1, 023 MHz) (rcF / 1,023 MHz.)² where F represents the modulation frequency. This feature has a half width at half maximum of approximately 450 kHz. Thus, the function value is approximately 0.5 for F = in 450 kHz, while the value is one for F = 0. The power spectral density of 'n (t) corresponding to the modulation signal known in the GPS literature as the exact code or The P code, is proportional to its² (ttF / 1 0.23 MHz) (kF / 10.23 MHz)²

Thus, half the width at half maximum of the power spectral density for n (t) is approximately 4.5 MHz.

For the L1 band, the 1575.42 MHz signal, the squared mean of n (t) is usually equal to half the value of m (t); thus, &lt;n²(t)&gt; = 0.5 &lt;m²(T)&gt;.

(It is possible for a GPS satellite to be operated in extra ordinary modes, where the ratio of the square averages or power ratio 30 deviates from 0.5; in particular, a value of 0 is possible.)

Thus, the ratio of the power spectral density of n (t) to m (t) is usually equal to approximately 0.5: 10 = 0.05 for a value of F close to zero, so that if a bandpass filter adjusted to the spectrum of m (t) , centered on the L1 carrier frequency, approximately 90 percent of the power contained in this filter's output will be derived from the m (t) signal component and less than 10 percent will be derived from the n (t) component. Therefore, to simplify the remainder of this description, it will be assumed that the GPS-L140 signal has no n (t) component and that it has the

460 685'

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simpler form:

tί s (t) = m (t) cos (2πίθΐ + φ).

In general, the received carrier phase, φ, is a slowly varying function of time, so that the actual received / carrier frequency is obtained by the algebraic sum:

f = f_Q + (2π)^_1 / / dt), where f is the nominal carrier frequency and άφΜΐ is the time derivative of φ. By slowly varying is meant that the expression (2ir) 1 (d0 / dt) is very small in comparison with f and with the bandwidth of mft ^ The main reason for the time variation of φ is Doppler offset, which can cause f to deviate from f by plus or minus up to approximately 4.5 kHz.

The received signal s (t) does not contain any discrete spectral power component at the carrier frequency, since the mean of m (t) is zero. Thus, the carrier is completely suppressed and the power spectral density function of the L1 signal s (t) is equal to the power spectral density function of the modulation m (t), converted from baseband to the received carrier frequency

f. Since m (t) is a real value time function, its power spectral density is an evenly symmetric function of frequency. Thus, the power spectral density of the signal s (t) has uniform symmetry with respect to carrier frequency f and is said to be a double sideband spectrum. The portion of this power spectrum corresponding to frequencies higher than f is referred to as the upper sideband; the part that corresponds to lower frequencies is the lower sideband. (The slight asymmetry, at most approximately 3 parts of 10 6, between the upper and lower sidebands due to the Doppler stretch of the signal is not significant here.)

According to the present invention, an antenna is provided at each end of a baseline vector. The signals received by each antenna are separated into upper and lower sideband components.

These separate components are filtered, converted into digital one-bit form and multiplied with each other. Their product is digitally analyzed by correlation with the square output signals of a local oscillator to determine the power and phase relative to the local oscillator of the carrier, which is implicit in the dual sideband signal received from each satellite.

460 685

Doppler displacement differences are used to distinguish different satellites' carriers. Thus, the effects and carrier phases of the signals from several satellites are simultaneously measured and numerical data representing the measurement results, S is obtained at each measurement mark. The measurements are performed in real time at each mark without reference to signals received at any other location and without knowledge of any of the coded signals modulating the GPS carrier waves. Data from the measurements taken simultaneously but independently at two measurement marks, once per second for a period of time of sufficient duration, e.g. for approximately 5000 seconds, is then processed together to determine the baseline vector extending from one tag to the other. Two methods of treatment are shown. In both methods, a triadic function, which is .15, is a function of the measured data and of an experimental value b for the baseline value. The vector space for 6 is systematically examined

A to obtain the unique value of b, which maximizes the calculated function. This value of 6 is considered to be the desired determination of the unknown baseline vector

Referring to Figure 1, a system for determining a baseline vector É of the present invention is illustrated. The baseline vector &amp;, which is also sometimes referred to below by the term baseline, is the relative position vector of one measurement mark SM-2 with respect to another measurement mark SM-1. The baseline 25 extends from the measurement mark SM-1, which is at the beginning or at one end of the baseline, to the measurement mark SM-2, which is at the end of or at the other end of the baseline. The system 11 comprises two rationally operating interferometer field terminals 13-1 and 13-2, wherein a terminal is located at each end of the baseline, and a computer which can be structurally and functionally included and form part of one of the terminals 13 or which may form a separate unit 15, as shown.

The system, for its normal operation, requires some numerical data from external sources. It also requires some means for transmitting numerical data between the computer 15 and each term 13 before and after or (as the case may be) during baseline measurements.

Before measurements to determine the baseline are started, 40 data are input from a first data memory .17, which represents the circuitry.

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460 685. The 12 orbits for a plurality of GPS satellites, two of which, identified as GPS-1 and GPS-2, are illustrated for illustrative purposes, in the computer 15 along with approximate data representing the locations of the measurement marks SM-1 and SM-2. obtained from a second data memory 19. The latter data, for example, could represent the measurement mark sites with an accuracy of a few kilometers. From this satellite or measurement station data, the computer 15 in tabular form as a function of time generates a prediction of the Doppler frequency offset that the 1575.42 MHz signal transmitted by each GPS satellite will receive when it is received at each measurement mark. . The computer 15 also generates a table prediction of the power level of the signal to be received from each satellite at each mark. The predicted effect is zero if the satellite will be below the horizon; and it is a function of the predicted elevation angle of the satellite above the horizon, depending on the angle dependence of the gain of a receiver antenna (at the mark) and usually to a lesser extent of the transmitting antenna [on the satellite]. The tables for predicted frequency offsets and effects over a period of time, comprising the anticipated measurements for all GPS satellites expected to be visible at each measurement mark, are now transmitted to and inserted into memory by some known means, such as by telephone or radio telephone link. of a smaller computer, which is located in the special interferometer field terminal 13, which will be placed or which may already have been placed at the measuring mark. Alternatively, the frequency and power prediction tables can be generated by the computer in the interferometer field terminal.

The Doppler frequency predictions are calculated according to formulas which are well known in the art. The magnitude of the errors in these predictions is of the order of 1 Hz per kilometer error in the assumed position of the measurement mark. The additional error in frequency prediction due to errors in extrapolation of the satellite orbit is normally of the order of 1 Hz or less for predictions made, at least one day in advance. Frequency prediction errors of up to several Hz are tolerable within the scope of the invention. The predictions for the effect received need not be very accurate; errors of several decibels could be tolerated, since these predictions were not used for any very critical purpose. They mainly serve to enable

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46Π 6 «5 for the field terminal computer to check whether the desired signal_?no false signal is received. At some possible expense of reliability, the power prediction tables could be eliminated.

An interferometer field terminal 13, which has been placed at a measurement mark, now receives the 1575.42-MHz signals from a plurality of satellites, up to 7, but in no case less than two, simultaneously. For a precise determination of the baseline to be obtained, it is essential that the terminals at both ends of the baseline observe the satellites simultaneously.

Electronic circuits (will be described below), in each terminal, the received signals separate into upper and lower sideband components and, using the predictions - analyze the Doppler frequency offset these sideband components' to determine the carrier power and phase implicitly in it from each satellite received signal. Data from these power and phase determinations are stored in the field terminal and returned to the central computer 15 months by some conventional means.

Data from the two interferometer field terminals 13-1 and

13-2 must be processed together to obtain a precise determination of the baseline vector.

It should be noted that means for communication or transmission of long distance data are not required for the operation of this system. The terminals 13-1 and 13-2 can be physically transported to the same location as the computer 15 and where the prediction tables can be transferred from the computer 15 to the terminals 13. Then the terminals 13, which in their memories contain the tables, can be transferred to the measurement marks SM-1 and SM -2, where the satellites are observed. Upon completion of these observations, the terminals 13 can be returned to the location of the computer 15, where carrier phase data can be transferred from both terminals to the computer for processing.

Referring now to Fig. 2, the main components of an interferometer terminal 13, also referred to as field terminal, are now illustrated. Each field terminal Ί3 has an antenna unit 21 which is connected to an electronic unit 23 by means of a coaxial cable 25.

Each antenna unit 21 includes an antenna 27 and a preamplifier unit 29. The antenna is located on the measuring mark

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460

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the heat when setting its phase center does not exceed a few millimeters.

The antenna 27 receives the 1575.42 MHz radio signals transmitted by the GPS satellites. The received signals are amplified by the preamplifier 27 and fed via the coaxial cable 25 to a receiver unit 31 of the electronic unit 23, the receiver unit 31 comprising a sideband separator 33, a receiver power circuit 34 and an oscillator circuit 35.

The upper sideband portion of the signals, including the portion of the signals received from all satellites in combination, which occupies a range of radio frequencies which

extending upwards from 1575.42 MHz, is tapped into the sideband separator from the lower sideband portion, which corresponds to radio frequencies below 1575.42 MHz. To effect this separation, the sideband separator 33 uses a 1575.42-MHz reference signal: fed by the oscillator circuit 35.

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Receiver unit 31 supplies three signals in analog form to a digital electronics unit 37. One analog signal, denoted u (t), represents the upper sideband component of the received radio frequency signals, converted to baseband. The second analog signal, designated δ (t), also represents the lower sideband component 25 converted to baseband. Each of these two signals contains contributions from all visible satellites. The third signal transmitted to the digital electronics unit 37 is a sinusoidal signal having a frequency of 5,115 MHz which is the output of a self-oscillating, stable quartz crystal oscillator in the oscillator circuit 35. The output of this oscillator frequency is multiplied by a determined integer frequency

308 in the oscillator unit to obtain the reference frequency

1575.42 MHz used by the sideband separator. The accuracy of the frequencies 9 generated by the oscillator unit 35 is normally approximately a part of 10, although an accuracy Q of a part of 10 could be tolerated.

In the digital electronics unit 37, each of the three analog inputs is converted into a digital-logic signal.

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The digital signals are processed under the control of a field terminal computer 39 for generating carrier power and phase data. The digital electronics unit 37 is connected to the field terminal computer 39 by means of a bidirectional data bus 41. The field terminal computer 39 may be a digital equipment of type Digital Equipment Corporation (DEC) model LSI-11/2; in this case, the data bus 41 may be the DEC-Q bus.

Carrier phase data is stored in the memory of the field terminal computer 39 until it is desirable to transmit said data to the central computer 15 for processing. As indicated, the central computer may be eliminated and processing performed in one of the field terminal computers 39. Phase data may also be printed by the field computer 39 on a data storage medium, e.g. a magnetic tape, cassette or disc (not shown). The data can also be transmitted via direct electrical connection or via a modem and telephone connection or via any other standard means.

Referring now to Figure 3, the components of the antenna unit 21 are now shown in greater detail. The unit 21 comprises an antenna 27 which, as mentioned, is so constructed that its phase center can be accurately aligned with respect to the measurement mark.

The 1575.42-MHz radio signals received by the antenna 27 are supplied to the preamplifier circuit 29, whose function is to raise the signals power level sufficiently to eliminate the attenuation of the coaxial cable 25 which connects the antenna unit 21 to the receiver unit 25, and to eliminate the background noise. generated in the input amplifier of the receiver unit 31.

In the preamplifier circuit 29, the signals received from the antenna are first filtered by a bandpass filter 43 having a bandwidth of approximately 50 MHz and centered at 1575.42 MHz.

The function of the filter 43 is to prevent overloading of the receiver unit 31 by strong false signals which may occur outside the GPS signal band. The output of the bandpass filter 43 is fed to a passive diode limiter 45, which can serve to protect a low noise amplifier 47 from being burned by possibly very strong signals, e.g. those which could be transmitted by nearby high power radar systems. The low noise amplifier 47 is a gallium arsenide field effect transistor (FET) amplifier of standard type with a noise factor of approximately 2 db.

DC power for the low noise amplifier is supplied via coaxial460 ίο

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^Jkaoel η 25, which is connected to the preamplifier unit 29, from the receiver unit 31, via a radio frequency choke 49 and a voltage regulator 51.

Referring to Fig. 4, the components of the receiver unit 31 are shown in greater detail. The receiver unit 31 includes a receiver power circuit 34, a sideband separator 33, and an oscillator circuit 35. The receiver power circuit 34 provides direct current for the operation of the oscillator circuit 35, the sideband separator 33 and, via the coaxial cable 25, the low noise amplifier 47 in the antenna 47. Oscillator circuit 35 provides a reference frequency of 1575.42 MHz for the sideband separator 33 and a reference frequency of 5,115 MHz for the digital electronics unit 37. The sideband separator 33 separates the signals received in a radio frequency band which is centered at 1575.42 MHz and extends upward. down from this frequency, in separate upper and lower sideband components with baseband.

Receiver power circuit 34 contains regulated DC power sources 61 and additionally an accumulator battery 63. Battery 63 allows to supply power without interruption to the crystal oscillator 65 in the oscillator circuit 35, to the real-time clock of the digital electronics unit 37, and to the data memory of the field terminal board 39, and external source of electrical power. Thus, the frequency stability of the oscillator will be maintained; the clock time setting will not be lost and the data stored in the computer memory will also not be lost.

Oscillator 65 in oscillator circuit 35 is a quartz crystal oscillator, e.g. one frequency 100 oscillator of the Frequency and Time System (FTS), which provides an output frequency

Q of 5,115 MHz within a portion of 10 or less. The FTS model 1001 has a stability of approximately 1 part of 10 µ θ per day and a part of 10 over time intervals from 1 to 10 seconds and is therefore more than suitable for this application. Oscillator 65 provides two identically similar outputs, one passing to digital electronics unit 37 and the other to a 1575.42-MHz synthesizer 67 in oscillator circuit 35.

The 1575.42-MHz synthesizer 67 contains a voltage controlled transistor oscillator (VCO) 69 which oscillates with a frequency 40

460 685 is of 393,855 MHz, which is 77 times 5,115 MHz. The phase of this oscillator is stabilized with respect to the phase of the 5,115-MHz reference by the action of a phase-locking loop consisting of the oscillator VCO 69, a coupler 71, a divider 73, a phase frequency field detector 75 and a loop filter 77. A portion of the output power of the oscillator VCO 69 is coupled by the switch 71 to the input of the frequency divider 73, which consists of integrated circuits with standard emitter coupled logic (ECL) which divide by 11 and then by 7. The output of divider 73 is the variable input signal and the oscillator 65 5,115 MHz output is the reference input to the phase frequency detector 75, such as the standard ECL integrated circuit, e.g. of Motorola type No. MC 12040. The output of the detector 75 is low-pass filtered in the loop filter 77 to obtain the control voltage which is input to the oscillator VCO 69. The output of the oscillator VCO 69 is quadrupled in frequency by a sequence of two frequency doublers 79 with balanced diodes of standard type and amplified by an amplifier 81. obtaining the 1575.42-MHz output frequency that drives the sideband separator 33.

The signals in a band centered at 1575.42 MHz received from the antenna unit 21 via the coaxial cable 25 at the input 83 of the sideband separator 33 are coupled through a direct current capacitor 85 through a bandpass filter 87 and amplified by an input amplifier 89 pre-amplifier.

29 (in the antenna unit) is coupled to the coaxial cable 25 via a radio frequency choke 91 from the receiver power circuit 34.

. The radio frequency power divider or hybrid 95, 1575.42MH-Z local oscillator hybrid 9.5, with 50 ° phase offset, the two double balanced blends 97 and 9'9 'together with the broadband frequency frequency 101 with 90 ° phase offset in the sidebar separator comprises with single sideband radio frequency-to-baseband converter or conventional phase-type demodulator. Such a demodulator has been described, for example, in an article in Proceeding of the IEEE, volume 59 (1971), pages 1617-1618 by Alan EERogers. Its operation here can be described as follows.

Let f indicate the frequency of the reference signal supplied to the sideband separator 33 by the oscillator circuit 35. Nominal is equal to 1575.42 MHz, which is equal to the nominal carrier frequency of the GPS satellite's L1 band transmissions before

460 685 (first order) Doppler offset. In this case, the outputs 102 and 103 of the phase shift hybrid 95 can be written as their 2πΐθΐ and cos 2πίθΐ, respectively. These outputs, which are in 90 ° phase difference, constitute local oscillator inputs to the mixer 97 and 99, respectively. The radio frequency inputs to the two mixers are identically the same. Accordingly, the baseband output signals of the mixers are identically identical except for a phase shift of π / 2 radians. (With baseband we refer to the range of frequencies, closer to zero than f, to which moto responds the difference between the input frequency and f_Q.) The direction of this phase shift, positive or negative, depends on whether the input frequency is above or below f. Thus, it is possible to select inputs with either upper sidebands (higher input frequency) or below sidebands and discard the opposite sideband by shifting the phase of a mixer output by adding π / 2 radians, after which the two mixer signals are either added or subtracted ( depending on which sideband is desired).

The 90 ° phase shift hybrid 101, which has two inputs 1u9 and 111 as well as two outputs 105 and 107, performs this π / 2 phase shift as well as addition / subtraction. The upper output 105 of the hybrid 101 is obtained by the arithmetic sum of the upper input 109 and the lower input 111, both inputs having been delayed in phase by an amount depending on the frequency, the phase offset of the lower input being, however, greater than the phase offset of the the upper input signal with a constant corresponding to π / 2 radians regardless of frequency. The lower output 107 is obtained by the arithmetic difference between the same two differentially phase-shifted input signals _Lp, where f_credits = 10 kHz is much smaller than _Lp is approximately equal to the one-sided of the GPS system's C / A modulation m (t), as before. The design of a 90 ° phase shift has these properties, obtained in the stated Rogers.

Now, the outputs of the 90 ° phase shift hybrid 101 are amplified separately by identically similar video amplifiers 113 and 115 and in

109 and 111, the difference being taken in the direction of the upper minus the lower. The specified phase shift with π / 2 radians (one quarter cycle) is accurately maintained for all frequencies between fpjp and at least f_p, where fj_[P = f ^_p = 450 kHz, and f the bandwidth discussed hybrid, as the article by

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460 685 are filtered by high-pass filters 117 and 119 and low-pass filters 121 and 123. The filters 117 and 119 are identical to high-pass filters with low-frequency disconnection at f ^ p. lower than the maximum possible size of Doppler offset that a GPS satellite signal could have.

It is desirable to eliminate all such components, as they could otherwise interfere with the subsequent determination in the digital electronics unit and the computer of the field terminal of the received Doppler offset carrier phase.

•15

Such potentially interfering signals could include low frequency flash noise generated in the mixes themselves or they could be obtained by a combination of mixer balance and (undesired) low frequency amplitude or phase fluctuations of the 1575.42 MHz reference signal or amplification. of any radio frequency signal amplifier placed before the mixes. Another potential source of low frequency interference is hum or pulsation with respect to the power source voltages or currents. Another source could be a frequency close to f · o

Low pass filter with bandwidth sideways bandwidth of m (t). The characteristics of each as a function of frequency are adapted to power spectral interfering unmodulated carrier signal

121 and 123 are identically the same as f p, which is equal to that of the low-pass filter density of m (t). The purpose of these filters is to eliminate noise and interference outside the bandwidth of m (t). Note that the broadband GPS-P code modulation signal n (t) here could normally be a source of interference. Most, approximately 80 '_O, of the power derived from n (t), is eliminated by these low-pass filters. This degree of elimination is sufficient to ensure that the P-code interference has a negligible effect. However, we note that if the narrowband m (t) modulation was coupled to bor-t in the GPS satellites, the broadband n (t) modulation would no longer represent an undesired interference signal. It would be the desired signal. Such coupling in the GPS signal structure could be satisfied by increasing the bandwidth of low pass filters 121 and 123 by

460 685 a factor of 10 for adapting the same to the new signal. The output signal u (t) from the low-pass filter 121 represents the down-converted and filtered upper sideband component of the original signal s (t); and the output signal δ (t) of the low pass filter 125 represents the lower sideband. It should be noted that the spectrum of u (t) will be shifted upward in frequency and that the spectrum of £ (t) will be shifted downward in frequency relative to the spectrum of the original modulation m (t) by an amount equal to (ff) corresponding to the difference between the actual received carrier frequency f and the local oscillator frequency f_O'. / If the doppler offset of the carrier, (f- £) is negative, then the u (t) spectrum is shifted downward and 2 (t), upward. / The magnitude of this offset is assumed to be less than f +_p. This assumption will apply if the frequency offset primarily stems from the doppler offset, which can never exceed 5 kHz in size, provided that f f p is set approximately equal to 10 kHz. Any offset of the frequency of the reference crystal oscillator 65 from the desired frequency of 5,115 MHz will also cause a (308 times greater) offset of spectra u (t) and £ (t). Normally, however, such a shift will be much smaller than f ^ p. In addition to the frequency shift of the upper and lower sideband outputs u (t) and £ (t), there is a frequency dependent dispersive phase shift of each output depending on the 90 ° phase shift hybrid 101. however, in particular 90 ° phase-shifting hydride, designed by Rogers (op.cit.), this phase shift is too small to be significant. Similarly, additional phase shifts introduced through bandpass filter 87 as well as high and low pass filters 117,

119, 121 and 123 to be trivial if standard filter designs are used. Each of these effects also tends to be eliminated when the difference between terminals is obtained in the subsequent data processing. The elimination is not exact because no two filters are ever exactly the same; also the dPppler firings at different locations are different at each

Si 'has been shown by direct calculation and confirmed by real experiment.

Referring now to Fig. 5, a block diagram of the digital electronics unit 37 is now shown.

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460 685

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the unit 37 includes a signal conditioning device 125, a correlator unit 125 comprising a group of seven identically similar correlators, a numeric oscillator unit 129 comprising a corresponding group of seven identically similar numerical oscillators, and a real-time clock 131, wherein the correlator unit 127, the numeric oscillator 129 and the real time clock 131 via a data bus 133 are connected to each other and to the field terminal computer 39. The first function of the signal conditioning device 125 is to convert the analog upper sideband signal u (t), the analog lower sideband signal δ (t) and the analog sinusoidal 5,115 MHz signal separately into a digital or logical binary value signal which is suitable for processing conventional transistor-transistor (TTL) circuits.

The signal conditioning device 125 produces two output signals. One is a binary valued periodic TTL logic level square waveform · with a frequency of 10.23 MHz, which is produced by frequency doubling of the 5,115-MHz input signal. This 10.23-MHz output serves as a clock signal to control the timing of all subsequent digital circuits. This clock signal is divided by 1023 (= 3 x 11 x 31) in real time clock 131 to obtain a ticking per 100 microseconds; further divisions with successive factors of 10 then give a full decimal representation of the time in seconds, the least significant number representing units of 10 ^ seconds. Time is always readable in this form via the data bus 133. The operation of the correlator unit 127, the operation of the numerical oscillator unit 129, and the operation of the field terminal computer 39 are all controlled by the real time clock 131 via the data bus 133.

The second digital output of signal conditioning device 125 is obtained from the analog u (t) and £ (t) inputs and is a binary valued non-periodic TTL logic level waveform. This output is produced by a logical TTL35 -EXNQR gate which has two inputs: one input represents the sign of the u (t) input and the other sign of £ (t). Thus, the gate output is true (T or binary 1) if and only if the analog u (t) and £ (t) signals have the same sign.

Figure 6 shows a block diagram of signal conditioning.460 685 ²² device 125. The analog signal u (t) is input to a comparator 135, the output of which is a TTL logic level signal, which is true when u (t) is positive and false when u (t) is negative. This TTL logic signal is transmitted as an input to a TTL5 EXNOR gate 137. The analog signal 1 (t) is similarly fed to a comparator 139, whose output is transmitted in the same way as the second input of EXNOR gate 137. The sine-5,115 MHz signal obtained from the crystal oscillator 65 is input to a conventional analog frequency doubling circuit 141, the output of which is fed to a third comparator 142 to provide a

The 10.23 MHz square wave TTL level output 10.23 MHz output is also used as a clock input to a flip-flop 145 which samples and holds the output of gate 137. Thus, flip-flop 145 output is the EXNOR function for the characters of u (t) and 2, (t) sampled at a uniform rate of 10.23 x 10 3 times per second and retained between the sampling times. It is well known in the field of radio interferometry, as used by, for example, JM The moral of an article in Methods of Experimental Physics, volume 12, part C, pages 228-260, is that the binary time-dependent function UQL has a Fourier transform or a Fourier spectrum, which is a good approximation both in phase and relative amplitude to Fourier spectrum of the analog product · u (t) £ (t). The accuracy of the approximation depends on the analog signals, which are random and Gaussian. In addition, the correlation coefficient between the two inputs must be much less than 1. (In fact, the noise shakes out the non-linearities of the comparators. EXNOR gate 137 can be considered as a multiplier in which each input has values ​​of +1 and -1.) These conditions are well met in the present system. Thus, in the following, the logic level from flip-flop 145 is simply represented as the product u (t) £ (t).

The USL product from the signal conditioning device

125 constitutes an input signal in parallel form to each of seven identically similar correlators in the correlator unit 125.

Before describing the construction of the correlator unit 127, its operating principles will be briefly explained.

*

In the correlator, the u (t) £ (t) product is correlated with binary approximations to the sine and cosine * functions of time, which are generated by a corresponding oscillator of the seven numeric oscillators. The frequency of the oscillator is controlled by the field terminal.

460 685 computer 39 according to the time indicated by the real time clock

131 levels. At any given time, the oscillator frequency is set equal to twice the predicted Doppler frequency offset of the 1575.42-MHz carrier transmitted by one of the satellites.

An oscillator and a correlator are associated with each of the satellites within sight, up to a maximum of seven satellites. If more than seven satellites were ever within sight, in principle several numerical oscillators and correlators could be used in the system. In practice, seven satellites are enough). If the predicted doppler offset is close enough to the actual doppler offset, the correlator outputs will accurately measure the power and phase of the signal from the particular satellite for which the prediction was made, and the measurement will not be significantly affected by the presence of signals from other satellites. , which has other doppler offsets.

In mathematical terms, the operation of one of the numerical oscillators and its associated correlator is described as follows: As a function of time, t, which is indicated by real time clock 131, the predicted doppler frequency shift of the satellite carrier of fp (t) is given. The value of fp (t) is interpolated from the table by pre-calculated values, which were previously stored in the memory of the field terminal computer. The numerical oscillator generates two time functions: £ 2φ (t)] and its [2φρ (ΐ) 3, with a phase difference of 90, where 4&gt; p (t) represents a predicted phase, which is a function of time.

The function &lt;J&gt; p (t) is initially equal to zero at time t, when the numerical oscillator begins to oscillate;

and at any subsequent time &lt;J&gt;_p(t) the time is obtained through the integral

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where £ (t ') represents the instantaneous value of f at an intermediate time t'. The factor 2π is necessary if, as is usual, the frequency fp is measured in units of periods per unit of time and the phase φ is assumed to be measured in units of radians instead of by periods.

Now it forms the correlator, which operates between the times t_Q and tp the quantities a and b of their inputs [u (t) £ (t) J,

460 685

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cos ^{and its} O (t)] according to the formulas ^u 1 a = fu (t) £ (t) cos Ρφ Ct) | dt t P and b = Ju (t) £ (t) sin £ 2φ (t)] dt. r. P

The time interval for the integration, ΐ ^ -ΐθ, equals 1 second and the suggested integrations are performed every second. With each one-second tick from the real-time clock, the values ​​of the integrals are strobed into the memory registers, the integrations are reset to zero, the numerical oscillator is restarted and a new integration period begins. Thus, at the end of each second, the correlator outputs the signals a and b, which represent the time averages during the previous one-second interval of the product cos [2φ (t) J and the product u (t) 2 (t) sin [2φ (t) J, respectively. These outputs represent the correlations of the product u (t) £, (t) with the cosine and sine functions.

During the one-second interval, the oscillator frequency fp (t) is updated for every tenth of a second by the computer while controlling the tenth of a tick from the real-time clock. This update is necessary, as the satellite's doppler displacement changes depending on the satellite's movement relative to the field terminal on the ground and the varying projection of the relative velocity along the target line at a rate that can be a substantial part of one Hz per second.

Now the correlator outputs a and b can be combined to obtain the power and carrier phase counts of the signal from the particular satellite for which the prediction, f_p(t) was made.

Define a complex number c, whose real part is equal to a and whose imaginary part is equal to b. Thus c = a + jb where j is the square root of minus one. Then c - C &lt;m applies²&gt; &lt;exp [2j (ψ-φρ)]&gt;

where C is a positive real constant scale factor: where &lt;m&gt; is the time mean over the integration interval from t_Q to t₁

460 685 of the square of the GPS modulation function m (t); and where &lt;exp [2j (φ-φ) J&gt; is the time mean over the same interval of the complex exponential function θχρ | _2] (φ-φ) J. provided

P that the difference (φ-φ_ρ) between the received GPS carrier phase φ = φ (ί) and the corresponding prediction φ_ρ = φ_ρ(ΐ) does not vary by a substantial portion of a period during the integration time, the magnitude of c is approximately proportional to the mean of the received power:

and the angle of c is approximately equal to twice the mean of the phase difference (Φ-Φ):

i - ^ψρ '/ c = tg^_1(b / a) - 2 &lt;(φ-φ)&gt;.

Note that the angle of c is determined separately from b and a with respect to the module 2π radians. Thus, the difference (φ-φρ) with respect to the 'module π radians is determined.

In order for the received signal power and the received carrier phase (with respect to module π) to be accurately determined from a and b according to these formulas, two conditions must be fulfilled: firstly, as mentioned, the actual phase φ (ΐ) must differ from the predicted phase φρ (ί) with an amount that changes by much less than a period during the one-second integration time; secondly, the noise ratio must be from the generator, which is given through

SNR_C = (2 / π) (π / 4) (B_effT_int)^1/2 F ^{fl / 2KB}eff^Tinb^1/2

F, be much greater than 1, where b b is the effective bandwidth of signals u (t) and Z (t) equal to approximately c

5x10 Hz; where is the integration time, which is equal to one second, and where F is the part of the power that occurs in uu (t) and l (t) and derives from the GPS-m (t) signal and not from noise. The factor (2 / π) is responsible for the loss of correlation between u (t) and δ (t) caused by the analog-to-digital conversion of these signals through the comparators in the signal conditioning device. The factor (π / 4) accounts for the loss associated with the use of square wave40 approximations for the sine and cosine functions of the correlator.

460 685 26

The square root of the product is approximately equal

700th Therefore, the relationship applies:

SNR_C = 350 'F.

The part F of either sideband effect derived from the GPS satellite depends on the receiver antenna gain and the receiving system's noise factor. For the MITES antenna and the receiver system described above and for a satellite elevation angle above 20 °, it is known from experiments that F exceeds approximately 0.03.

That's why it applies

SNR_C &gt; 10, which is sufficient for accurate power and phase measurements. The standard deviation of the noise in each part, real and imaginary, of the complex quantity C is obtained by o_c - | c | / SNR_c.

The former condition for accuracy in the power and phase measurements, namely that (φ-φθ) does not vary by a substantial portion of a period during the one-second integration time, is equivalent to the condition that the difference between the actual received carrier frequency f and the local reference frequency f does not deviate from the predicted (numerical oscillator) frequency fp with a substantial part of a Hz. This condition is met in the present system by applying feedback control with respect to the frequency of the numerical oscillator to keep this frequency close to the actual received carrier frequency. This control is exercised by a simple program executed by the field terminal computer 39. A description of this program follows.

The complex number c, formed by the a and b outputs of the correlator at the end of the kth one-second integration interval, is denoted c (tj, where tj represents the time at the middle of this interval.) To the numerical oscillator frequency for it (k +1): take the interval a corrective impact part is added

K * / [c (t_k) c * (t_k_₁)] / 2K Hertz, where K is a positive real constant less than 1, where /.[J ^AnS^your

460 685 the angle of the complex magnitude enclosed by the parentheses []; and where c (t ^ _ p is the complex conjugate of the complex number c from the nearest preceding one, (k-1): take the interval.

The principle of operation with this program can be understood by the following example: if the frequency prediction is, for example, too low by 0.1 Hz, the angle of c will be ahead by a tenth period in a second and the complex magnitude *

c (t ^) c (t | __p will have an angle (+ 0.1) x (2π) radians (plus some zero mean noise). The addition of the impact portion * which in this case is positive will reduce the magnitude of the negative the frequency prediction error from (0.1 Hz) to (1-K) x (0.1 Hz).

The value of K must be greater than zero for otherwise no reduction of a frequency prediction error will be obtained through the feedback. The value must be smaller than otherwise the feedback will result in unstable oscillation with respect to the error due to the delay in applying the correction. The exact value is not critical and the optimum value can be determined by experiment. A nominal value of 0.5 was used in the present system.

An important other factor for this frequency feedback is that the frequency of the numerical oscillator will be drawn toward the actual received carrier frequency from an initial frequency, which may be as many as several Hertz above or below it. This retraction phenomenon is well known in the art of phase or frequency tracking feedback loops, as discussed, for example, in the book entitled Phaselock Techniques by Floyd M. Gardner, published by John Wiley &amp; Sons, Inc. New York 1966.

The significance of the withdrawal phenomenon for the present system is that the knowledge, a priori, of the measurement mark situation need not have less uncertainty than a few kilometers.

A potentially adverse side effect of the retraction phenomenon in the present system is that the numerical oscillator assumed to follow a particular satellite may instead be drawn to the frequency of another satellite if the latter's frequency is close to that of the latter and if the latter's signal is strong. compared to the last one. To limit the damage that could occur from these events, the field terminal computer program contains a provision that limits the size of the accumulated

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460 685 actuating portion, which can be added to the previous frequency prediction, to approximately 10 Hz. Since the difference between the frequencies of two satellites normally changes by approximately one Hz per second, it follows that only approximately 10 sec.

S of measurement data or less than approximately 1 percent of the total data obtained at a field site can be eliminated by tracking the wrong satellite. Experience indicates that this percentage is insignificant.

Referring now to Fig. 7, there is now shown a block diagram of a correlator module 149 which is one of the seven identically similar modules in the correlator unit 127. All seven modules have the same input signal UQL, which is the USL output of the signal conditioning device 125. Each module also receives a cosine input and a sine input from a corresponding module of the seven numeric oscillator modules. The UQL input and the cosine input pass to an EXNOR gate 151, the output of which is the input of a clocked digital counter 153. The UHL input and the sine input pass to another EXNOR gate 155, the output of which is the input of another counter 157. Once per second, the counters registers 153, 157 in the respective output buffer 159, 161 are locked by a pulse from the real-time clock 131 in the digital electronics unit 37 and the counters are thereby reset to zero. At a rate of 10.23 MHz, which is controlled by the clock signal from the signal conditioning device 125, each counter 153,

157 a supplement with one if and only if its input from its associated EXNOR gate 151, 155 is true. Thus, at the end of each one-second interval, the contents of the output buffers 159, 161 indicate the number of times between zero and 10,230,000 that the UQL and cosine / sine inputs matched during the previous one-second time. The content of the output buffers 159, 161 of each counter is coupled to the data bus 135, through which the field terminal computer 39 reads the content every second. Each counter / latch can be a single integrated circuit,

e.g. the 32-bit model of model No. LS 7060, manufactured by LSI Systems, Inc.

The magnitude a, previously defined by the cross correlation between [u (t) £ (t) J and cos [2 &lt;j&gt; (t)], is obtained in the field terminal computer 39 by subtracting 5,115,000 from the output of cosine calculator 40 and the result divided by 5,115,000. The magnitude b is obtained

460 685 similarly by subtracting 5 1 1 5 000 from the output of the sine '' counter and dividing the result by 5 115 000. (Thus, reference size for a or b represents perfect correlation between Qu (t) Ä (t) J and cosine. Before these results are stored in the memory of the field terminal computer 39, each number can be reduced to only four bits to save memory space.)

Referring now to Figure 8, a block diagram of one of the seven identically identical numeric oscillator modules 163 of numeric oscillator unit 129 is illustrated, each module 163 transmitting a cosine and a sine input to a correlator module 149. Each numeric oscillator 163 comprises a binary phase register 167 and a binary frequency register 169; a binary adder 171; an EXNOR gate 173; a converter 175; and a frequency divider 177.

The phase register 167 and frequency register 169 each have 32 bits and the adder 171 is a 32 bit adder. The binary number found in phase register 167 at any point represents the phase of the oscillator output, the most significant bit representing half a cycle, the second most significant bit a quarter of a cycle, and so on. The binary number present in frequency register 169 similarly represents the frequency of the oscillator, the most significant bit in this case having a value of 155,000 Hz, which equals one 66-part cycle per 10.23-MHz period The clock signal from the signal conditioning device 125. The adder 171 adds together the numbers contained in the frequency register 169 and the phase register 167. The sum is entered in the phase register 167 and replaces the previous content once per cycle for the output of the divider 177 which divides the 10.23 MHz clock signal by a determined factor 33. The phase register 167 is thus updated at a rate of exactly 310 000 per second. . The amount by which the phase is increased with each update is obtained by the contents of the frequency register 169. The frequency register 169 is updated, as mentioned, 10 times per second via the data bus 153 by the field terminal computer 39. (Both negative and positive frequencies are represented by the content of the frequency register using the conventional two-complement method. According to this convention, the negative portion of a binary number is formed by completing each bit, then one is added.

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685 accordingly, by leaving the most significant bit zero and all other bits one. The fact that the most significant bit is one means that the number is negative.)

The sine output of the numerical oscillator 163 is obtained from converter 175 which converts the most significant bit of phase register 167. The sine output has a value one when the phase is between zero and half a cycle, and a value zero when the phase is between half and one cycle. (which is the same as the phase is between minus one and a half and zero cycles). The cosine output of the numerical oscillator is obtained from EXNOR gate 173, whose inputs are the most significant and the second most significant bit in the phase register. The cosine output only when the phase is within plus with respect to zero.

Referring to the figure of the field terminal computer 39. The computer (CPU) 181, a program memory 183, a data memory 185, two-way data port 187 connected to an operating terminal 189, and an external two-way data port 191 connected to a modulator. demodulator (modem) 193, which in turn is connected to a telephone line, a radio telephone or some other interconnected by a data bus 133, which also serves to connect (see Fig. 5).

Central unit 181 may be of type Digital Equipment Corporation (DEC) model LSI-11/2 (part number KD11-GC); the program memory 183 may be a programmable permanent 32-K character group memory e.g. of type DEC with part number MRV11-C; the data memory 185 may be a permanent 32-K character group direct memory, e.g. type DEC with part number MXV11-AC; the two outer two-way data ports (187 and 191) may be the RS-232 series data ports included in the MXV11-AC; the operating terminal 189 may be DEC model V * T-100 or any equivalent series-ASCII terminal which, like said model VT-100, can be connected to the device's MXV11-AC RS-232 series data interface or to the computer via any other suitable external data port device; modem 193 may be a standard has a value one when and or minus a quarter cycle is now shown a block diagram includes a central unit an external telecommunication link 195. The computer 39 parts are the computer 39 to other parts of the field terminal in

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460 685 modem which is compatible with the RS-232 device and can be completely eliminated if the field terminal computer 39, as mentioned, is connected directly to the base terminal computer 15. The data bus 133 can be the LSI-11-Q bus. The real-time clock 131, the numeric oscillator unit 129, and the correlator unit 127 may be connected to the Q bus by being mounted on standard circuit boards, which directly plug into short edge switching portions of the back plane of an LSI-11 computer system. Such circuit boards are provided from DEC equipped with special integrated circuits which can handle all data communication between the Q bus and the special interferometer terminal circuits mounted on the cards. ' The measurement data stored in the memory 185 of the field terminal computer 39 includes a time series of complex numbers for each of up to seven observed satellites, such a number being obtained every second. This data is obtained over a period of approximately 5000 seconds, during which at least two satellites are always observed, with the average number of observed satellites being at least four. For the lithe satellite at time t, complex data is denoted A (ft the angle of speech is equal to twice the carrier phase measured for the same satellite at the same time, the phase of each satellite being referred to the same local reference oscillator signal, namely the 1575.42-MHz signal generated by field terminal 13-1 oscillator circuit 35.

Complex data A ^ (t), where i = 1 ..., 7, is obtained by the field terminal computer 37 from the a and b outputs of the seven correlators 149 in the correlator unit 127 as follows. The first correlator applies;

A_in(t) = (a (t) + jb (t) J exp [2j $_p(t)], wherein a (t) and b (t) represent the normalized a and b outputs, respectively, for the 1-second integration or counting interval centered at time t; where j is the square root of minus one; and wherein 2φ_ρ(t) is twice the predicted carrier phase of the farthest satellite at time t. Note that the complex number A

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460 685 complex numbers c, obtained from the 1st correlator output, multiplied by [2j4 (t)]. The A-angle of the number represents / p (twice) the received carrier phase referred to the (twice) 1575.42-MHz local reference phase, whereas the c angle of the number is assigned to (twice) the sum of this reference oscillator phase plus the phase of the numerical oscillator.

For the purpose of this explanation, it is considered that the data set (A ^ (t)} is that generated by the field terminal 13-1 which is at the beginning of the baseline vector. The second field terminal 10-2, which is the field terminal at the end of the baseline vector, is observable. the same satellites at the same times as the first terminal, leaving data corresponding to A ^ (t), designated B ^ (t). The same satellites are observed, since both terminals were assigned prediction data from the same central computer 15, which numbered the satellites 1-7 in just one way. The observations at the two terminals are completely simultaneous, since the clocks of the two terminals were synchronized immediately before the observations, and the clock speeds differ from each other by a trivial amount. (The main effect of the speed difference between the crystal oscillators controlling the clock speeds is to vary the phase difference between the 1575.42-MHz references.) It does not matter if a particular satellite is visible from one terminal at a particular time but hidden with respect to on the other. The strength of either A 2 (t) or

B 2 (t) in this case will simply be zero or almost zero.

The operations of the central computer 15 to complete the determination of the baseline vector for the interferometer, which has obtained the power and phase measurement data collected by two field terminals 13-1 and 13-2 located at the ends of the baseline vector, will now be discussed.

The first step in the processing of A 2 (t) and B 2 (t) data in the central computer comprises multiplying the complex conjugate of A 2 (t), denoted A * (t), by (t). The product³⁵ S (t) BiCt) has an angle, / S 2 (t), which is equal to twice the difference between the measured phases of the carrier signals received from the satellite at the two terminals, each phase has been measured with respect to the local reference oscillator

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460 685 in the respective terminal. Accordingly, the angle of S t) is related to the difference between the phases of the local oscillators and to the baseline vector between the terminals with the theoretical relation yS_±Ct) = Δφ_Ε0 + (4π £_ί/ ο) £ · s_in(t), where Δφ ^ θ represents the local oscillator phase difference, whereby the received frequency of the ith satellite is almost equal to 1575.42 MHz, c is the light speed, where är is the baseline vector, and s ^ (t) is a unit vector. in the direction of the giant satellite, viewed at time t from the center of the baseline vector. (This relation gives the angle / S.(t) in radians instead of cycles. Since the frequency f 1 is given in cycles, rather than in radians, per second, a factor of 2π must be included. The condition that 4π instead of 2π occurs here is that each field terminal measures twice the received signal phase.) This relationship is approximate as it ignores the second-order parallax, propagation medium effects, multi-path effects, relativistic effects, disturbances etc. These small effects are neglected here for clarity. The error associated with the failure of these effects is equivalent to a baseline error of less than approximately 1 cm for a baseline length less than approximately one kilometer. (With the exception of the effect of noise, which is completely random, it is possible to shape the effects that we have neglected above to obtain a more accurate theoretical representation of / S 2 (t). This form is described, for example, in an article of II Shapiro, entitled Estimation of astrometric and geodetic parameters from VLBI observations, in the publication Methods of Experimental Physics, volume 12, part C pages 261-276, 1976.).

Theoretically, the size of S is obtained by | S_±| = CG²(cos6_in) where C is a constant and G is the directional power gain of a receiver antenna, written as a function of the cosine of the zenith angle Θ of the 1st satellite. G is assumed to be azimuth independent and is normalized so that the power received by an isotropic antenna with adapted circular polarization is equal to 1. For the MITES design:

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460 685

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G (cos0)^c'(1.23) · (1 + cos0) 2' sin ^ ((3/4) cos0), o ° &lt;0 &lt;9O °;

G (cosO) - 0.9O ° &lt;0.

The value of this function is approximately 2.46 at zenith (6 = 0); it has a maximum of approximately 3.63 at 0 - 40 °, it has a value of 1 at Θ - 72 °, and it approaches 0 when Θ approaches 90 °.

The following step in processing measurement data obtained from the two interferometer terminals involves summing the complex numbers S 2 (t) for all in order to maintain a sum S (t) for each measurement time t:

n

S (t) = Σ S. (t), i = 1 ¹ the sum covering all the satellites observed at time t.

The next step in processing measurement data involves selecting a sample value, b, of the baseline vector F and calculating from this value b a function of time S (t) which theoretically represents the value that S (t) would have had the true value, i) for the baseline vector would be equal to the sample value, b:

~ n _

S (t) = Σ | A- (t) I · | B. (t) | · Expr-j4Kb-s. (T) /Ä.ji=1^{1 1} ‘ ^{1 1} the radio wavelength corresponding to the received carrier frequency. Thus = c / f ^. The method of selecting a value of b is described below. Note that in the theoretical function S (t), as opposed to the function obtained by measurement S (t), there is no term representing the local oscillator phase difference. In addition, the constant scale factor C is omitted.

Next, the size of S (t) is multiplied by the size of S (t) and the product of these sizes is summed over all the measurement times to obtain a value, R (b), which depends on b as well; Of course, on the measurements:

RCb) = Σ | S (t_£) | - | SCt_A? | , £

whereby t₀ represents the SL in the group of approximately 5000 measurement times. R (b) is called an ambiguity function.

The next step in the treatment involves:

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460 685 repeating the calculation of R (b) for different values ​​of b and determining the special value of b, for which the function of R (b) has the greatest value. This value of b represents the desired determination of the baseline vector K.

The sample value b for the baseline vector is initially chosen to be equal to the best estimate, a priori, or even, which is available from independent information about the locations of the measurement marks, e.g. the locations obtained by identifying landmarks on a map. The maximization of R (b) with respect to b is accomplished by examining a three-dimensional volume, which is centered on this initial value of b and which is large enough to span the uncertainty of the initial estimate. In the investigation, each point is examined by a uniformly spaced three-dimensional grid for locating the single point, where R (b) is maximum. The grid space is initially a meter. Then the volume, which extends two meters from the single Λ point of maximum R (b), is examined by examining a grid with

Λ a gap of 20 cm. The maximum value for RCb) is found at this grid at smaller intervals. Then the grid space is halved and the linear extent of the grid is also halved and the examination is repeated. This process of halving continues until the grid space is less than 1 millimeter. The value A Λ of b, which finally maximizes R (b), is considered the desired determination of the baseline vector ί. Using a number of satellites, where n equals 5, a baseline vector determination can be obtained by the method of the present invention, which determination has an accuracy of approximately 5 millimeters in each coordinate for a baseline length of approximately 100 meters.

The method described above for processing measurement data from a pair of interferometer terminals to determine the baseline vector between the terminals represents a specialization of the general method, described in an article by Charles C. Counselman and Sergei A. Gourevitch entitled Miniature Interferometer Terminals for Earth Surveying : Ambiguity and Multipath with Global Positioning

System, published in IEEE Transactions on Geoscience

460 685

5 &lt;= * w

and Remote Sensing, volume GE-19, no. 4 pages 244-252, October 1981.

In another embodiment of a method for processing measurement data according to this invention, an ambiguity function R (b) is also formed of said measurement data and of a sample value,

A b, for the baseline; however, the method of forming the function is different. In this embodiment, as in the previous embodiment, the complex conjugate of A ^ (t) is multiplied by Ib (t) to obtain a complex product S ^ (t):

S ^ t) = A? (T) B_in(t) wherein A 2 (t) is a complex number representing the measurements of the signal received from the i.th satellite at an interferometer terminal at time t, the size of A 2 (t) being proportional to the received the effect and wherein the angle M ^ (t) is twice the phase of the carrier relative to the local oscillator of the terminal, wherein B ^ (t) is the same as A ^ (t) except that the value is obtained from the other terminal at the other end of the baseline vector .

Subsequently, S. (t) is multiplied by a single complex exponential function of a sample value, b, for the baseline vector and the product is then summed over all satellites observed at time t to obtain a sum S (t)&gt; which is a

A function of time and of sample value b: n -.

S (t) · Σ S. (t) exp | -j4irb «s. (T) / X. | i = 1^{1 1} wherein s ^ (t) is a unit vector in the direction of the i.th satellite at time t and wherein is the wavelength of the signal received from the i.th satellite. (Note that if b «b, then the angle of each term in the sum is equal to Αφ ^ θ independent of i.)

Then the magnitude of S (t) is taken and summed over all observation times to obtain the function R (b):

R (b) = Σ | S (t_p) |, £ * whereby t₀ is the £ th of the approximately 5000 measurement times. Finally, the value of b, which maximizes R (b), is found by the same examination procedure as that described in connection with the original data processing method. This

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460 685 value of b represents the desired determination of the baseline vector

This latter embodiment is more efficient in computational terms than the first described embodiment.

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Wi '

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