System for transmitting signals over an optical link.

The present invention relates to a measuring device with a transmitter side and a receiving side, and arranged for transmitting at least one measuring signal, for example the output signal from a measuring sensor, from the transmitter side of the measuring device to its receiving side via an optical link.

It is known that measurement signals and other signals can be transmitted via an optical link. Such a link consists of a light emitter, light transmitting means and light detecting means. The light emitter comprises means for modulating the emitted light in a suitable manner so that the emitted light contains information about the current value of the measuring signal. The light transmitting means, which may consist of a light guide (optical fiber), transmits the light signal to the detecting means, where the information about the measuring signal is converted into a suitable signal, for example to an electrical voltage proportional to the measuring signal. A measuring device of this kind has several advantages. Galvanic separation is obtained automatically between the transmitter side and the receiver side, which can thus be located at widely differing potential levels. Furthermore, the optical link, in contrast to electrical signal lines, is completely insensitive to electromagnetic interference. Another advantage is that in the case of an optical link, in contrast to electrical wires, sparks or short circuits can not occur, which is of value, for example in an explosive environment.

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In the case of a measuring device of the specified type, it is essential to reduce the power consumption on the transmitter side as far as possible, since the energy supply of the transmitter side means, for example in cases where they are located at high electrical potential, is often very complicated and costly. power requirements can be kept at a very low level. This makes the use of digital transmission of the measurement information via the optical link less attractive, since such a transmission normally requires power-decreasing analog-to-digital converters and light modulators on the transmitter side.

However, a conventional analog transmission depends on the gain / attenuation of the optical link. This can vary due to several reasons. Thus, the attenuation of a light guide varies, among other things, with changes in its radius of curvature, and the gain of the light emitting and light detecting means (LEDs, photodiodes) varies with temperature and time (aging). This has the consequence that the gain of the measuring device can vary, which results in low accuracy of the output signal of the measuring device.

Another disadvantage of a conventional analog transmission is that its dynamic range becomes small if the available power on the transmitter side is low.

The invention relates to a measuring device of the type indicated in the introduction, in which the transmitter side has a very low power consumption, at the same time as high accuracy, high dynamics and large bandwidth can be obtained.

What characterizes a measuring device according to the invention is stated in the appended claims.

According to a preferred embodiment of the measuring device according to the invention, it is provided with means with controllable gain (or attenuation), preferably on the receiver side, this gain being arranged to be controlled so that the total gain between the regulator and the comparating means is kept constant. This can be done by means of special calibration signals, which can be either time or frequency multiplexed with the measurement signal. This makes it possible to keep the gain of the measuring device constant with high accuracy.

The very low power consumption which can be achieved on the transmitter side of a measuring device according to the invention enables that in an embodiment of the invention the energy supply of the transmitter side means takes place by means of optical

7809961-1 energy transfer, which has obvious advantages in the applications relevant for a measuring device according to the invention.

The invention will be described in more detail in the following in connection with the attached Figures 1 and 2. Figure 1 shows a measuring device according to the invention with time division multiplexed calibration, and Figure 2 shows a measuring device according to the invention with frequency multiplexed calibration.

Fig. 1 shows an optically coupled measuring device according to the invention for transmitting an analog electrical input signal U1 to an analog electrical output signal U1. The feeder consists of a transmitter side S and a receiver side M, which are connected to each other by means of the four light guides LC1-LC4. The transmitter and receiver sides may be located at a distance from each other of, for example, a few meters up to several kilometers or longer. They can also be located at different electrical potentials, whereby the measuring device can, for example, transmit a current measured value from a high-voltage line (transmitter side) down to ground level (receiver side). Alternatively, for example where only a not too large potential difference is to be bridged by the measuring device, the transmitter and receiver side can be arranged in connection with each other, whereby the light conductors with associated light and photodiodes can be constituted by optocouplers.

The measuring device works according to the compensation principle. A comparing means, the summing circuit S1 on the transmitter side, compares the measuring signal U_in with a feedback signal U The difference ϋθ between these two signals is transmitted to the receiver side and controls a controller there (F7-C1). The output signal of the controller constitutes the output signal of the measuring device, and the feedback signal is a linear function of this signal. The controller operates until the said difference becomes zero, the output signal U of the measuring device. will be proportional to its input signal U. ·

Uu in

The input signal of the measuring device U_in is applied to the summing circuit S1 via a field effect transistor FBT1. This is controlled by a control logic SL on the receiver side. In the summing circuit, the input signal U1 is compared with the feedback signal U1 and the difference U controls the current through an amplifier F1 through an LED LD4. The voltage across the resistor is a measure of the current through the LED and is supplied with the changed sign back to the summing circuit S1. In this way, proportionality is achieved between the output signal U of the summing circuit and the current of the LED, which gives a good accuracy in the area c in the vicinity, of zero where the output signal U of the summing circuit and thus the diode current

C will be during operation of the measuring device. Via the light guide LC2, a photodiode PD2

7809961-1 and an amplifier F4, the comparison signal U is transmitted to the receiver side, where it is denoted U '_c. This signal is applied to a controller (amplifier F7 and capacitor C1) with integrating characteristics. The output signal from the controller constitutes the mains output signal U. This signal is applied via a field effect transistor FET2 and a summing circuit SJ an amplifier F6 with controllable gain. The output signal of the amplifier is applied to the buzzer of the transmitter side in the input field a1 S1 via a summing circuit S2, an amplifier F5, an LED LD2, the light guide LCJ, a photodiode PD3 and an amplifier F2 (whose output signal is called U The light from the LED LD2 is also applied to a photodiode PD5 of the same type as the photodiode PDJ. The output signal from the photodiode PD5 is applied to the summing circuit S2 with the character changed. As a result, a linearizating feedback is obtained which means that the output signal from the diode PD3 is a linear function of the signal U

To control the gain of the amplifier F6, the comparison signal U * is applied via a field effect transistor EEP3 to a second controller, the amplifier F8 with the feedback capacitor C2, with integrating characteristics. The control output U of the controller is applied to the amplifier F6 and controls its gain, p

The field effect transistors FEM, FET2 and FET3 operate as switching means and are controlled from a control logic circuit SL. This can for instance consist of an astable multivibrator (or other type of oscillator) and emits alternately signals at the two outputs a and b. The signals from the outputs a and b are applied to the transistors FEP2 and EET5. The signal from output a is applied to the transistor FEM via an amplifier F9 »an LED LD3», a light guide LC4, a photodiode PD4 and an amplifier F3 · When the signal is output on output a, the transistors FEM and FEP2 are controlled to conducting states. During this interval, the measuring interval, the controller F7-C1 adjusts so that there is proportionality between the output signal U and the measuring signal U The transistor FEPJ is non-conductive during the measuring range, ie the output signal of the controller F8-C2 and thus the gain of the amplifier F6 is kept constant. During the next interval, the calibration interval, an output signal is instead emitted at the output b of the control logic, whereby the transistor FET3 becomes conductive and FEM and FET2 become non-conductive. In this case, the input signal U1 is disconnected from the summing circuit S1 and the comparison signal U 'is connected to the input of the controller F8-C2, whereby the gain of the amplifier F6 is adjusted in the manner described below. Since gain changes in the optical link LD2-LC3-PD3 can be expected to take place slowly, the measurement intervals can, if desired, be made long compared to the intermediate calibration intervals. More sophisticated methods are also conceivable, where for example the time interval between the calibration intervals is determined by how fast the transmission characteristics between S2 and S3 vary and where the calibration5

The length of the interval 7809961-1 is determined by how fast the controller F8-C2 adjusts, whereby an adaptive continuous control of SL is performed on the basis of the output signals from F7-C1 and F8-C2.

During the calibration intervals, only the two accurately constant reference voltages υθ and U2 are applied to the summing circuits S1 and S3. These can be obtained from suitable reference voltage sensors, eg temperature compensated zener diodes. The controller F8-C2 is supplied with the comparator signal U *_c, which is proportional to the signal and thus to the difference between υθ and the return signal U * ^. The controller reaches steady state when U ^ U '^ O, i.e. when Men^U'fb'^{F is} the gain / attenuation in the transmission link for the feedback signal between the summing circuits S3 and S1. When the controller is set, it is therefore U_n »F'U, ie the reinforcement in^U0 the said transmission link is F = During the calibration intervals, the gain of the amplifier F6 is thus set so that the total gain of the feedback signal is always kept at the constant value specified above, accurately determined by the reference voltages ·

During the measurement interval, the transistors FET1 and FET2 are conducting and the transistor

FET 3 non-conductive. Regulator F8-C2 output U_g and thus the gain in the amplifier F6 is therefore constant and equal to the value set during the immediately preceding calibration interval. During this interval, the controller F7-C1 operates until its input signal and thus the comparison signal U / U¹ . cc becomes zero. Then is^Uin^{+ n}0 “^ * fb ⁰

But and which gives ie ^U'fb ”F * ^ i^{+ D}out) ^unow . - · u.

ut υθ in ^Uut ·

The gain of the measuring device = - is kept accurately constant in this way ^uof gain variations in the input optical transmissions.

The LD4-LG2-PD2 transmission from the transmitter to receiver side only acts as a zero point detector. The LED LD4 can therefore have a very small dynamic range and thus a very low power requirement. The only thing required by this transfer is

7809961-1 that it should have a defined zero point, which can easily be obtained by means of the current return R1-S1 described above.

Otherwise, the transmitter side consists of passive components (PDJ, PD4) and of the amplifiers FT, F2, F3, which can be designed as very power-efficient circuits. In addition, the amplifier F3 can be avoided if FEP1 is constituted by a photofield effect transistor and the function of F2 can be performed by F1, if the photodiode current from PDJ is used directly as an input signal to the summing point S1. The power demand of the transmitter side is therefore very low with a measuring device according to the invention, and so low that the required energy can be transmitted to the transmitter side by optical means. An example of such an optical energy transfer is shown in Figure 1. One rectifier or amplifier LR supplies a number of series-connected LEDs LD1 · Via one or more 1 light conductors LC1 the emitted light is transmitted to the receiver side, where it is converted by the series-connected photodiodes PD1 into electrical energy, which is supplied to a power supply circuit SA. By series connection of a suitable number of photodiodes PD1, the desired voltage level can be obtained.

The requirement for high dynamics and thus a large power requirement is with a measuring device according to the invention transferred from the LED (LD4) on the transmitter side to the LED LD2 on the receiver side. Pen later is generally located at ground potential or elsewhere where the energy supply offers no problems. Slightly reduced power requirements on the transmitter side generally bring very great benefits.

The dynamics of the measuring device are determined by the dynamics of the feedback link LB2-LC3-PD3.

Since the power supply of the LED LD2 does not present any problems, the dynamics of the measuring device can be made high.

The accuracy of the measuring device is also determined by how carefully the gain in the feedback link can be kept constant. By the above-described automatic calibration method, the gain of the feedback link can be kept very constant, and the accuracy of a measuring device according to the invention can therefore be made very good.

In some applications, the gain of the feedback link can be expected to be constant with sufficient accuracy, without the calibration procedure described, and the means for this can then be omitted.

The controller F7-C1 practically retains its output signal during the calibration intervals

7809961-1 unchanged, especially if another FET is introduced and used to disconnect the input signal to F7 during the calibration interval, and therefore no renewed oscillation sequence is needed at the beginning of each feed interval, which increases the upper limit frequency of the measuring device and enables simpler circuits for signal processing. of measure the output of the device.

The controllers F7-C1 and F8-C2 are shown in Fig. 1 as controllers with pure integrating characteristics. Alternatively, the controllers can consist of controllers with, for example, proportional or proportional-integrating characteristics.

Inputs U_in to the measuring device is obtained in the typical case from some form of sensor. In Fig. 1, the sensor has been assumed to emit an electrical voltage to the measuring device, but other types of input signals to the measuring device are also conceivable.

The transmission link LD3-LC4-PD4 from the control logic SL to the transistor FET1 can in some cases be eliminated if there is access on both transmitter and receiver side to some form of common sensor, eg mains voltage, for controlling the switching between measurement and calibration intervals.

Likewise, in some cases the energy supply to the transmitter side can take place in another way than that shown in Fig. 1, whereby the transmission link LD1-LC1-PD1 can be abolished.

If desired, a measuring device according to Fig. 1 can be used for transmitting several measuring signals. The transmitter and receiver sides are then provided with coupling means, which in turn connect the various input signals to the summing circuit S1 and in turn thereby switch the output voltage of the controller F7-C1 between corresponding outputs from the measuring device.

The terms “light guides, light detecting means, etc.” used above refer to devices that work with electromagnetic radiation both within the visible range and within adjacent wavelength ranges.

Figure 1 shows a measuring device according to the invention, where measurement and calibration take place alternately and can therefore be said to be time-multiplexed. Fig. 2 shows an alternative embodiment where the measurement and calibration signals are transmitted simultaneously but have different frequencies, ie they can be said to be frequency multiplexed.

In the same way as in Figure 1, measure the signal with the feedback signal^r in the summator 81 · The result, the comparator signal U1, is transmitted via the link 194-102-19) 2 to the receiver array, where it controls the controller F7-C1, the output signal of which is the measuring device. Via pre-torque crown F6 mod controllable reinforcement

7809961-1 and the link LB2-LCJ-PBJ transmit the feedback signal to the transmitter side. A second controller F8-C2 controls the gain of the amplifier F6 so that the total gain of the feedback signal between SJ and S1 is kept constant. The principal mode of action is thus the same. as for the measuring device in Fig. 1.

The transmission link for the comparison signal is chopper stabilized · From an oscillator 0SC1, which operates with a high frequency f in relation to the other frequencies present in the measuring device₁, a pulse train is obtained with the frequency f The transistor chops up the comparison signal with the frequency, f, and the thus chopped signal is transmitted via the link LB4-LC2-PD2 and an amplifier F4 to a demodulator M2, which may be a multiplier circuit. The modulator is controlled by the output signal of the oscillator OSC1 and emits a direct voltage signal which corresponds to the component with the frequency f Benna signal is applied to a low-pass filter LP1, where the high-frequency demodulation ripple is filtered out. In this way a safe detection of the zero level c of the comparison signal U is obtained independent of operation at the zero levels of the constituent components, for example due to the dark current of the photodiode PD2.

The calibration of the feedback link SJ-S1 takes place with the aid of signals with a frequency f_Q which is lower than f ^ but higher than the measurement signal U ^_n upper limit frequency. The oscillator 0SG2 emits an output signal with the frequency and an accurately constant amplitude υ ^ θρ is applied to the summing circuit SJ · The output signal from the oscillator is controlled via the LED LBJ and the light guide LC4_f and the AC voltage component V 'of the modulated signal_RE£ is supplied via a decoupling capacitor CJ to the summing circuit S1 · A component with the frequency f_Q is therefore included in the comparison signal U and is fed via the chopper stabilized c link FET4-F1-LB4-LC2-PB2-F4-M2-LP1 to a demodulator M1 which is controlled by the oscillator's OSC2 output signal · The modulation ripple is removed in the low-pass filter

LPJ, the output signal of which consists of a direct voltage with an amplitude corresponding to the amplitude of the component with the frequency ίθ in the comparison signal. The feedback signal U¹^ to the summing circuit S1 will contain a component with the frequency ίθ. The controller F8-C2 will control the gain of the amplifier F6 so that the input signal to the controller becomes zero, ie so that U^f »= V 'But since U¹. ^ = F is the gain in the feedback link between SJ and S1, and since V '«raf

F - art 7—.

ref konst · V_REp ”becomes

The gain of the feedback link is thus automatically kept at a constant value determined by the two reference voltages.

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The measuring signal U_in is compared in the summing circuit S1 with the feedback signal and the low frequency component of the comparison signal U^{1 * * *} fed via one. low-pass filter LP2 (with a cut-off frequency lower than Γθ and f) to the controller F7-C1 · The controller operates until its equalization U. is such that

U f U “

V =^Vin 'ie the gain of the measuring device ~ · «¹ becomes constantly independent due to, for example, changes in the gain of the optical link LD2-LCJ-PD3.

Instead of transmitting the output signals of the oscillators OSC1 and OSC2 to the transmitter side, separate oscillators can be arranged on the transmitter side. These oscillators are then suitably, like OSC1 and OSC2, crystal controlled so that the oscillators maintain the same frequency in pairs. Then, however, demodulators according to M1 and M2 can not be used, but instead bandpass filters and rectifier circuits must be used for the demodulation.

Transmission of energy to the transmitter side takes place, as in Fig. 1, via an optical link, which, however, is slightly differently designed. An oscillator OSCJ emits a pulsed DC voltage which, after amplification in an amplifier F12, supplies a light emitting diode (or semiconductor laser) LD1, the light of which is supplied to a photodiode ED1 via a light guide LC1. The alternating voltage component one in the current of the photodiode is transformed up to the appropriate voltage level by a transformer TR, after which it is rectified by a peak value rectifier, consisting of a diode D and a capacitor 04, and supplied to the power supply circuit axis SA for the transmitter side circuits.

The light and photodiodes used in the above-described embodiments of the invention can of course be replaced by other means for emitting or detecting electromagnetic radiation. The light guides LC1-LC5 can, for example at a short distance between the transmitter and receiver side, be abolished or replaced by, for example, a lens system. This can be advantageous, for example, if the transmitter is located on an object that is movable in relation to the receiver side. According to another alternative, several of the signals can be transmitted on the same light guide, for example by using light of different wavelengths or frequency modulated light with different modulation frequencies for the different signals.

In the embodiments described above, the comparison signal U is formed_c as the difference between the measurement signal and the feedback signal Alternatively, the comparison can be made, for example, by forming the ratio between the two signals, or in another way.

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CLAIMS '