Ofdm synchronization using two pilot symbols having a predetermined frequency shift between each other

[0001] Embodiments are generally applicable to communication networks. More specifically, the embodiments relate to synchronization of orthogonal frequency division multiplexing.

BACKGROUND [0002] Digital broadband broadcast networks enable end users to receive digital content including video, audio, data, and so on. With the help of the mobile terminal, the user can receive digital content via a wireless digital broadcasting network. Digital content can be broadcast in a cell within the network. The cell may represent a geographical area that may be covered by the transmitter in the communication network. The network can have many cells, and the cells can adhere to other cells.

[0003] A receiving device, such as a mobile terminal, may receive a program or service in a data or transport stream. The transport stream carries individual program or service elements, such as audio, video and data components from a program or service. Typically, the receiving device locates various components of a particular program or service in the data stream through program-specific information (PSI) or service information (SI) embedded in the data stream. However, PSI or SI signaling may be insufficient in certain wireless communication systems, such as DVB-H (Digital Video Broadcasting-Handheld) systems for digital terrestrial television. The use of PSI or SI signaling in such systems can lead to suboptimal end-user experiences because the PSI and SI tables carrying PSI and SI information can have long repetition periods. In addition, PSI or SI signaling requires a large bandwidth, which is expensive, as well as reduces system performance.

[0004] US 2004/0141457 A1 discloses a method of determining an adaptive signal to facilitate acquiring symbol synchronization, frequency offset estimation, and channel estimation in an OFDM system.

BRIEF SUMMARY OF THE INVENTION [0005] The following is a simplified summary to provide background knowledge of certain aspects of the invention. The summary is not a comprehensive discussion of the invention. Nor is it intended to identify key or critical elements

- the invention, or define the scope of the invention. The following summary only illustrates several concepts of the invention in a simplified form as a prelude to the more detailed description below.

[0006] The scope of the invention has been defined by means of the attached independent claims.

[0007] Exemplary embodiments relate to the first and second OFDM pilot symbols. The first and second pilot symbols have respectively the first and second sets of allowed, forbidden and active carrier frequencies. The second sets of carrier frequencies are formed by frequency shifting of the respective first sets by a predetermined frequency, such as the frequency difference between adjacent carriers. The embodiment relates to frequency translation of a portion of the first received pilot symbol by one carrier space in a first direction, frequency translation of a second pilot symbol portion received by one carrier space in a second direction that is opposite to the first direction,

BRIEF DESCRIPTION OF THE DRAWINGS [0008] The invention and its advantages may be more fully understood by referring to the following description in connection with the accompanying drawing figures, where like reference numerals indicate similar elements, in which:

Fig. 1 illustrates a corresponding digital broadband broadcast system in which one or more illustrative embodiments of the invention may be implemented.

Fig. 2 illustrates an example of a mobile device according to the subject matter of the invention.

Fig. 3 schematically illustrates an example of cells, each of which may be covered by another receiver according to the subject matter of the invention.

Fig. 4 shows a frame and a super-frame of symbols, synchronization symbols used to search channels and detect services and data.

Fig. 5 shows how the center frequency of the signal may coincide with or be offset relative to the center frequency of the channel.

Fig. 6 is a block diagram illustrating steps performed by a receiver.

Fig. 7 shows an example of the size of the pilot bandwidth relative to the signal bandwidth and the channel raster bandwidth.

Fig. 8 illustrates a rare pilot interval in a pilot sequence for a pilot symbol.

Fig. 9 is a block diagram illustrating steps performed by a receiver to perform frequency domain correlation to detect the coarse offset used.

Fig. 10 is a block diagram that illustrates the steps of performing the time domain detection service correlation.

Fig. 11 shows an example of a pilot / signaling symbol.

Fig. 12 is a block diagram illustrating the steps of a method performed by means of a transmitter.

Figures 13 and 14 illustrate the relationship between the symbols P1, P2 and DATA.

Fig. 15 shows an exemplary frame and slit structure including symbols and OFDM cells.

Fig. 16 illustrates the coherence bandwidth and the differential modulation in one pilot symbol.

Fig. 17 depicts the differential modulation between two P1 symbols.

Fig. 18 shows two symbols 1k with guard interval 1/1 and differential modulation between symbols.

Fig. 19 shows the calculation of sums of received energy from one or more pilot symbols.

Fig. 20 shows the transmitter.

Fig. 21 shows a receiver.

Fig. 22 is a flowchart that shows the steps that can be performed by a receiver.

Fig. 23 is a graph of autocorrelation / cross-correlation between pilot sequences and their frequency shift version.

Fig. 24 is an enlarged version of the graph of Fig. 23 showing the low cross-correlation range for frequency shifts.

Fig. 25 is a graph that shows the envelope amplitude of the first pilot symbol signal.

Fig. 26 is an enlarged version of the graph of Fig. 25.

Fig. 27 shows an example of a 2k symbol (P1) according to an embodiment.

Fig. 28 shows a sync symbol P1 having two successive OFDM symbols (P1a and P1b) that have the same FFT size according to an embodiment.

Fig. 29 shows an example P1 in which each of the pulses P1a and P1b has been divided into two parts according to an embodiment.

- Fig. 30 is a diagram of the operation of a part of a receiver correlator according to an embodiment.

Fig. 31 is a diagram of the operation of a part of the receiver correlator according to an exemplary embodiment.

Fig. 32 shows the steps of a detection sequence according to an embodiment.

DETAILED DESCRIPTION OF THE INVENTION [0009] The following description and figures illustrate embodiments of the invention that include features from the independent claims, as well as other embodiments of related methods and devices that do not include all of the features of the independent claims, but are useful for a better understanding the claimed invention.

[0010] Embodiments are related to service discovery and channel search in digital broadcasting networks. Relatively fast detection of services is desirable from the user's point of view. Of course, when the receiving device is used for the first time, a blind service discovery / channel search is performed. Moreover, when the terminal is turned off and moved to another location, a new blind search is also performed. Typically, the mobile TV application also performs searching for channels in the background from time to time to detect possible new services. Blind service detection should last only a few seconds, so as not to annoy the end user and allow frequent re-scanning.

[0011] The challenges regarding conventional detection of digital video transmission services include the following. The DVB-H standard offers great flexibility in terms of signal throughput, FFT sizes, guard compartments, internal modulations and the like. Operators can use offsets for the DVB-H signal, i.e. the signal is not at the nominal center frequency of the channel, but is shifted by a certain amount. Different countries use different bandwidths of channel and signal raster. TPS (Transmitter Parameter Signaling) is included in the standard to help synchronize the receiver and search for channels. Unfortunately, the receiver must know several parameters before it can decode TPS information. Signal bandwidth, frequency shift, the FFT size and the guard interval must be known before the TPS can be decoded. Most UHF channels do not include the DVB-H service. Channels other than DVB-H are detected by trial and error (trying to get a lock with all mods), and this consumes a lot of time. Time to detect services other than DVB-H in fact mainly determines the achievable speed for channel search, because usually most channels are empty or contain a service other than DVB-H.

[0012] An example of a calculation for blind service detection is as follows: number of channels in UHF 35, (Channels 21-55, 470-750 MHz); the number of frequency shifts 7 (-3/6, -2/6, -1/6,

- 5 0, +1/6, +2/6, + 3/6 MHz); number of signal bands 3 (6MHz, 7MHz, 8MHz. 5MHz is a separate case only for receivers in the USA); the number of FFT 3 sizes (2K, 4K, 8K); number of guard compartments 4 (1/32, 1/16, 1/8 and 1/4); and average time to TPS decoding for one 120ms module (2K 50ms, 4K 100ms, 8K 200ms). The numbers are examples.

[0013] The obtained time for blind service discovery would be: * 7 * 3 * 3 * 4 * 120 ms = 1085.4 seconds = 17.64 minutes.

[0014] According to the embodiments, various methods can be used to shorten the time it takes to perform channel search / service discovery. The basic assumption of various methods is the introduction of a part of the signal (e.g. symbol (symbols) of initialization / synchronization), which has known properties and remains the same with different modes of operation of digital video transmission. Therefore, a known part of the signal can be decoded without having to resort to trial and error methods. The known signal part contains parameters for the rest of the signal; therefore, the rest of the signal can be decoded without trial and error methods after decoding the known part. The known signal part comprises a subset of available subcarriers and their modulation. The combination of predefined subcarriers (number of subcarriers) and their modulation is selected such that for the combination to be unique for each FFT offset-size pair (or unique only for different FFT sizes) and which combination can be used to identify the signal as a desired signal for digital video transmission. Furthermore, channels containing digital video transmission services can be effectively detected using a known signal portion. If the known constant part is not found on the basis of the signal under test, then the signal will be considered as a signal other than for digital video transmission or an empty channel, and the receiver can immediately proceed to the next channel / frequency. Thus, detecting channels other than for digital video and empty transmissions becomes relatively fast.

[0015] Fig. 1 illustrates a corresponding digital broadband broadcast system 102, in which one or more illustrative embodiments may be implemented. Systems such as the one illustrated here may use digital broadband broadcasting technology, for example, DVB-H (Digital Video Broadcast - Handheld) or next generation DVB-H. Examples of other digital broadcasting standards that can be used by digital broadband broadcast system 102 include digital terrestrial transmission (DVB-T), digital broadcasting with service integration for terrestrial broadcasting (ISDB). -T, Integrated Services Digital Broadcasting - Terrestrial),

- 6 Digital Audio Broadcasting) and digital radio broadcasting on long, medium and short waves (DRM, Digital Radio Mondiale). Other standards and digital emission techniques, currently known or later developed, may also be used. The objects of the invention may also be used in other multi-carrier digital transmission systems, such as, for example, T-DAB, T / S-DMB, ISDB-T and ATSC, proprietary systems such as Qualcomm MediaFLO / FLO, and non-traditional systems, such as 3GPP MBMS (Multimedia Broadcast / Multicast Services) and 3GPP2 BCMCS (Broadcast / Multicast Service).

[0016] Digital content may be created and / or provided by digital content sources 104 and may include video signals, audio signals, data, and so on. The sources 104 of digital content may provide content for the digital television transmitter 103 in the form of digital packages, e.g. Internet Protocol (IP) packets. A group of related IP packets sharing a unique IP address or other source identifier is sometimes described as an IP stream. The digital television transmitter 103 can receive, process and transmit for transmission a plurality of streams of digital content from a plurality of digital content sources 104. In various embodiments, data streams of digital content may be IP streams. The digitized content may then be transferred to a digital television tower (or other physical transmission component) for wireless transmission. Ultimately, terminals or mobile devices 112 can selectively receive and consume digital content from digital content sources 104.

[0017] As shown in Fig. 2, the mobile device 112 may include a processor 128 connected to a user interface 130, memory 134 and / or another storage device, and a display 136 that may be used to display video content, service guide information. and the like for a user of a mobile device. The mobile device 112 may also include a battery 150, a speaker 152, and an antenna 154. The user interface 130 may further include a keyboard, a touch screen, a voice interface, one or more arrow keys, a joystick, a virtual glove, a mouse, a trackball, a touch screen or the like.

[0018] The instructions performed by the computer and the data used by the processor 128 and other components in the mobile device 112 can be stored in the memory 134 read by the computer. The memory may be implemented using any combination of read-only memory modules or random access memory modules, optionally including both volatile and non-volatile memory. The software 140 may be stored in memory 134 and / or data storage device to provide instructions to the processor 128 to allow various functions to be performed by the mobile device 112. Alternatively, some or all of the instructions performed by the computer from the mobile device 112 may be implemented in hardware or software (not shown).

[0019] The mobile device 112 may be configured to receive, decode and process digital broadband broadcast transmissions that are based, for example, on a DVB (Digital Video Broadcast) standard such as DVB-H or DVB- T, through a specific DVB 141 receiver. The mobile device can also be equipped with other types of receivers for digital broadband broadcast transmissions. In addition, the reception apparatus 112 may also be configured to receive, decode and process transmissions via the FM / AM radio receiver 142, the WLAN transceiver 143 and the telecommunications transceiver 144. In one aspect of the invention, the mobile device 112 may receive radio data stream messages ( RDS, radio data stream).

In the DVB standard example, one DVB 10 Mbit / s transmission may have 200 audio program channels of 50 kbit / s or 50 video program (TV) channels of 200 kbit / s. The mobile device 112 may be configured to receive, decode and process the transmission based on the digital terrestrial television standard for reception by portable devices (DVB-H) or other DVB standards, such as DVB-MHP, DVB for satellite broadcasting (DVB-S) or DVB for terrestrial broadcasting (DVB-T). Similarly, other digital transmission formats can alternatively be used to provide content and information about the availability of complementary services, such as the ATSC (Advanced Television Systems Committee), NTSC (National Television System Committee), ISDB-T (digital broadcasting with the integration of services intended for terrestrial broadcasting), DAB (digital broadcasting), DMB (transmission of digital media transfers), FLO or DIRECTV. In addition, digital transmission can be quantized in time, as in DVBH technology. Time quantization can reduce the average power consumption of the mobile terminal and can enable smooth handover. Time quantization entails sending data in series with a higher instantaneous bit rate compared to the required bit rate if the data were transmitted using a traditional streaming mechanism. In this case, the mobile device 112 may have one or more buffer memories for storing decoded temporally transmitted transmission prior to presentation. Time quantization can reduce the average power consumption of the mobile terminal and can enable smooth handover. Time quantization entails sending data in series with a higher instantaneous bit rate compared to the required bit rate if the data were transmitted using a traditional streaming mechanism. In this case, the mobile device 112 may have one or more buffer memories for storing decoded temporally transmitted transmission prior to presentation. Time quantization can reduce the average power consumption of the mobile terminal and can enable smooth handover. Time quantization entails sending data in series with a higher instantaneous bit rate compared to the required bit rate if the data were transmitted using a traditional streaming mechanism. In this case, the mobile device 112 may have one or more buffer memories for storing decoded temporally transmitted transmission prior to presentation. if the data was transmitted using the traditional streaming mechanism. In this case, the mobile device 112 may have one or more buffer memories for storing decoded temporally transmitted transmission prior to presentation. if the data was transmitted using the traditional streaming mechanism. In this case, the mobile device 112 may have one or more buffer memories for storing decoded temporally transmitted transmission prior to presentation.

[0021] In addition, an Electronic Service Guide (ESG) can be used to provide information about a program or service. In general, the Electronic Service Guide (ESG) enables the terminal to communicate which services are available to end users and how to access services. ESG covers independently the existing parts of the ESG fragments. Traditionally, ESG fragments include XML and / or other binary documents, but recently included a wide range of items such as, for example, the SDP (Session Description Protocol) format, text file or image. ESG snippets describe one or several aspects of the currently available (or future) service or broadcast program. Such aspects may include, for example: description with any text, timetable, geographic availability, price, method of acquisition, genre and complementary information, such as image preview or clips. Audio, video and other types

- 8 data covering ESG fragments can be transmitted through different types of networks according to many different protocols. For example, data can be transmitted through a set of networks usually referred to as "Internet" using protocols from the Internet protocol suite, such as the Internet Protocol (IP) and the User Datagram Protocol (UDP). Data is often transmitted via the Internet addressed to a single user. However, they can be addressed to a group of users, which is commonly known as multicasting. In the case where the data is addressed to all users, this is referred to as broadcast transmission.

[0022] One of the methods of data transmission of data is the use of an IP data transmission network (IPDC). IPDC is a combination of digital broadcast and Internet protocol. With such an IP-based broadcast network, one or more service providers may provide various types of IP services including magazines, radio and internet television. These IP services are organized in one or more media streams in the form of audio, video and / or other types of data. To determine when and where these streams occur, users refer to the Electronic Service Guide (ESG). One type of DVB is terrestrial digital television for reception by portable devices (DVB-H).

[0023] DVB transport streams provide compressed audio and video data to the user via third party delivery networks. MPEG (Moving Picture Expert Group) is a technology by which encoded video, audio and data in a single program are multiplexed, with other programs, to a transport stream (TS). The TS is a packet data stream, with fixed length packets including the header. Each of the individual program, audio and video elements is transferred in packets having a unique packet identification (PID). To enable the receiving device to locate various elements of a specific program in the TS, the program-specific information (PSI) that is included in the TS is provided. In addition, the TS includes additional information about the service (SI), a set of tables belonging to the private syntax of the MPEG section. This allows the receiving device to correctly process data contained in the TS.

[0024] As mentioned above, ESG fragments can be transported via IPDC over a network, such as, for example, DVB-H to target devices. DVB-H may include, for example, separate audio, video and data streams. The target devices must then re-arrange the order of the ESG fragments and put them together in useful information.

[0025] In a typical communication system, a cell may define a geographical area that may be covered by a transmitter. The cell can be any size and

- 9 may have neighboring cells. Fig. 3 schematically illustrates an example of cells, each of which may be covered by another transmitter. In this example, Cell 1 represents a geographical area that is covered by a transmitter in the case of a communication network. Cell 2 is next to Cell 1 and represents the second geographical area that may be covered by another transmitter. Cell 2 may be, for example, another cell in the same network as Cell 1. Alternatively, Cell 2 may be in a different network than Cell 1. In this example, Cells 1, 3, 4 and 5 are neighboring cells for Cell 2.

[0026] According to one or more embodiments, data used in channel lookup and service discovery are signaled by means of symbols at least at the start of the data frame carrying the multimedia and other data for the services. In other embodiments, one or more of these symbols may also be inserted into a data frame. In addition, one or more of these symbols may be introduced at the beginning and / or within a super-frame of two or more data frames.

[0027] In one embodiment, the symbols comprise a first symbol that can be used to identify if the signal is of the desired type. In addition, the first symbol can be used to detect the offset from the center frequency of the radio channel. Symbols may contain a second symbol that can carry data on modulation parameters that are used in subsequent data symbols. In a further embodiment, the symbols comprise a third symbol that can be used to estimate the channel.

[0028] Fig. 4 shows a frame and a super-frame of symbols, synchronization symbols, S1-S3, used to search channels and detect services, and data D.

[0029] In various digital broadcasting networks, the multi-carrier signal may be set relative to the channel raster such that the center frequency of the signal (SCF) coincides with the center-center frequency (CCF) or may be shifted from the center frequency of the channel. The center frequency of the signal can be shifted due to reasons for using the frequency spectrum (e.g., interference from an adjacent channel). For the first signal, not all available subcarriers are used. In various embodiments, the subcarriers that are selected for the first symbol may be evenly spaced and may be symmetrically aligned with the center frequency of the channel or the offset frequency of the signal.

[0030] Fig. 5 shows how the center frequency of the signal can coincide with or be shifted relative to the center frequency of the channel (CCF). In Figure 5 SCF A and its corresponding CSF converge, SCF B and SCF C are shifted relative to corresponding CSF. The rectangles in Fig. 5 illustrate the subcarriers selected for the first symbol among the available subcarriers. In the case of SCF A, SCF B and SCF C, the selected subcarriers are centered around the respective SCF. However, the selected subcarriers for SCF D are centered around CCF, unlike SCF.

[0031] For the first symbol used to search channels and detect services, the subcarriers may be chosen such that they can be found independently of the offset. In the first symbol, a fast Fourier transform (FFT) can be used. The FFT constant can be selected from the available FFT sizes, which in current digital video transmission systems include 2K, 4K, 8K, but may also include 1K at the bottom and 16K at the top. In one embodiment, the lowest available FFT is used. In addition, the first symbol may use a guard interval (GI) that can be selected from the GI used for symbols that carry data. The first symbol may, in one embodiment, have no guard interval.

[0032] The number of subcarriers for the first symbol may be less than half of the available subcarriers.

[0033] When the first symbol is used for channel shift signaling, carriers may be modulated using Binary Phase Shift (BPSK) or Quadrature Phase Shift Keying (QPSK). The chosen pilot pattern may vary for different channel shift values, wherein the pilot pattern and the subcarrier modulation may be selected in one embodiment so that the different pilot patterns are orthogonal to each other and maximally different from each other, thus allowing interference immunity when detecting. In one embodiment, the different pilot patterns can only signal the size of the FFT, and the frequency offset is detected by detecting the shift from the nominal central frequency.

[0034] For the second (and third if present) symbol, the full signal bandwidth (essentially all available carriers) can be used. In an embodiment, the second (and third) symbol may use the same FFT size and the guard interval as the first symbol. In certain embodiments, not all of the available subcarriers are used for the second (and third) symbol. In one embodiment, the second and third symbols may have the same subcarrier as the pilot subcarriers, and in a further embodiment have additional subcarriers used as remotes. In one embodiment, the second symbol also carries signaling data, and further may carry forward error correction (FEC) data for signaling data.

[0035] According to the embodiments, a portion of the signal (e.g., an initialization / synchronization symbol) is introduced, which has known properties and remains the same with different modes of digital video transmission. The known signal part contains parameters for the rest of the signal; therefore, the rest of the signal can be decoded without trial and error methods after decoding the known part. Furthermore, channels containing digital video transmission services can be effectively detected using a known signal portion. If the known constant part is not found on the basis of the tested signal, then the signal will be considered as

- 11 signal other than for digital video transmission or an empty channel, and the receiver can immediately proceed to the next channel / frequency.

[0036] Fig. 6 is a block diagram illustrating the steps performed by a receiver according to at least one embodiment. The frequency synthesizer in the receiver is programmed to the nominal center frequency of the channel, according to the channel raster, as shown at 602 for signal reception on the channel. An attempt is made to determine if the received signal is of the desired type and whether offset is applied by comparing the received signal with a stored set of known signals, as shown at 604. If a match is found, the signal is determined as the desired type and the FFT offset and size can be determined. for the signal. A determination is made regarding whether a fit has been detected as shown at 606. If no match is detected,

Otherwise, if the match is detected, then following the "yes" branch of 606 follows, a specific frequency offset is used to re-program the frequency synthesizer as shown at 610. The next sync symbol is demodulated to detect the modulation parameters for data symbols as shown at 612. Finally channel estimation and correction and data demodulation are performed as shown at 614.

[0038] In the event that re-programming the frequency synthesizer takes a relatively long time, the receiver can wait for the next set of initialization / synchronization symbols and demodulate the modulation parameters from that set.

[0039] Fig. 7 shows an example of the size of the pilot bandwidth relative to the signal bandwidth and the channel raster bandwidth. In an embodiment, the first symbol is a pilot symbol for coarse frequency and time synchronization. The bandwidth of the pilot symbol is smaller than the actual data symbol. eg in the case of an 8MHz data symbol, the pilot symbol would have a width of 7MHz. The center frequency of the pilot symbol may be the same as the frequency for the data symbols, i.e. when using an offset for data symbols, the offset may also be used for the pilot symbol. With a smaller pilot bandwidth, the RF portion of the receiver can be programmed to the nominal center frequency of the channel during the initial synchronization phase and will still be set to receive the entire pilot band bandwidth. Without a smaller bandwidth of the pilot symbol, the receiver's RF channel selection filter would filter out a portion of the pilot symbol.

[0040] In an embodiment, the pilot symbol may use a selection of known (fixed) FFT and guard interval. Also the number of pilots used can be different than in

- 12 data symbols, i.e. some of the pilots can be blacked out, eg 256 pilots can be used. The pilots can be modulated using a known sequence.

[0041] Fig. 8 illustrates a rare pilot interval in a pilot sequence for a pilot symbol. The "unique structure" of the modulation sequence for the pilot pattern can be known by the receiver. In addition to modulation, the subcarriers in the pilot symbol may have different gain levels as illustrated in Fig. 8.

[0042] Fig. 9 is a block diagram illustrating steps performed by a receiver to perform frequency domain correlation to detect the coarse offset used. The part of the radio frequency of the receiver (frequency synthesizer) is programmed to the nominal center frequency (according to the channel raster) of the channel as shown at 902.

[0043] The FFT is calculated using a predetermined FFT size as shown at 904. The pilot symbol width is less than the channel bandwidth. Therefore, the FFT is able to capture the pilot symbol even when the initial setting for the frequency synthesizer is bad due to the offset.

[0044] The frequency offset is detected based on the shift of the pilot synchronization symbol in the frequency domain, as shown at 906. If no frequency correlation is detected, the signal is not a digital video transmission signal and the channel search may proceed to the next channel.

[0045] The offset is compensated by reprogramming the receiver's frequency synthesizer, as shown at 908. The next sync symbol is demodulated to detect the modulation parameters for the data symbols as shown at 910. An estimation and channel correction is performed based on the channel estimation symbol, as shown at 912, then the data is demodulated, as shown at 914. In an embodiment, the receiver can wait for the sync symbol in the next set of synchronization symbols, thus allowing reprogramming of the frequency synthesizer to the center frequency of the signal.

[0046] Various pilot sequences (unique structures) may be used based on the offset used. For example, if 7 offsets are possible (± 3 / 6MHz, ± 2 / 6MHz, ± 1 / 6MHz, 0), you can enter 7 different pilot sequences. Several methods can be used to build a pilot sequence, including, but not limited to, a pseudorandom sequence, inversion of every other element, center carrier amplification, and the like. According to an embodiment, the receiver performs a correlation in the time domain to detect the pilot sequence used, and thus the offset applied. The unique structures may be used according to one or more embodiments directed to perform correlation in the time domain. In contrast, in embodiments in the frequency domain, the offset can be detected using

- a sliding correlator in the frequency domain, i.e. a single unique structure can be used. In addition, information such as the FFT size for embodiments in the frequency domain may be encoded if different unique structures are used, e.g. for different FFT sizes. Then, you can correlate in the frequency domain with several unique structures. In an embodiment, if several unique structures are used, the received unique structure can be compared simultaneously with several stored unique structures. The received pilot sequence may be translated in the frequency domain gradually through the channel bandwidth, wherein a high correlation signal is generated when the pilot sequences converge.

[0047] Fig. 10 is a flowchart that illustrates steps according to an embodiment of the service detection correlation in the time domain. The part of the radio frequency of the receiver (frequency synthesizer) is programmed to the nominal center frequency (according to the channel raster) of the channel as shown at 1002.

[0048] In one embodiment, the correlation of the received pilot sequence is performed in the time domain with known pilot sequences to detect the applied offset as shown at 1004. For example, if seven offsets are used, seven different pilot sequences (unique structures) are defined. . Each coarse offset corresponds to a specific unique pilot sequence structure. Based on the correlation, the unique structure used, i.e. the offset applied, can be detected. The pilot sequence will be in the nominal central channel frequency (according to the channel raster). In one embodiment, the set of pilot symbols is defined such that each corresponds to a pair frequency shift-size FFT, wherein both the offset and the size of the FFT can be detected based on the detected correlation.

[0049] The frequency offset is detected based on the identified unique pilot sequence structure as shown at 1006. If neither of the pilot sequences shows a correlation, then the signal is not a digital video transmission signal and the search may proceed to the next channel.

[0050] The offset is compensated by reprogramming the receiver's frequency synthesizer as shown at 1008. The next sync symbol is demodulated to detect the modulation parameters for the data symbols as shown at 1010. A channel estimation and correction based on the channel estimation symbol is performed, as shown at 1012, and then the data is demodulated, as shown at 1014. In one embodiment, the receiver can wait for the next set of synchronization symbols to allow reprogramming of the frequency synthesizer.

[0051] After detecting the shift and re-programming the frequency synthesizer, the second symbol (i.e., the symbol following the pilot symbol) may use a fixed FFT selection and a guard interval, but would use the full signal bandwidth. The second signal can

- 14 then contain specific information about the modulation parameters for the subsequent data symbols. In a further embodiment, the second symbol may use the FFT, which is signaled in the first symbol.

[0052] An optional third symbol may be inserted before the data symbols to facilitate channel estimation.

[0053] Fig. 11 shows an example of a pilot / signal symbol sequence. The pilot symbol 1102 and the signaling symbols 1104 and 1106 may be repeated at a transmission frequency sufficiently, e.g. every 50ms, to enable the detection and synchronization of the signal as quickly as desired. The first pilot symbol 1102 is used for coarse frequency and time synchronization, and in addition can also carry information about the FFT size for the following symbols. FFT, guard interval and modulation are fixed for the first symbol. In one embodiment, the second symbol 1104 includes the same pilot subcarrier as the first symbol, but may have additional subcarriers that are used as the pilot subcarrier. The second signaling symbol also carries signaling data including the FFT size, guard interval and modulation parameters. The third signaling symbol contains even more pilots that are used for channel estimation and accurate time coordination.

[0054] Modulation parameters for data symbols (such as constellation, QPSK vs. 16QAM vs. 64QAM) can often be changed because repeated signaling symbols carry information about selected parameters.

[0055] Fig. 12 is a block diagram illustrating the steps of a method performed by means of a transmitter. A symbol sequence is created that includes the pilot symbol configured to carry the coarse frequency and time synchronization information as the first symbol followed by the next signal symbol configured to carry modulation parameters as a second symbol followed by a plurality of data symbols as shown at 1202. In one embodiment, a third signaling symbol may follow the second signal symbol. The symbol sequence is then broadcast in the broadcast channel at the pilot bandwidth that may be less than the bandwidth of the data signal, which may be even smaller than the bandwidth of the broadcast channel channel raster,

[0056] Fig. 13 and Fig. 14 illustrate the relationship between P1, P2 and data symbols (i.e., OFDM symbols) as an example. From Figures 13 and 14, it can be seen how the data has been divided for the duration of P2 symbols and data. The data packets can be placed directly after the P2-ni packet, both are transferred in the "data symbols".

[0057] Fig. 15 shows an exemplary frame and slit structure. In Figure 15, frame 1502 may include one or more slots 1504. For example, frame 1502 includes slit 1506 through slit 4 1512. Each slit 1506-1512 may include several symbols

- 15 OFDM (orthogonal multiplication in the frequency domain), usually from several symbols to several dozen symbols. Services are assigned to these slots, so that one or more slots are used for the service. For example, the slot 1506 may include a number of OFDM symbols 1514 through 1524. In addition, each OFDM symbol may comprise a plurality of OFDM cells. For example, the OFDM symbol 1514 contains OFDM 1526 cells through 1534.

[0058] Embodiments are directed to the initial detection of services in the next generation system of terrestrial digital transmission intended for terrestrial broadcasting (DVB-T2). The DVB-T2 system may comprise a preamble, which is intended to efficiently identify available T2 signals. The preamble should not use too much capacity, but should be compatible with the different sizes of the fast Fourier transform (FFT) (2k, 4k, 8k, 16k and 32k). Minimization of the overhead led to the use of 2k symbols (P1) for each FFT size and signaling of the actual FFT size in this symbol by modulation of carriers using different pseudo random binary sequences (PRBS). To determine the FFT size of the following symbols, the receiver detects the modulating PRBS. This PRBS also indicates the frequency shift in the whole number (the DVB-T2 signal can be shifted by +/- 1/6, +/- 2/6, +/- 3 / 6MHz compared to the nominal central frequency). In summary, the P1 symbols are used in the initial scan to: (1) detect the presence of the T2 signal; (2) estimate the frequency offset; and (3) detect the FFT size used.

[0059] After the initial scanning, the P1 symbol can not be used during normal data reception or connection switching, because the parameters conveyed by P1 (i.e., the FFT size and frequency offset) remain constant. Regarding the connection switching, these parameters are the same between radio frequency (RF) channels or they are signaled before the connection is switched (eg in the program's relevant information / service information (PSI / SI) according to ETSI EN 300 468 Digital Video Broadcasting (DVB); Specification for Information Information (SI) in DVB systems). However, P1 can be used during normal data reception to, for example, detect the beginning of the frame or improve the synchronization and channel estimation algorithms.

[0060] The detection of P1, and thus the detection of the DVB-T2 signal, is based on a guard interval correlation (GIC). In the GIC, the guard interval is correlated with the end of the symbol. The peak level in GIC indicates a potential DVB-T2 signal that can be verified based on the P2 symbol. The first problem is that the guard interval should be long to provide interference-proof detection (i.e., a long guard interval provides a higher signal-to-noise ratio). However, a longer guard interval, and thus a longer P1, reduces data capacity.

[0061] Since P1 is the first symbol to receive, there is usually no prior knowledge of channel conditions. Therefore, the P1 symbol should include some measures to prevent channel disturbances. In practice, this would mean using, for example, additional pilot carriers for channel estimation or differential modulation between subcarriers.

[0062] Due to the smaller size of the FFT, the spacing between the carriers of the P1 symbol may not be as dense as in the following data symbols (e.g., 2k for P1 and 32k for data). For the successful detection of PRBS in P1, the coherence bandwidth of the channel should be less than the space between subcarriers of the 2k symbol. However, the network can be designed for 32k mode, and long single frequency network (SFN) latencies can provide much higher frequency selectivity.

The received complex signal signal at the carrier k index can be expressed as rk = hksk + nk, where sk is a data symbol (e.g., using square quad keying (QPSK)), hk is the channel response at the k index k, and nk is the noise term.

[0064] In coherent demodulation, hk is first estimated by means of pilots, and then the operation of the channel is compensated by, for example, dividing hands by estimated hk.

[0065] If you consider DVB-T2 and the P1 symbol, there will be no pilots to estimate hk. Therefore, non-coherent demodulation will usually be used, without channel estimation. This can be done using differential demodulation (e.g., differential binary square keying (DBPSK), where information is encoded to the phase difference between two adjacent carriers. These two neighboring carriers can be expressed as rk = hksk + nk and rk + 1 = hk + 1sk + 1 + nk + 1. The transmitted symbol can be decoded from the phase difference between the two received carriers: rk + 1-rk = hk + 1sk + 1-hksk + n.

[0066] Fig. 16 illustrates the coherence bandwidth and the differential modulation in one OFDM pilot (PI) symbol. It is assumed that the channel response phases hk and hk + 1 are approximately the same as shown in the upper graph of Fig. 16. However, in a highly frequency-selective channel (e.g., the bottom graph of Fig. 16) the correlation between adjacent channel responses is relatively low. This makes the use of differential modulation between carriers unfeasible.

[0067] The width of the coherence band (i.e., the bandwidth, where the channel response is W - highly correlated) can be approximated by means of ^coil r_d where τd is the fading of the channel delay. The coherence bandwidth of the channel should be smaller than the distance between carriers for DBPSK applications between carriers. The FFT P1 size is 2k, and the carrier space in the 8MHz channel is 4.46kHz. Of these carriers, it is used

- every third or ninth carrier. Therefore, the real distance between carriers can be as much as 40.1 kHz. On the other hand, the delay blur in a large SFN network can be 448 μs (16k mode with% guard interval), resulting in a coherence bandwidth of 2.2 kHz.

[0068] According to the associated method, two P1 symbols are used, e.g. symbol 1k with GI = 1/1. Both symbols are used separately in the GIC. When GI = 1/1, the entire duration of the symbol can be used in GIC.

[0069] According to a related method, differential modulation is applied between two P1 symbols as shown in Fig. 17. Since the differential modulation is now performed according to the sub-carriers, there are no requirements for the coherence bandwidth. (Alternatively, the first P1 symbol may be used to estimate the channel, which would allow coherent demodulation for the second P1 symbol).

[0070] The time interval of two P1 symbols is relatively short such that the channel does not change from the first symbol to the second symbol. Therefore, according to one or more embodiments, the differential modulation can be performed in the time domain between carriers having the same number of carriers.

[0071] Exemplary embodiments also support a mobile reception. According to the embodiments, the coherence time of the channel is longer than the duration of the two symbols P1. This makes the correlation between rk (1) and rk (2) high. The coherence time of a channel can be approximated by means of^coli ~ where Fd is the channel Doppler scattering and is given as

<img file="PL2215794T3_D0001.tif" />

where v is the speed of the receiver, c is the speed of light (3 * 10^AND8m / s), and Fc is the carrier frequency. If v = 120km / h and Fc = 666MHz, then Fd = 74Hz and Tcoh = 13.5ms, which is much longer than the duration of the P1 symbol (eg 280μs).

[0072] According to one or more embodiments, synchronization of the P1 symbol can be improved. The P1 symbols may have a guard interval of 1/1, which would improve the symbol synchronization and maximize the length of the guard interval correlation with the overhead. The P1 symbols may use 1k FFT, which would reduce the overhead compared to the two 2k symbols.

[0073] Correction of the guard interval (GIC) is the basic method of synchronization in orthogonal frequency reproduction (OFDM) symbols. Since GI is a cyclic copy of the last part of the real OFDM symbol, the receiver is able to find the beginning of the OFDM symbol by detecting this correlation. In practice, the receiver continuously correlates two blocks of the received signal, which are separated by N samples (N is the FFT size, as well as the number of data samples). The correlation peak level is detected at the correct position.

[0074] Fig. 18 shows two symbols 1k with guard interval 1/1 and differential modulation between symbols. As can be seen, the guard interval 1/1 means that the GI and the data part have the same length and the samples are also equal. Equivalently, it can be said that the symbol 1/1 has two equal symbols without a guard interval.

[0075] Due to the differential modulation, the consecutive symbols, P1 and P1 ', are different, which means that a normal GIC should be used in each P1 symbol. However, the length of the correlation is doubled compared to the symbol GI 1/4 2k (1/4 * 2048 = 512), and correlations from the two symbols can be combined for further improvements. The GI 1/1 1k symbol is also desirable because the correlation of the guard interval does not match the data modes (2k, 4k etc.).

[0076] A further embodiment speeds up the initial scanning. It is desirable to quickly detect signals other than T2 so that the receiver can tune to the next frequency. This can be done by detecting zero carriers in the P1 symbol by: (1) calculating three sums (see Fig. 19, which shows the calculation of sums of energy received from P1 according to one or more embodiments) of received energy for carriers that belong to the subsets r3k , r3k-1 and r3k + 1, where r is the k-th carrier from the symbol (s) P1, ak = 1, 2, 3 ...; and (2) detecting the presence of the T2 signal by comparing the received energy in the three subsets; and (3) determination of the energy threshold (e.g. 5 dB below the highest); and (4) if only one sum exceeds the threshold value, a possible T2 signal is detected.

[0077] Fig. 20 shows a transmitter according to one or more embodiments. The first P1 is modulated according to the BPSK according to the reference sequence, and the second P1 is modulated as follows: if PRBSk = 0 bk, 2 = bk, i; if PRBSk = 1 bk, 2 = -bk, 1 (or vice versa), where PRBSk is the k-th element of PRBS, and bk, m is the transmitted symbol on the k-th carrier at the m-th symbol P1. The transmitter then connects the original reference sequence and delayed differential modulated sequence before introducing the inverse fast fourier transform (IFFT) and the guard interval. N refers to the size of the FFT.

[0078] Fig. 21 shows a receiver according to one or more embodiments. The receiver performs the reverse operation of the transmitter discussed above in connection with FIG. 20. This means that the receiver removes the guard interval from the Pi symbols (first and second pilot symbols), performs fast Fourier transform on symbols P1, and then demodulates the P1 symbols differentially to obtain estimation of the transmitted pseudo-random binary sequence. The receiver does not need to know the reference sequence.

[0079] FIG. 22 is a flowchart that shows the steps that can be performed by a receiver according to one or more embodiments. In the initial scan the receiver can be tuned to the nominal center frequency of the channel and can start looking for the P1 symbol. The following procedure can then be repeated with selected channels (and bandwidths) - but not necessarily with each frequency shift, because the P1 symbol may

- 19 be detected at the nominal center frequency irrespective of the offset applied.

[0080] The first task after choosing the bandwidth and nominal center frequency is to find the presence of the T2 signal. The symbol P1 can be found, for example, by the correlation of the guard interval that is resistant to the frequency shift. The use of the guard interval correlation also helps to detect the T2 signal, because the absence of the 2k symbol implies a channel other than T2.

[0081] The correlation of the guard interval is intended for situations where the channel delay deflection remains within the guard interval, which may not be the case for the P1 symbol in large scale SFN (e.g., with 32k mode). In this case, the delays longer than the delays especially of the guard interval, which are multiples of the useful symbol, cause a false correlation.

[0082] However, it should be noted that the time synchronization of symbols in the presence of a strong SFN echo is not just a P1-specific problem, because in any case the receiver should be able to synchronize to the correct path. The difference is that the P1 correlation has a higher noise level due to the shorter GIC window.

[0083] Coarse timing and frequency synchronization is obtained from the correlation of the guard interval. These are coarse estimates that apply to the P1 symbol itself and can be specified using the following symbols. It is assumed that these estimates are accurate enough to detect one of the PRBS patterns to detect the size of the FFT.

[0084] In the case of initial rapid scanning, channels that do not include a T2 signal should be rejected relatively quickly. The preamble structure according to the embodiments supports step-wise detection, where channels other than T2 can be confirmed by reading static L1 signaling.

[0085] The first elimination can be made by the correlation of the guard interval. The P1 signal can be repeated every frame (about 200 ms) and is quite resistant to SNR requirements, so testing two successive P1 positions can be sufficiently reliable to detect the T2 signal. It would take about 500 ms per RF channel. The receiver can then decide if any P1 symbol has been detected. If this was done on 39 UHF channels and even with 3 channel bandwidths, the total time used for scanning is approximately 58 seconds. It should be noted that an attempt to scan different bandwidth at the same time does not actually help, because channel rasters are different.

[0086] When a possible P1 symbol is detected, the receiver can perform coarse synchronization and FFT. The receiver can then use a rare carrier raster to distinguish between T2 signals and others. Thus, signals other than T2 may most likely be detected based on the first received P1 symbol.

[0087] The detection of the frequency shift is based on the detection of the shifted pilot pattern. Detection of frequency shift and FFT size can be separated by first applying power to the adopted pilot carriers to detect the correct offset and then calculating the correlation to five PRBS. On the other hand, PRBS can already be used when detecting the frequency offset. A rare carrier raster reduces the complexity of search algorithms.

[0088] Once the frequency offset is detected, the receiver may be tuned to receive data symbols. The next task is to detect the protection interval used to decode the P2 symbol. Since the P1 symbol does not carry any signaling information from the GI, the receiver can detect this by using normal OFDM symbols during the frame. The P2 symbol immediately after the detected P1 can not be decoded. However, there is enough time to detect the GI before the next frame, because you can apply the entire 200 ms frame duration. This adds another 200 ms until the acquisition of the signal, but this is most likely only with DETECTED T2 signals, not with each channel tested. Since the maximum number of parallel multiples is usually on the order of 7 to 8,

[0089] In the case where the frame duration can be configured, frame synchronization can be obtained by recognizing the next P1 symbol. Detected parameters from static L1 signaling in P2 symbols are then confirmed.

[0090] In one embodiment, the first P1 is used to estimate the channel, which is then used to balance the second P1. This re-applies the basic assumption of various embodiments, although the implementation differs. N refers to the size of the FFT.

[0091] According to the DVB-T2 standards, the symbols P1 and P2 are shown as a solution for initial scanning and broadcasting of signaling. According to embodiments, the difference module between the two P1 symbols can have advantages in highly frequency-selective channels.

[0092] As discussed above, the symbols P1 are used in the initial scan to: (1) detect the presence of a T2 signal; (2) estimate the frequency offset; and (3) detect the FFT size used. A possible way to estimate the frequency shift (and to some extent detect the presence of the T2 signal) is to use the comb domain, i.e. to use a subset of the available subcarriers in the OFDM symbol. Assume that L subcarriers are available (= FFT size with subtracted guard bands). In addition, assume that every third subcarrier is available for this pilot / synchronization application, so there will be L = ^ L / jJ + and active subcarriers for the synchronization signal. Mathematically combs can be represented by the sequence P (0), P (i), K, P (L'-1) bits. Here the bit P (k) informs

- 21 '0' indicates a subcarrier with no power, and '1' indicates a subcarrier containing a signal modulated by BPSK. The assumption is that when the operator uses the channel frequency offset, the comb is shifted accordingly. Thus, after reaching the time synchronization and frequency synchronization in fractions, the receiver can perform FFT and search for frequency shift in whole numbers. Here, the receiver can apply the received power to the adopted pilot carriers (i.e., the comb) and detect the frequency offset without demodulating the pseudo-random binary sequence. The correct frequency shift in whole numbers (= multiple of the total number of subcarriers) can then be detected by the presence of a relatively good fit with the offset comb and the measured power of the subcarrier signal. The size of the FFT (selected from for example 5 options) is then indicated by the selection of 5 BPSK Sm (0), Sm (1), K Sm (L'-1), for m = 1,2,3,4 or 5.

[0093] The frequency offset (after adjusting to its fractional part) is equal

L - \ - n adding constant offset n to subscripts. Sum^sandⁿand⁼ ^ P ^ Peak + n) then calculates £ = 0 number of collisions between the comb and its shifted version, and S (0) = N is equal to the number of N subcarriers in the comb. In order for the frequency shift detection in the whole number to work, the collisions of the collisions S (n), n / 0 should be small compared to the correct matching of N.

[0094] Ideally, the structure of the P1 signals should be such that it also supports other detection methods, thus providing the equipment constructors with freedom of choice. Another approach to the problem of detecting the presence of P1 signal is based on time domain correlation. To support this alternative approach, real signals £ -1

ΣP (k), S_m(k) ex.p (2n j \ f + (w + 3ł) 4f] /) should have good cross-correlation properties - not only for different m values, but also for different pairs (m, n), for different values combinations (FFT size, frequency offset).

[0095] Other properties required by the set of signals are the properties of the justified autocorrelation of the time domain and the properties of a reasonable peak-to-average power ratio (PAPR). Ideally, it should also be possible to quickly and effectively generate both combs and BPSK sequences without resorting to large look-up tables.

[0096] The embodiments refer to: 1) combs limited to every third subcarrier and 2) combs that contain approximately half of the other subcarriers, so the number of active subcarrles N should be approximately L / 6. With such assumptions shorter comb patterns / sequences with length L = -Ll3 ^ +1 are desirable.

[0097] According to embodiments, the binary sequence mo of the appropriate length is used to generate the comb, and the selected cyclic shifts of the same sequence m (now

- 22 interpreted as + 1 / -1 as opposed to 0/1) is used to generate 5 BPSK patterns.

[0098] Six bit patterns are specified, each containing r bits, not all zeros, which will be referred to as grains. The grains are then expanded to a sequence of 2^r -1 by using the recursive formula defined by the primitive polynomial of degree r. It should be noted that the same recursive pattern is used to create each of the 6 sequences. One of the sequences is signaled to determine the comb, and the remaining 5 determines the BPSK patterns by reinterpreting '0' as +1 and '1' as -1. Ideally then L '= 2^r -1. Different use cases and an alternative way of constructing the comb may also be used.

In the particular case of using DVB-T2, L = 1531 subcarriers can be used, i.e. L '= 511 = 2⁹-1, r = 9 and primitive polynomian feedback 1 + x^four x +⁹. An example of a set of grains contains 100,000,000 for combs and 000 110 101, 110 001 100, 101 111 101, 101 101 111, 111 100 111 (all interpreted as +/- 1s) for 5 BPSK standards. They are extended to the P and Sm sequences for m = 1, 2, 3, 4 and 5 by repeated use of recursive formulas P (k) = P (k-4) + P (k-9) (mod2) and Sm ( k) = Sm (k-4) * Sm (k-9), for k = 9.10, ..., 510.

[0100] The design criterion in the selection of grains is that while the obtained sequences are cyclic shifts of one another, the magnitude of the shift that is to be taken from one to the other should be relatively large. Likewise, the grains may be designed such that one of them can not be produced from the bit XOR comb sequence and the subsequent sequence by means of a short (e.g., below 45 positions) circular sweep.

[0101] If the number of available carriers L 'is not in form 2^r-1, but is relatively close to such a number, then the comb and sequences can be shortened by cutting off a small segment from the end of the tail of the sequence m or the pattern can be extended by cyclically repeating it for a relatively short time. In the above example, the number of subcarriers can be reduced from 1531 to 1507 by cyclically shifting the comb pattern, as well as the BPSK sequence by one position. To achieve this, 9-bit grains can be expanded to 10 bits by applying a recurrent equation once. After that, the first bit can be left, thus producing a 9-bit grain. Thus, grains 000,000 001 for comb and grain 001 101 010,100 011 000, 011 111 010, 011 011 110, 111 001 111 for BPSK sequences would be used instead of the above suggestions. Then the comb will start from 8 zeros, ie 24 empty subcarriers, and the P1 signal will be narrowed to 1507 more carriers. It should be noted that the role played by the available bandwidth is less important because in the use of a narrower band (e.g., 5 MHz), the subcarrier gap is also narrower and there is still room for approximately the same number of subcarriers.

[0102] An alternative way to generate the frequency domain comb is to use a square residue sequence (= QR sequences) that are known in the art.

- state of the art. The resulting comb shares collision statistics between shifted versions with a m-sequence comb. This alternative method has the advantage that the length of the QR sequence is a prime number p adjoining 3 modulo 4. Therefore, the set of available lengths is more flexible when using QR sequences. Cyclic shifted versions of the same sequence can also be used here to construct a BPSK sequence. However, generating relatively long QR sequences on a regular basis is more computationally demanding and, in practice, it may be necessary to use a relatively large look-up table.

[0103] According to at least one embodiment, the proposed P1 signals

510

PI (t) = y £ (k) S_m(k) exp (27rj [f + (n + 3k) &amp; f] t) " _{| Eggs} λ ς m + · "ιι · + are for m = 1,2,3,4 and 5. Here n stands for the integer part?

frequency offsets. It is counted as a multiple of the space between subcarriers, so in the proposed use case n = ± 37, ± 75, ± 112 correspond to frequency shifts ± 1/6, ± 1/3, ± 1 / 2MHz (note that fractions between the subcarriers are previously supported regardless of whether they are the result of a rounding error or a discrepancy of the clock between the receiver and the transmitter). On the other hand, the illustrated structure actually allows any integer values ​​n up to 134. Here, P and Sm for m = 1, 2, 3, 4 and 5 are sequences 511 of the length discussed above. These signals occupy 256 subcarriers in the range of 1531 consecutive subcarriers.

[0104] There are various other options for grains that work equally well. For example, each of the 6 m sequences can be cyclically shifted by the same amount without changing the correlation value. Exemplary grain values ​​work well when the integral part of the frequency offset is below 3 * 45 = 135. In this respect, cross-correlations among the sequence shift versions remain low. Searching the computer revealed other grain sets with equally good results. The possibility of even a slightly wider range of low correlation has been completely ruled out, but it is known that if n will be up to 3 * 51 = 153, such a small correlation range can not be achieved by this method, regardless of how exactly the grains are selected.

[0105] Interval in multiples of 3 allows relatively fast detection of the total part of the frequency shift because there are no collisions between the real comb and the tested version, unless the difference between the tested and the actual total offsets is equal to a multiple of 3. If this condition is met, the number the collision is 256, when there is a correct shift, otherwise the number of collisions is in the range of 119 ... 128, i.e. near the optimal pseudorandom half-way point 128. For a randomly generated comb with a similar structure (= limited to every third subcarrier) (= generally, on average one of the 6 subcarriers is "active"), the expected range (standard deviations +/- 2 from the expected value) for the number of collisions is between 104 and 144,thus, the use of the sequence m improves this by introducing variability into a narrower range.

[0106] The basic algebraic structure of the m sequence helps to ensure that almost all sequences so produced have in the justified way good PAPR properties (the exception is the use of the same seed for the comb and for the sequence) and in a justifiedly good autocorrelations in the time domain. Careful selection of grains also helps to ensure good cross-correlation properties among the versions of the shift of different sequences. In fact, non-trivial correlations are very close to zero as opposed to randomly fluctuating up to +/- 2SD level 32.

[0107] Fig. 23 is a graph of autocorrelation / cross-correlation between pilot sequences and their frequency shift version according to one or more embodiments.

[0108] Fig. 24 is an enlarged version of the graph of Fig. 23 showing a low cross-correlation range for frequency offsets.

[0109] Fig. 25 is a graph that shows the envelope amplitude of the first P1 signal (a single symbol calculated at a center frequency of 666 MHz and a carrier distance of 4464 Hz, sampled at 25 MHz to produce these numbers) according to at least one embodiment . The scale is chosen so that the average square amplitude is equal to one.

[0110] Fig. 26 is an enlarged version of the graph of Fig. 25. Together Fig. 25 and 26 show the validated PAPR properties of the set.

[0111] In the discussion of the sequence of BPSK and P1 that follows, F = GF (512) will mean a finite field of 512 elements, ag will be a primitive element F that satisfies the equation l + g^five+ g⁹= 0, so that the power of g^and pass through non-zero elements F when the exponent i takes the values ​​i = 0, l, ..., 5l0. It should further be noted that g^-and then it will be the root of the previous 1 + x return equation^fourx +⁹= 0. Let tr: F ^ GF (2) be a tracking function. The previous sequence is 0 / l and all its cyclic shifts are obtained as sequences ma (i) = tr (ag^i-1), for i = 1,2, K, 511 and α ^ 0. E (x) = (- 1) is saved^{tr (x)} and ω = e^lπJj511. Thus you can choose the elements aeF and β / eFj = 1,2,3,4,5, so that the comb of zeros and ones is obtained as P (i) = tr (ag^i-1) = (1-e (at '^-one)) l2 and that the BPSK sequences are obtained as Sj (i) = e (ejg^i-1). The P1 sequences are therefore obtained using the formula E1_s(i) = (the 'ag' ')) e (¡g' ')^{/ 2} [0112] We have the identity e (x + y) = e (x) -e (y) and sum

<img file="PL2215794T3_D0002.tif" />

(hereinafter referred to as equation (1) or sum (1)), when γ is a value other than zero, and so-called Gaussian sums

510 s (k,₇) = ^ e (yg (hereinafter referred to as equation (2) or sum (2)) that have a complex absolute value ^⁵¹² when both γ and k are non-zero and below when one of them, but not both, is zero.

[0113] At this point, it should be noted that the proposed comb corresponds to the option α = 1.

- [0114] Consider the number of collisions between the pattern P (k) = (1-e (ag)) l2 and its shifted version P (k + n), where n indicates the offset amount (at most ^ 112/3 ^ = 37). If we were to continue the pattern of this comb cyclically with the period of 511, then it would be possible to calculate the number of collisions. Let us denote the variable x = g^k and let's assume a typical convention, such that F * is a set of non-zero elements in field F. Then the number of collisions modulo 511 'is (so k + n is the calculated modulo 511)

<img file="PL2215794T3_D0003.tif" />

Here the first sum is equal to 511. Because t &lt;511, coefficients a, agn, a (1 + gn) are non-zero, and equation (1) informs us that all other sums are equal to -1 (adapting to this that the term e (0) = 1 is missing in totals). In total, it is obtained that the shifted comb has 512/4 = 128 collisions with a periodically extended comb. When we take into account the effect of the tail caused by the sum of k + n overflow&gt; 511, we will see the expected fall in the number of collisions. At n = 1 2, 3, 4, 6, 7, 8, 128 collisions occur and this number drops approximately linearly when n increases. When n reaches the maximum value of 37, the number of collisions is 125. The lowest collision value 119 is achieved by shifting n = 36. So with this crest, the number of collisions between the two shifting combs will be close to the ideal point halfway 128.

[0116] It is possible to calculate the cross-correlation between two P1 sequences (in the f domain, since according to Parseval's theorem, it does not matter whether this is done in the frequency or time domain) as ⁵¹⁰ i ('(Pl, ·, Pl, ·.) = y Σ (i - - Ą,) g') = X e ((¡- ¡.) x) - £ &lt;(«+ ¡- ¡.) x ) ² i = 0 ' ² &lt;xeF xeF, so the sum (1) informs us that this cross-correlation is equal to zero, provided that βΐβΐ is non-zero (in other words, two sequences differ) and that α- ()_{C. whereas,}(β is non-zero (in other words, two sequences do not complement each other bitwise.) A practical test for this is that two sequences of this type are orthogonal, their initial segments are different, and that the bit-by-XOR of their initial segments is different from the initial ones comb segment P.

[0117] As in the calculation of the number of collisions, the sequence is first cyclically expanded in the field f, the cross correlation between such expanded signal pair is calculated and more or less ignored short "tail", which is the sum of several pseudorandom terms and will not contribute significantly. Thus, the cross-correlation (domain f) between the P1 signal and the next P1 signal offset from the previous position is as follows

<img file="PL2215794T3_D0004.tif" />

- 26 (hereinafter referred to as equation (3)).

[0118] It should be noted here that the indices jij 'can be equal, i.e. we are also interested in the correlation between the sequence and its version of the offset. On the basis of equation (1), we can see that this main term is zero, unless one of the coefficients in square brackets is zero. Since n assumes values ​​in the range of about zero, we have to choose the coefficients βι, Κ, β5 so that discrete logarithms of base g only of coefficients as well as sums a + e1, K, a + es are as far away from each other (cyclic modulo 511) . Since there are a total of 10 elements of the field, the minimum separation between discrete logarithms can not be greater than ^ -511 / 10 ^ = 51. When choosing α = 1 = g⁰ of an exemplary construction, a small heuristic search is provided by the set used in the above discussion: e1 = g³³And g = e1 +¹⁸¹, β2 = ξ¹³⁵ And g = e2 +⁴⁹⁹, e3 = g²⁴⁵, α + β3 = g³⁹⁸, β4 = g³⁴⁹, a + β4 = g⁸⁵, β5 = g⁴⁴⁵, α + β5 = g²⁹⁶. Here, discrete logarithms create a list {33, 135, 245, 349, 445, 181, 499, 398, 85, 296} - the first five discrete logarithms determine the elements β1, Κ, β5, and the last five list discrete logarithms from elements α + β1 , Κ, α + β5. The smallest cyclic separation of 45 occurs here between 499 and 33, because 33-499 + 511 = 45. The next sequence of discrete logarithms, which also has the smallest cyclic separation of 45, is {33, 135, 233, 339, 447, 181, 499, 388, 286, 80}. It is not known whether there are options leading to even greater cyclic separation. Because 3 * 45 = 135 (subcarrier separations) is greater than 112, this is sufficient for our purposes.

[0119] These numbers explain the gaps in Fig. 23. There are no matches with offsets up to 44 in any direction, so the width of the zone close to zero in Fig. 28 is 2 * 44 + 1 = 89 carriers. It should be noted that separation 45 corresponds to terms with a negative sign in equation (3). The smallest cyclic separation corresponding to the positive sign term is 96 and occurs between pairs (445, 349) and (181, 85). This explains why all the closest side lobes are negative, as well as explaining the wider space of 2 * 96 + 1 = 193 carriers above the x axis.

510 [0120] Here we show how the constraint S (Y) = ^^is(Y) Y⁷ means that the autocorrelation of / = 0 of proposed signals remains at a low level at least for a discrete set of time shifts. The version of the time domain of the proposed signal P1 is Ρΐ₇(0 = / 1)₇(A :) i?²®^{(/ + ΔΔ /) ζ} where for convenience we can enable the shift k

frequency wfi allow for the distance Δf between two possible carriers of the signal P1 (= 3 times the space between subcarriers of the OFDM symbol 2k). Suppose we have an At time error, which is smaller than the guard interval. Then the correlator of the time domain sees

- 27 K_ω, p_one;&lt;₊A)&gt; = ^ J « _s(A:) Pl₇^ ') E^{2; !!}'^{(/ UV)}'IS'^{2? N}'^{(/ + And}'^{¥) (z + Ai}With period kk '= K'e-²'^ref^ Pl_j(K)²is^and'^{! I}^^l}k (hereinafter referred to as equation (4)).

[0121] Here, the coefficients K and K 'serve to normalize and include an increase in power as well as constants derived from DFT and integration. The absolute value of this term depends the same (up to scaling) only on the sum. Assume that At is of such magnitude that the product Af At = n / 511, for a certain integer n. This means that the time error is the total multiple of 1/511 of the common subcarrier period. So you can save =_m^nk. Considering that P1j (k + 1) = (1-e (ag^k)) / 2 depends only on the comb pattern (and not on the BPSK modulation), we see that with these time error values ​​the cross-correlation is equal to

510 (/&gt; l_s(Z), Pl, (/ + «/ (511 · Δ /))) = tf" ((1 - e yyT £ = 0 (hereinafter referred to as equation (5)).

[0122] The sums in equation (1) and (2) inform then that (forgetting the multiplier K "- its absolute value is independent of n), the sum has the value 256, when n = 0 (i.e. when there is no timing error), otherwise it has a value of ^ 512 «22.6

To sum up: with our symbols there is a relatively dense discrete set of time errors, which will lead to autocorrelation values ​​of about 10 dB below the synchronized value. While this is not final, it is highly suggested that the autocorrelation properties of our proposed signals are relatively good.

[0123] Again, the sums (1) and (2) are essential for our respect. When two different signals P1 - P1j and P1j 'are compared, the calculation that leads to equations (4) and (5) above will give this time

510 (? L_s(/) H_s. (/ + «/(511-A/)))-tf"^(le(agń)e((Ą-Ą.^AND&gt; '^l, k = Q [0124] Let us remind you that we work on the assumption that ()_{C. whereas,}(β- a. If here n = 0, then this sum has the value 0 based on the formula (1), otherwise we have two Gaussian sums, so based on the inequality of the triangle we can estimate

<img file="PL2215794T3_D0005.tif" />

In other words, with this discrete set of time errors, cross-correlations are at least 7 db below the ideal match of 256 * K '.

<img file="PL2215794T3_D0006.tif" />

[0125] Again, the sum (2) allows us to provide a relatively accurate estimate of envelope power at sampling moments of At = n / (511Af) for all n = 0.1, ..., 510. we have

510 k = 0 [0126] Since α ^ β /, we obtain zero at n = 0, and by the result of equation (2) from Gauss sum the sum in absolute value characters is limited from the top by 2X12. The total sampled boundary is the same as at most 1 Here the total signal energy is 256, so the average power is equal to 7256/2048 = 1/7. Thus, at this sampling frequency, the envelope's maximum to average power ratio is at most

There is a general limitation that informs us that the continuous peak-to-average envelope power ratio is at most (2ln (511) + 1.132 + 4/511) 78 in the worst case (and in practice, probably a little better).

[0127] As discussed above, a relatively fast recognition and synchronization of OFDM signals can be obtained by using special synchronization signals or specially designated symbols, according to various embodiments. For example, the P1 symbol may be defined as predetermined, e.g. the OFDM symbol 2k, with a special structure using relatively rarely assigned carriers (e.g., every third position is allowed, as discussed above).

[0128] A potential problem with this approach is that under certain multitrack conditions the energy of the synchronization symbol can be greatly reduced, particularly on selected active carriers. For example, if there is a two-track channel with a 0 dB force and a delay that is 1/3 of the OFDM symbol length (useful part without a guard interval), then every third carrier will be largely canceled. With proper, unsuccessful phasing, these invalidations can occur in the locations of selected active carriers. The invalidation will be partial, but it can still be potentially harmful.

[0129] Another potential problem is the continuous interference of waves. It can be difficult to control with the above approach.

[0130] Other approaches to synchronization symbols have been used in the past, such as having a longer sequence of known waveforms (such as sinusoids) or pseudo random codes. The main drawback of such approaches is that they normally use quite long periods, and thus are relatively inefficient. Data capacity is reduced. What's more, a short matching / sync period is beneficial in mobile channels. Then, the channel may remain more or less stable during the synchronization symbol, which increases the detection performance of the synchronization.

[0131] In the embodiments, fairly short symbols (such as 2k or 1k symbols for DVB) are used, respectively defined in the frequency domain, including

- 29 rarely distributed carriers at pseudo-random locations, with relatively robust modulation (such as binary quadrature phase keying (BPSK) or quadrature phase keying (QPSK)). In addition, further short OFDM symbols may be used in the embodiments, both carrying a rare active carrier ("pilot"), the second symbol having active carrier locations shifted by a predetermined distance after frequency. For the sake of implementation, the travel may, in one embodiment, be one spacing between the carriers of the OFDM symbol. The exemplary embodiment applies that only the locations of carriers that originate from the regular structure (every second, every third or the like) are admitted. Active carriers were distributed (pseudo) randomly in these locations.

[0132] Fig. 27 shows an example of a symbol (P1) of a 2k symbol, according to an embodiment, with a length Tu of 224 μs and a guard interval of length Tg, which is a% of the length of Tu, a useful symbol. Ts, which is not shown in Fig. 27, refers to the length of the OFDM symbol = Tu + Tg. Locations of carriers are taken from every third raster, leading to a periodicity of 3 in the useful part of the signal.

[0133] The periodicity shown in Fig. 27 can be used to detect a signal efficiently and reliably. The receiver can accept a correlation between the first and second cycles to detect the presence of a signal. It should be noted that in normal data carrying OFDM symbols these correlations would be small - close to zero. The immunity of detection to interference can be increased by taking into account many simultaneous correlations, e.g. taking into account the correlation between the first cycle and the third cycle. You can also add a correlation between cycle 2 and cycle 3 for additional reliability.

[0134] The previous solutions are based rather on the guard interval correlation (GIC), because the data in the guard interval (cyclic prefix) is the same as the data at the end of the useful symbol (end of cycle 3 in the above example). However, according to embodiments, higher signal energy is available in correlation and it is possible to use several simultaneous correlations at different parts of the signal, which provides interference and noise tolerance. In addition, you can also make a correlation between the guard interval and any of the n sections.

[0135] In an additional aspect, some information transfer may be included in the synchronization symbol. One possibility is to use a known BPSK sequence (or QPSK) in OFDM symbols. The sequence can be one of a set of several sequences. These sequences may be the same or different for the first and second pulses. These sequences are correlated in the receiver with known sequences to determine which has been sent, and

- 30 the same few bits of information can be transferred. For example, if the number of possible BPSK sequence combinations is 8, then 3 bits can be transmitted. [0136] The modification is that the sequence in the first symbol can be used as a reference. Thus, the sequence in the first pilot symbol (P1a) is previously known to the receiver. The second pilot symbol may then have, say, m possible sequences. The sequence in the P1a symbol is used as the coarse channel estimation for the P1b symbol. The receiver decides which sequence was sent in P1b based on the phase information it received from the first symbol. Ld (m) (ie log2 (m)) bits are transferred again.

[0137] An additional modification is that information in the P1b sequence may be encoded by differential modulation, such that the encoded value in the P1b sequence is given as the phase difference as compared to the phase value at the corresponding carrier position in the sequence at P1a. Typically, this phase difference is 0 or 180 degrees in differential binary quadrature phase keying (DBPSK). You can also apply differential binary square keying (DBPSK).

[0138] According to embodiments, the receiver may use a periodicity that results from a rare sub-sampling. The receiver may perform frequency translations in several (i.e. two or more) subsections of received pulses, as discussed in more detail below.

[0139] According to embodiments, the transmitter may include synchronization signals in the transmitted signal. One preferred form of implementation is when the synchronization signal is determined based on a fixed FFT size (e.g., 1k in DVB-T2 / H2). The data carrying the OFDM symbols following the sync symbols (some other synchronization signals may also be certain) may also be another form having a different FFT size, symbol length, guard intervals (GIs), modulation etc. This preferred structure of the synchronization symbol will be described later. Second, the receiver can use the sync symbol properties in different ways. Some innovative receiver algorithms will also be described.

[0140] The resistance of P1 sync symbol to interference can be increased by dividing it into two parts, P1a and P1b, where both parts are FFT 1k symbols with relatively small guard intervals (GI) (such as 1/16) or even without any GI . Short guard intervals are sufficient because, in any case, the detection must be based on the resistance of P1 and we can not have long GIs to avoid inter-symbol interference (ISI, Intersymbol interference). However, strengthening the carriers in P1 counteracts this. Strengthening is due to the fact that only a relatively small number of active carriers are used. In one embodiment, an appropriate value would be that on average a carrier is used. In the case of 1k FFT, this would mean a row of 128 pseudo-randomly located active carriers.

- a relatively small number, such as 2, 3, 4, 5 or the like. The rest of the carriers can have a zero value.

[0141] Fig. 28 shows a sync symbol P1 having two consecutive OFDM symbols (P1a and P1b) that have the same FFT size according to an embodiment. Locations of active carriers are illustrated by means of wide continuous lines in Fig. 28. As shown in Fig. 28, the locations of active carriers in P1b are frequency-shifted relative to the location of active carriers in P1a. The narrow continuous horizontal lines in Fig. 28 show the acceptable locations of the supporting FFT symbols, and the dashed horizontal lines represent forbidden locations of carriers.

[0142] The location of the active carriers of the second part (P1b) would be shifted some distance after frequency, in one embodiment, by one location of the carrier compared to the first part P1a. This would have the advantage of not interfering with P1a to P1b, since the excess (due to multitrack propagation) from P1a would have entered unapplied carrier locations in P1b. Carrier locations 1-5 are indicated in Fig. 28. Carrier locations 1, 3, 5, ... are referred to as odd-numbered carrier locations, and carrier locations 2, 4, 6, ... are referred to as even carrier locations.

[0143] Several embodiments will now be provided using examples for DVB-T2 / H2. In the first example, due to the resistance to interference, the P1a structure could have: 1k FFT, length GI Tg = 1/16 * Tu (corresponds to 64 samples). The basic raster for 128 carriers is' / 2, an uneven pseudo-random pattern, the active carriers would have the average pseudorandom BPSK 1/6 coding (e.g., only even carrier carriers). P1b may be otherwise similar, but the locations shifted by one, i.e. only odd-numbered carrier locations are used. The BPSK sequence may be the same or different from that in P1a. Various uncorrelated BPSK sequences may be used to indicate e.g. the size of the FFT subsequently incoming data symbols or some other useful parameters.

[0144] In addition, in a further embodiment, the BPSK sequences in P1a and P1b can be selected such that the sequence in the first P1a symbol remains the same (and was known to the receiver). Additional information may be coded to select the BPSK sequence of the second P1b symbol. The active carriers in the first symbol may then serve as pilot values ​​for the second symbol. Since it is assumed that the length of the symbol is short in relation to the channel changes, it can be assumed that the channel remains more or less unchanged by the time of the second symbol. Moreover, since the frequency shift is one carrier gap (or some other small number), the channel phase can change by a relatively small amount from P1a to P1b (at corresponding locations).

In the second example of DVB-T2 / H2, since the synchronization symbol P1 is robust due to the gain of the active carrier (mean power may be the same as in actual data symbols), the guard intervals may be omitted. Thus

- 32 it would be possible to use the following structure: P1a: 1k FFT, 0 GI, 128 active carriers, pseudo-randomly located, using only even load locations, modulated according to BPSK using a random sequence. P1b: like P1a, but the locations of active carriers are shifted up or down by frequency by one carrier space after frequency, i.e. using only odd carrier locations, if P1a uses even carrier locations. FFT size information from data symbols can be coded for BPSK sequence selection. It can be, for example, 5 or 6 different sequences, which should differ from each other. The determination of these sequences can take place using the normal coding logic - maximizing Hamming distance between sequences.

In the third example of DVB-T2 / H2, the pulse structure is the same as in the second example, but short guard intervals (like 1/16) are used in both parts P1a and

P1b.

[0147] The receiver algorithms according to the embodiments will now be discussed. The pulse structure P1, as described above, is suitable for a variety of detection algorithms. At least the following information can be extracted from such a pulse structure: the presence of a DVB-T2 / H2 signal (or any other designated system). Because the signal has unique features in the time domain (periodicity n) and in the frequency domain (due to the known BPSK sequence), the pilot symbol can be reliably detected with a relatively low probability of false detection.

[0148] Coarse timing of the signal can be obtained. The P1 correlation properties provide a good candidate for proper time coordination. Numerous correlations and / or the use of most of the signal energy in correlation makes it reliable.

[0149] An estimate of the delay profile in a multi-channel channel can be obtained. The time correlation properties also provide a coarse multi-path channel structure. In particular for SFN networks, this can be used to estimate the difference in delay extremes in a multi-path profile (useful in locating the FFT window position for detecting useful data).

[0150] An estimation of the signal frequency shift can be obtained. This is based on the FFT in the first P1a symbol and / or the second P1b symbol. For example, by correlating received power on carriers with known active carrier locations, an offset can be solved.

[0151] A small amount of information encoded for selection of the BPSK sequence can be transferred (two methods as described above). For example, the size of the FFT useful data symbols in the frame can be signaled.

[0152] For detection, the correlation of half the symbol and preferably simultaneously P1a and P1b may be used.

[0153] The carrier shift may be solved taking FFT 1k into P1a and / or P1b (P1b may be used for additional noise immunity). Correlation with known BPSK candidate patterns would resolve the size of FFT useful data.

[0154] In the case of SFN, the delay value 1/2 of the length of the symbol 1k may create, for example, a partial elimination of the symbol P1a. However, in this case the P1b symbol would remain unchanged (constructive addition of a delayed component). In principle, the value of the symbol 1 delay could create blank values ​​for each 1k carrier (for continuous sine waves), but then it would be innocuous, since the delayed interference would go beyond the useful symbol period. Thus, as a result, the delay values ​​would not cause significant problems.

[0155] Embodiments of an exemplary receiver algorithm will now be discussed. In this particular case of the above second and third examples, it is worth noting that P1b is a transformed version of P1a. Thus, if the receiver would translate the P1b pulse by a known frequency distance, then P1a and P1b would be the same. This would mean that the correlations could be retrieved by P1a and the P1b version translated into frequency (even with the included guard interval). This may be beneficial for counteracting some interference, such as CW interference in one frequency. Frequency translation would mean that such interference would be virtually divided into two parts that differ in frequency. The correlation of such signals would then be close to zero,

[0156] A further embodiment of an exemplary possible receiver algorithm operates in cases such as Examples 2 and 3 above, where the BPSK (or QPSK) sequences in P1a and P1b are the same.

[0157] Fig. 29 shows an example P1 in which each of the symbols P1a and P1b has been divided into two parts according to an embodiment. Assuming that admitted carrier positions are taken from every second grid (e.g., even carrier numbers for P1a and odd carrier numbers for P1b), a useful part of the symbols includes two identical parts (i.e., P1a1 and P1a2 and P2a1 and P2a2 respectively). The receiver can now accept the mixing correlations of the parts P1a and P1b (see Fig. 30). For correlations, the following adaptations can be made in operations that extend over the time interval covered by the total length P1: samples belonging to the first period including the guard interval GI1 and P1a1 (in the third example it will mean 64 + 512 = 576 samples) will remain unchanged ; samples belonging to the second period (Pla2) will be frequency-translated, i.e. each sample is multiplied by exp (j2n1 / N), where N is the FFT size (1024 in this example), and i is the sample index (576 ... 1087) this will result in the desired frequency shift by one carrier space. Samples belonging to the first part of the second pulse including the guard interval GI2 and P1b1 are translated after

- 34 frequencies in the opposite direction by multiplying these samples by exp (j2n1 / N), where i goes from 0 to 575. Samples belonging to P1b2 remain unchanged.

[0158] A correlation is now formed by multiplying the modified samples corresponding to the parts P1a with the conjugate complex number of the modified samples corresponding to the parts P1b. The results from pairwise multiplications are summed up and a correlation result is obtained. The result will be maximized if the calculation period (in this example, 2174 samples) will be for the received P1.

[0159] Fig. 30 is a flowchart of a part of the receiver correlator according to an exemplary embodiment. The blocks designated GI1-P1b2 represent the received P1 symbol from the OFDM signal frame. The data is fed into the buffer memory and has a place of multiplication in pairs and summaries for each time of the incoming sample. The characters * in Fig. 30 represent a complex conjugate. The expressions T. \ p (jot) 'and TAp (-je) t)' mean frequency translation up and down respectively.

[0160] Fig. 31 is a flow diagram of a part of the receiver correlator according to an exemplary embodiment. This system leads to a narrow level of peak correlation. The correlation parts in pairs differ from those in the embodiment of Fig. 30. In contrast, the operating principle and operations are more or less the same. Protective compartments are not used in the arrangement of Fig. 31. Again, the terms 'H \ p (jo) t)' and 'H \ p (-je) t)' mean frequency translation up and down respectively.

[0161] In the arrangement of Figure 31, the guard compartments are not used in correlation, which makes the buffer memory a bit shorter. Importantly, the parts of the symbol that are correlated in pairs are on the one hand P1a1 vs. P1b2, on the other hand P1a2 vs. P1b1. It can be shown that this leads to a narrower peak correlation level than the system of Fig. 30. The disadvantage of the embodiment of Fig. 31 is that energy from the guard compartments is not used. Thus, it would be advantageous to make GIs much shorter than the length of the symbol or even to use zero GI.

[0162] The embodiments of Figs. 30 and 31 provide the following results: the modifications (translations) in P1a and P1b are symmetrical (frequency shifts up and down), which may improve the cancellation of small identical errors in processing; modifications take place at (about) half of the symbol's length interval, which helps to "tear out" a possible continuous wave (CW), multiple path and other interferences in correlation; all the pulse energy is applied - also the guard interval in the embodiment of fig. 30; and the correlation peak level will be quite narrow (about half the width, if P1a and P1b would be correlated like frequency translation takes place only in P1b (or P1a)).

[0163] Fig. 32 shows the steps of a detection sequence according to an embodiment. Fig. 32 is shown in the context of signal processing according to the third example described above.

[0164] The correlation is taken from a period that corresponds to the length of the P1a symbol, as shown at 3202. Samples that are taken at the delay Ts (= length P1a) are corrected by multiplying them by a complex coefficient e \ p (-jo) t). where ω is the frequency difference between the carriers of the OFDM symbol (= 1 / Tu).

[0165] The above correlation result is compared to the rolling average of the latest correlation results as shown at 3204.

[0166] When the above comparison result exceeds the set threshold value, a decision is made that P1 is present and following the "yes" branch from 3206. Otherwise, the processing for the correlation period is terminated as shown at 3216.

[0167] The receiver determines the local maximum correlation position and determines the origin of the P1 symbol based on the local maximum position as shown at 3208.

[0168] The receiver takes 1 k FFT through the signal portion P1a, as shown at 3210. [0169] The receiver correlates different frequency domain BPSK sequence positions as shown at 3212. The maximum correlation ensures the frequency grid position in the FFT domain that can be used, to resolve the difference between the receiver's carrier frequency and the nominal frequency (carrier displacement). The sequence that maximizes the correlation indicates the size of the FFT of the actual data symbols.

[0170] The possible frequency offset is corrected for the detection of the following symbols, as shown at 3214, and the processing for the correlation period is terminated as shown at 3214.

[0171] According to the embodiments, the tolerance for the multiplier weakening is increased, because it would be highly unlikely that both portions P1a and P1b would be canceled at the same time. When the receiver detection is based on the correlation results using both P1a and P1b, at least one result will be much higher than just correlation with noise or random data.

[0172] In addition, due to the periodicity of P1 due to the rare use of bearing positions, the receiver can base its decision regarding the presence of signal P1 on correlations in periodic signal parts. This periodicity increases the applied signal strength (since virtually all of the signal energy can be used in detection) as well as increases diversity, since different parts of the signal can be used in the time domain, providing a variety of correlation results. These correlation results differ in the content of interference (or unwanted signals) and can be used to make more reliable decisions.

[0173] One or more embodiments of the invention may be implemented in computer-executable instructions, such as in one or more program modules, executed by one or more computers or other computers.

- 36 devices. In general, the program includes procedures, programs, objects, components, data structures etc. that perform individual tasks or implement individual abstract data types when executed by a processor in a computer or other device. Computer-executable instructions may be stored on a computer-readable medium such as a hard disk, optical disk, portable data carrier, permanent memory, RAM etc. As will be appreciated by those skilled in the art, the functionality of the program modules may be combined or arranged as desired in various embodiments. In addition, the functionality can be implemented in whole or in part in software or hardware equivalents, such as integrated circuits, directly programmable gate arrays (FPGAs),

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