Three stage power source for electric arc welding with the first stage having a DC-DC converter with a soft switching circuit

The invention relates to the field of electric arc welding and more specifically to the three-stage power source according to the preamble of claim 1 (see for example, WO 03/015973 A), for such welding and the new relationship between the first two stages of the three-stage power source.

BACKGROUND OF THE INVENTION [0002] Electric arc welding requires passing AC or DC current between a metal electrode and a workpiece in which the metal electrode is usually a hollow metal wire or a solid metal wire. The power source is used to create a given pattern and / or polarize the current between the electrode advancing wire and the workpiece so that the arc melts the tip of the advancing welding wire and deposits molten metal on the workpiece. Although various inverter technologies are used as the power source, the most effective is the inverter-based power source, in which the switching network includes high-frequency switches to produce the desired waveform or current level for the welding process. The inverter power source is described in US 5,278,390 (Blankenship) in which the inverter is controlled by "waveform control technology" paved by The Lincoln Electric Company of Cleveland, Ohio. The current waveform is generated by a series of short pulses formed at a frequency generally above 18 kHz, and the group of short pulses has a profile controlled by a waveform generator. According to the standard power source technology, the input signal to the power source inverter stage is current rectified from a sinusoidal power source. It is common practice to use a suitable power factor correction converter that is either part of the inverter switching network as shown in US 5,991,169 (Kooken), or located in front of the inverter stage as shown in US 6,177,645 (Church). Indeed, a power source with inverter or power factor corrected stage has been known in the field of welding for years. Other power sources using an input converter with power factor correction in the form of a boost converter are shown in the document

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US 6,504,132 (Church). Two Church patents and a Kooken patent form the background of the invention. In both documents US 5,991,169 (Kooken) and US 6,504,132 (Church) the actual welding current is regulated by the output contact chopper or lowering converter (buck converter) and the isolation is obtained by the transformer either at the output of the inverter stage or the output of the boost converter . These different topologies for power sources are well known in arc welding technology. In these prior art patents, the actual welding current, welding voltage or power is controlled at or before the output stage of a power source whose output stage is either an inverter or a contact interrupter. Both the inverter and the contact interrupter are unregulated to produce a constant low voltage DC bus to supply regulated welding.

[0003] The isolation of welding operations is characteristic of most welding power sources. The term "welding" includes "plasma cutting". In US 5,991,180 (Vogel), a preliminary control system using a boost converter is directed to the converter, which is disclosed as a contact chopper having an output isolation transformer located after the welding control and directly controlling the welding operation. In this power source, the contact chopper network is controlled to produce the desired controlled output welding current, and isolation is provided in the output stage. Similarly, US 5,601,741 (Thommes) shows a boost converter to power a controllable inverter with pulse width modulation providing an adjustable output signal for actual welding operations. In both the Vogel and Thommes documents, the second stage is regulated so that it directs the controlled power factor current from the pre-regulation system to the welding operation. Welding control occurs in the second stage and is usually directed by the pulse width modulation control circuit. Both Vogel and Thommes are state of the art technologies. In US 6,278,080 (Moriguchi), the power source of the inverter type is regulated to control the desired welding current. Isolation is obtained by a transformer between the controlled second stage inverter and the welding output, which is disclosed as a DC welding operation. A similar power source is shown in US 5,926,381 (Moriguchi) and US 6,069,811 (Moriguchi), in which current isolation

From the inverter stage occurs at the output of the inverter and directly manages the welding operation. Document US 5,926,381 (Moriguchi) discloses a common system for using the voltage at the first stage output of a booster converter to provide a control voltage for either the adjustable inverter stage or the booster converter itself. Three Moriguchi patents are prior art information showing a known power source in which an adjustable inverter is routed through an input boost converter or rectifier DC output to produce a controlled welding current directed to the isolation output transformer. The secondary AC signal of the isolation transformer is directly used for welding operations. There is no third degree topology like the one used in the new power source of the invention.

[0004] Turning now to non-welding technology, an aspect of the invention is the use of a synchronous rectifier device at the output of a second stage DC / DC converter. Synchronous rectifiers are commonly used and one such rectifier is disclosed in US 6,618,274 (Boylan). Document US 3,737,755 (Calkin), discloses a DC / DC converter in low power applications in which a steady regulated current is directed to an unregulated inverter to provide a constant DC output signal. Any control of the unregulated inverter is on the input side of the inverter so that the DC input signal is the only parameter that can be adjusted to control the fixed DC output signal of the inverter. This is a topography that requires control of the signal in the inverter so that the inverter provides a controlled steady output signal. Well-known technology not related to welding from Boylan and Calkin presented here to show a synchronous rectifier and a version of an unregulated inverter in which any adjustment is carried out in front of the inverter by controlling the input DC signal level. None of these patents relate to the power source for welding and are only presented by reference as general technical concepts such as synchronous rectifying devices and unregulated inverters. A two-stage AC-DC converter not intended for welding is shown in document US 5,019,952 (Smolenski) which introduces minimal harmonic distortion to the current flowing into the converter. The load is unchanging and does not require adjustment as it is

Required in the welding operation. This patent shows general technology not related in any way to the power source requirements for electric arc welding.

[0005] These patents form the background of the invention relating to a power source that must be regulated by a welding operation in which such adjustment takes place in a medium current, medium voltage and current welding operation feedback loop. Steady load power sources are not suitable for the present invention, only as general technical information.

[0006] In the past, an inverter in a power source produced a welding current regulated by a parameter in the welding operation, such as current, voltage or power. This inverter was normally controlled by a pulse width modulator in which the duty cycle of high frequency switches was controlled by feedback from the welding operation so that the duty cycle was regulated in a range substantially less than 100%. This type of PWM controlled inverter is referred to as a regulated single stage inverter. Such an inverter was the output of the power source and was the last stage of the power source. Lower fill rates resulted in higher primary currents and higher losses. The inverter performance varied according to the fill factor control caused by the requirement to adjust the single-stage output of the inverter to create an output signal suitable for welding. Using the power source in which the last stage is a regulated single-stage inverter resulted in heat loss, lower efficiency, high cost and increased size of elements. For this reason, certain welding source manufacturers have advertised power sources as better than inverter based power sources because they do not use inverters with the resulting high cost and other difficulties. The inverter stage was to be avoided, as it had the dual function of isolating the output and regulating the current to produce a welding current. See Hoverson 6,723,957 document presented here as a reference.

THREE-STAGE POWER SOURCE ENHANCED BY THIS

INVENTION

[0007] The present invention is used with a three-stage power source for electric arc welding (plasma cutting), in which the power source inverter is a second stage as it has been until now, but is unregulated such that a third stage can be added to provide real adjustment to create a welding current. Using this new three-stage concept, the inverter can operate at a very high switching frequency, while the output third stage can be a contact breaker operating at a lower switching frequency. Consequently, the switching frequency is optimized by a function performed by the stage as opposed to the need to use a high frequency in the inverter stage with a pulse-width modulation used to actually adjust the welding output current. In addition, the isolated, fixed DC voltage for the regulated third stage can be substantially lower than the DC voltage from the input stage of the transducer and much higher than the current output welding voltage.

[0008] A three-stage power source using the invention requires a new topography for a power source in which the inverter with modulated pulse width is simply a second stage for creating an isolated fixed DC output bus without feedback signal for an inverter with second stage modulated pulse width. This insulated rail is used in a third stage controlled by actual welding parameters to produce a welding current. Consequently, the invention uses an unregulated second stage not only providing the necessary insulation but also producing a constant DC output rail to be used by the third stage in which welding control is obtained. The unregulated second stage inverter operates at very high frequency with a duty cycle which is determined during the operation of the power source. The frequency is over 18 kHz and preferably around 100 kHz. The fill factor is set at different levels; however, the preferred duty cycle is close to 100% for maximum performance. The use of a fixed, high duty cycle minimizes the current flow time of the second stage inverter controlled by the phase shift modulator to substantially reduce heat and increase efficiency. The output of the second unregulated inverter stage can be rectified using well

Known synchronous rectifying devices, which devices are controlled by the secondary winding of the internal isolation transformer of the unregulated second stage inverter. By using synchronous rectifying devices on the second stage output, an additional improvement in the overall efficiency of the power source is obtained. The first stage is either an input rectifier or an input rectifier with a power factor correction converter. A converter with a first stage power factor correction is preferred. This converter is located behind the standard rectifier or can be connected to the rectifier. Of course, this converter can be a passive converter with power factor correction or an active converter such as a boosting, lowering or lowering increasing converter. The first stage of the invention produces a first DC bus at a fixed voltage. Using the standard first stage for a power source, the first DC output signal, which is the DC input rail for the unregulated inverter, can be adjusted and set to approximately 400-900 volts DC. The output of the unregulated, isolating inverter forming the second stage of the new power source is a fixed DC bus having a predetermined relationship with the first DC input bus. The voltage of the second DC bus or output is generally less than the voltage of the DC bus from the first stage. The power source therefore produces a second DC bus that has a mathematical relationship with the DC input bus from the power factor corrected converter. According to standard practice, the unregulated second stage inverter includes an isolation transformer having a primary winding and a secondary winding so that the secondary winding is isolated from the input of the power source. See US 4,864,479 (Steiger). The unregulated second stage inverter can operate at a switching frequency to optimize the operation of the second stage inverter. In this way, an extremely high switching frequency is used to reduce the size and cost of the components in the new unregulated second stage inverter. By using a fixed duty cycle with phase shift control, voltage and current surges in switching devices are reduced to ensure soft switching operation. In fact, in a preferred embodiment, the duty cycle is set to a value close to 100% so that the switches are fully on or fully off. This drastically reduces the current circulating in the other

And substantially improves the performance characteristics of the second stage inverter, which also provides the function of isolating the power source welding output from the AC source of the power source. Due to the fact that the switching devices in the second stage of the unregulated inverter operate fully switched on, this inverter has high efficiency and is very flexible in operation. The isolation transformer determines the relationship between the fixed DC bus on the input side of the unregulated second stage ("first DC output signal" from the first stage) and the DC output bus on the output of this second stage ("second DC output signal"). In some prior art power sources, the duty cycle on the primary winding of the isolation transformer in the adjustable inverter is regulated by the welding operation. In the new three-stage power source according to the present invention, there is no adjustment of the welding operation either in the first stage or the second stage.

[0009] A power source for electric arc welding having the property of active power factor correction and the tight control of the energy output directed to the welding operation requires at least two switching stages. These two stages ensure that the instantaneous energy sent to the power source and transferred from the power source can be adjusted independently with the appropriate energy storage elements. Thus, the power factor corrected power source for electric arc welding generally requires two independent switching control circuits. One of the control circuits is used to control energy or output current for welding operations. The second control circuit is used to control the DC signal from the active converter with power factor correction forming the first stage of the power source. Thus, power sources for electric arc welding with power factor correction capabilities require two switching networks, each with independent control requirements. The first switching control is suitable for the welding current output and the second switching control is suitable for power factor correction in the input power source. The second switching control ensures that the first stage output is a fixed DC voltage referred to as "DC bus". The DC bus voltage is itself used to control the first stage converter to ensure that the DC bus from this converter has a fixed voltage level. To sum up the power source based on the inverter for welding

An electric arc requires two separate switching networks and two control circuits for these networks.

[0010] A power source based on an inverter for electric arc welding has a different conceptual requirement. One of the stages in the power source must provide electrical insulation between the AC variable input signal and the adjustable output current suitable for welding. The isolation device is usually in the form of a transformer. In the prior art, two-stage power sources based on the inverter have two locations per isolation device. In the first example, the input factor with power factor correction is not insulated and the isolation transformer is provided in the second stage of the adjustable output inverter. In another example, the insulation occurs in the first stage of the power factor correction converter. In this second example, an uninsulated output inverter or other uninsulated inverter can be used as the second stage. The first example is more efficient than the second example because of the effect of 60 Hz on the RMS current on the input side of the power source. To sum up, insulation is the second conceptual requirement for a power source for welding.

[0011] Two requirements for an active power source with power factor correction for welding are (a) two separate and independent control circuits for two separate switching networks, and (b) a suitable structure for isolating the power source input from the power source output. These basic requirements for inverter-based power sources are implemented in a secondary three-stage power source. The unregulated second stage is an insulating stage between two adjustable uninsulated stages creating a unique system requiring a three-stage power source based on the inverter. The new three-stage power source is more efficient than the two-stage power source based on the inverter assuming that the same power factor correction pre-regulation system is used. Thus, the new three-stage power source is more efficient, but still has the main characteristics required by the power source used in electric arc welding. There are two independently controlled switching networks. There is an insulation step. These limits are provided in such a way as to increase efficiency and achieve better welding efficiency and better heat dissipation of the power switching elements.

[0012] Because the second unregulated stage of the three-stage power source inverter provides system isolation, many types of uninsulated transducers can be used as a power factor correction pre-regulation system. The booster converter is the most popular converter due to the current shaping function and continuous current characteristics on this type of processing line. However, the boost converter output voltage is higher than the peak of the highest line voltage whose peak can be 775 Volts. Thus, other active power factor correction controllers can be used in the invention, which is a three-stage power source in which the second stage is unregulated and provides insulation. One of the other options for an active input with power factor correction or first stage is an up / down converter so that the primary voltage bus or input bus to the second stage can be lower than the peak of the AC input voltage signal for the power source. This type of power factor correction converter still produces little harmonics. One such power factor converter is referred to as a pulse boost / drop converter. DC bus voltage from 400 Volts to 500 Volts used for the second stage is obtained with an AC input voltage in the range from 115 Volts to 575 Volts. Regardless of the AC voltage for the first stage, the output voltage of the active power factor converter is controlled to be at least between 400 Volts and 500 Volts. Other types of inverters with active or passive power factor correction can be used in the invention. A preferred transducer is active thus constituting a second switching network requiring a second control circuit. The use of the term electric arc welding also includes other output processes such as plasma cutting.

[0013] As explained so far, a three-stage power source using the invention uses a three-stage power source for electric arc welding. The third stage feedback control creates an output current suitable for welding. The input first stage is usually an active power factor correction converter requiring a second switching network and a second independent control circuit. This three-stage topography is not used in the prior art. With this topography, the second stage added is only used to transform the DC bus

High voltage on the primary side of the second stage on the DC bus of the lower voltage on the secondary side of the second stage insulated from the primary side. Thus, three stages require a DC bus on the secondary side of the second stage so that the rail can be used to adjust the welding power. The term "bus" means a DC signal that has a controlled level. The three-stage power source has a first DC bus from the input stage called the "first DC output" where the first DC output has a controlled DC voltage. There is a second DC bus on the secondary side of the second stage called the "second DC output" where the second DC output is also a controlled DC voltage level. The creation of a second DC bus on the secondary side of the unregulated inverter has advantages other than the benefits of using an unregulated second stage inverter as described so far. The secondary DC bus or secondary DC output is insulated from the primary side of the second stage so that there is no need for isolation in the third stage of the welding control circuit. In other words, an output control circuit, such as a contact breaker, has an input DC bus with a fixed voltage level. In practice, the contact breaker has a controller with control voltage, which is determined from the DC input to the contact breaker. This DC input signal is isolated from the input power supply. Consequently, the control voltage for the output stage controller or contact breaker can be determined from an uninsulated DC source. This is usually the input signal for the contact breaker. Separate control voltage isolation for the controller used in the output stage is not required. The use of a fixed DC bus from the second stage allows the DC voltage at the third stage output, which is regulated by welding operations, to be much lower than the normal primary DC input bus ("DC first output") of the power source. In the past, the output of the power factor converter was a relatively high DC signal based on the use of a boost converter. This high DC voltage was directed to an adjustable inverter stage for use in generating a welding current. Using the present invention, the high voltage from the power factor converter output rail is drastically reduced. It is more efficient to convert a 100-volt DC bus to a 15-volt controlled power supply than to convert a 400-volt DC bus to a 15-volt controlled power supply.

[0014] The second stage of the secondary three-stage power source is in the form of an unregulated DC-DC converter, has an input connected to the first DC output signal and an output in the form of a second DC output signal electrically isolated from the first DC output signal the value of a given ratio to the first DC output signal. The power source contains a third stage for converting the second DC output signal into welding current for the welding process. The third stage of the power source includes an adjustable inverter such as a contact breaker or inverter. When an inverter is used, the output is a DC signal directed to a polarizing network or switch, which switch allows DC welding through a power source. The polarization switch allows welding with either negative DC, positive DC or AC. The welding process using either a contact chopper or an inverter can be performed in shielding gas, such as in MIG welding, and can use any type of electrode, such as a tungsten, hollow wire or solid metal wire. According to an aspect of the invention, the output of the unregulated DC-DC converter is substantially smaller than the input of the second stage. In most cases, the input and output of the second stage are DC voltages with substantially fixed values.

INVENTION [0015] The invention provides a power source according to claim 1.

[0016] There are several advantages of operating inverters for welding at high switching speeds. For example, smaller magnetic components translate into better mobility. Another advantage is the potential of having a control system with a larger bandwidth, which will provide better arc performance. Thus, the new three-stage power source described above is improved by the present invention. The secondary three-stage power source has power switches operating at extremely high switching speeds exceeding 18 kHz. The boosting power switch for the first stage and the four power switches for the unregulated second stage all work at high frequency to obtain the benefit of high switching speeds. There is also a disadvantage of using such high switching speeds. Such switching speeds cause losses

Switching. If the switching losses are not reduced, power source efficiency and reliability are reduced. Switching losses are caused by the overlap of current and voltage during switching, either from on to off or from off to on. To reduce switching losses, either voltage or current must be kept close to zero during switching. The switching transition can be either zero voltage or zero current or both. This is called "soft switching". To achieve soft switching through zero voltage or zero current at high switching speeds, so-called resonance techniques or resonance acids were used. However, this type of known soft switching control often causes higher current and voltage loads due to sinusoidal waveforms and still has conduction losses. However, there are known soft switching circuits that use zero voltage transition converters or zero current transition converters in such a way as to reduce both switching losses and conduction losses.

[0017] It is known that the unregulated second stage inverter of the new three-stage power source to which the present invention is directed utilizes PWM phase shift to control output power. By setting the phase shift at a high level close to 100%, preferably above 80%, the switching losses in the second unregulated stage are limited. Using a PWM control with a fixed phase shift, the second stage works near full conduction to create low conduction losses. The second unregulated stage is actually softly switched. According to the invention, the three-stage power source described above has soft switching in the input stage. To date, the present invention requires the use of an active soft switching circuit so that the first input stage is connected to the actual soft switching of the second unregulated stage. This combination of added soft switching with proper soft switching has essentially the increased efficiency of the new three-stage power source to which the invention is directed.

[0018] The first stage active soft switching circuit is a typical circuit described in the 1991 article issued by IEEE entitled High Efficiency Telecom Rectifier using A Novel Soft-Switching Boost-based Input Current

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Shaper. This type of circuit is also described in the 2002 article by IEEE entitled A New ZVT-PWM DC-DC Converter. Another active soft-switching circuit is the voltage-current transition circuit described in article 2004 entitled A New ZVT-ZCT-PWM DC-DC Converter published by IEEE Transactions on Power Electronics published in May 2004. These articles describe an active soft switching circuit or circuits of the type used in the first stage of the three-stage power source. The invention combines active soft switching for the first input stage and an unregulated inverter of proper soft switching using PWM phase shift control. Document US 4,864,479 (Steigerweld) discloses a common unregulated inverter using phase shift control. This type of unregulated power stage has a topography that increases efficiency by minimizing circulating currents by using switching operations with a fixed high duty cycle. An unregulated inverter operating with a fixed duty cycle will achieve soft switching on all primary switches with a minimum of conduction losses. This concept is used in the second stage of the three-stage power source to which the invention is directed.

[0019] According to the invention, the high speed switching power switch from the first stage three-stage power source is softly switched by means of an active circuit to reduce both switch loss and output rectifier losses. In addition, the output stage of the soft switches is connected to a second stage having the capacity of proper soft switching using an unregulated inverter with a fixed duty cycle, phase shift. The combination of an active soft-switching circuit for the first stage combined with proper soft-switching of an unregulated inverter with a fixed duty cycle substantially increases the efficiency of the new three-stage power source to which the present invention is directed.

[0020] Utilizing the active soft switching circuit in the first input stage of the three-stage power source, the first stage pulse width modulator converter has zero voltage switching for the active converter switch and zero blocking recovery current for

Output diode. This soft switching occurs without increasing voltage or current surges, i.e. conduction losses of two components. This soft-switching circuit for the power switch (active) for the first stage includes a zero voltage transition using a network with inductance branch and capacitor branch both in parallel with both an active power boost switch with pulse width modulation and a passive output switch or output boost diode. The two branch network includes an inductive branch capacitive branch controlled by switching an auxiliary switch. The auxiliary switch is also connected in parallel with the boost power switch with pulse width modulation and is turned on for a short period of time just before turning on the switch with pulse width modulation. The network current of the coil increases until the output rectifier diode turns off, communicating it with the soft switching operation. The coil current continues to increase by bringing the voltage across the pulse width modulation circuit to zero before turning on the boost switch. The reverse diode of the switch with pulse width modulation is therefore polarized in the forward direction. An on signal is applied to the power switch while a feedback diode conducts to ensure zero switching of the modulation switch on power up. The auxiliary switch is then turned off and the modulation power switch is turned on. The auxiliary diode and capacitor provide a voltage suppressor on the auxiliary switch so that the auxiliary switch is not overloaded during shutdown. The current of the induction branch quickly drops to zero, during which time the auxiliary switch is turned off. The other operation is the same as the operation of a traditional boost converter with pulse width modulation, except that the energy stored in the two branch network is transferred to the load when the main switch is turned off. In some descriptions of these two branches, they are referred to as a resonant circuit, which may be technically true, but not necessarily for the soft switching function.

[0021] A circuit with two branches controlled by the auxiliary switch is used in the first stage of the present invention to provide soft switching of both the power switch and the output diode. Such a circuit is described in US 5,418,704 (Hua). Soft switching

First stage and natural soft second stage switching is the result of using the present invention.

[0022] According to the present invention, a three-phase power source is provided for the electric arc welding process. This power source contains an input stage having an AC input and a first DC output signal, a second stage in the form of an unregulated DC-DC converter having an input connected to the first DC output signal, a network of switches switched at high frequency with a given duty cycle to convert the input signal to the first internal AC signal, and an isolation transformer with a primary winding supplied by the first internal high frequency AC signal and a secondary winding to create a second internal high frequency AC signal and a rectifier to convert the second internal AC signal into a second stage DC output signal. The magnitude of the output signal for the second stage is related to the amount of overlap between the phase shifts with the switches that use the phase shift controlled by the pulse width modulator so that the second stage is properly softly switched. The third stage in the power source is used to convert the second DC output signal from the second stage to the welding output for the welding process. This three-stage power source is improved by providing a DC-DC converter in the first stage, where the converter has a power switch with a soft switching circuit. Thus, the first stage soft switching circuit complements the proper soft phase shift switching, the unregulated second stage to increase the efficiency of the first two stages in the three stage power source.

[0023] According to another aspect of the present invention, the soft switching circuit of the first stage three-stage power source is an active damping circuit with an auxiliary switch operating in accordance with the power switch to positively divert voltage towards zero during both switching transitions. The first-stage DC-DC converter has an output or boost diode which is also softly switched by the first-stage soft switching circuit. In accordance with another aspect of the invention, the first stage DC-DC converter has a positive and negative output terminal with a capacitor connecting the terminals

55 and the diode connecting the positive end of the auxiliary switch to the positive output terminal. A three-stage power source with a unique combination of active soft switching on the first stage and proper soft switching on the second stage is used in the third stage contact breaker. Optionally, the output contact breaker has a soft switching circuit for its power switch. All these features of the present invention improve the three-stage power source having, as a new feature, an unregulated middle insulation step increasing the efficiency of the power source while maintaining the benefit of its three-stage topography.

[0024] The present invention is a combination of an input stage and an unregulated center stage of a three-stage power source, wherein the first stage has an active soft switching circuit for a boost power switch and the proper soft switching for a phase shifted unregulated second stage. Consequently, the invention requires a two-stage AC-DC converter comprising an input stage having an AC input and a first DC output signal and a second stage. The second stage is in the form of an unregulated DC-DC converter having an input connected to the first DC output signal, a network of switches switched at a high frequency with a given duty cycle to transform the input into the first internal AC signal, Isolation transformer with primary winding powered by the first internal high frequency AC signal and secondary winding to create the second internal AC signal, and a rectifier to convert the second internal AC signal to the second DC output signal of the second stage. The magnitude of the output signal for the second stage is related to the amount of overlap between phase shifted switches. The input stage includes a power switch having a soft switching network, which network is an active damping circuit with an auxiliary switch operating in accordance with the first stage power switch.

[0025] The first object of the present invention is to provide a new three-stage power source in which the first stage has an active soft switching circuit for a fast switched network switch and the second stage is an unregulated inverter forming part of the insulation stage,

Which inverter has soft switching characteristics based on a fixed high duty cycle for several of its switches.

[0026] Another object of the present invention is to provide a two-stage inverter for use in power processing, which converter includes a power switch with an active soft-switching circuit and the second stage comprises an unregulated inverter with a fixed duty cycle controlled by phase shift.

[0027] Still another object of the present invention is to provide a three-stage power source as defined above, which three-stage power source also has an output stage in the form of a contact breaker with a contact breaker power switch having a passive soft switching circuit.

[0028] Still another object of the present invention is to provide a three-stage power source, as defined above, which power source comprises an active soft switching circuit for the first stage, a proper soft switching characteristic for the second stage, and a passive soft switching circuit for the third stage.

[0029] These and other purposes and advantages will be clear from the following description taken with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS (THREE-STAGE POWER SOURCE) [0030]

FIGURE 1 is a block diagram illustrating a three stage power source and showing an embodiment of the three stage power source improved by the invention;

FIGURE 2 and FIGURE 3 are block diagrams similar to FIGURE 1 showing further examples of the implementation of a three-stage power source;

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FIGURES 4-8 are partial block diagrams showing a three stage power source with various first stage embodiments;

FIGURE 9 is a block diagram showing the last two stages of a three stage power source in which the output stage provides AC welding current;

FIGURE 9A is a block diagram of a waveform technology control circuit for use in the three stage power source shown in FIGURE 9, together with graphs showing three welding waveforms;

FIGURE 10 is a block diagram illustrating the second and third stages of a three stage power source in which the output stage is DC welding current;

FIGURE 11 is a block diagram illustrating the topography of a three-stage power source for creating a current suitable for electric arc welding with two separate controller control voltage sources;

FIGURE 12 is a block diagram illustrating a specific three-stage power source using the topography to which the present invention is directed;

FIGURES 13-16 are wiring diagrams showing four different circuits for power factor correction in a first stage three-stage power source;

FIGURE 17 is a combined block diagram and wiring diagram illustrating the preferred embodiment of the unregulated inverter constituting the new second stage of the three stage power source to which the present invention is directed;

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FIGURES 18-21 are wiring diagrams showing several inverters used as an unregulated, isolation second stage inverter having a new aspect of the three stage power source to which the present invention is directed;

FIGURE 22 is a wiring diagram of a first input stage and a second insulating stage forming an embodiment of the present invention;

FIGURE 23 is a wiring diagram of a second embodiment of the present invention;

FIGURE 24 is a wiring diagram illustrating a three-stage power source in which the output stage is a contact breaker with a passive soft switching circuit;

FIGURE 25 is a wiring diagram showing the active soft handoff circuit used in the embodiment of FIGURE 22;

FIGURE 26 is a wiring diagram showing the active soft switching circuit used in the preferred embodiment of the invention; and

FIGURE 27 is a graph of the voltage curves and trip signals for the main power switch and the auxiliary circuit switch shown in FIGURE 26.

THREE-STAGE POWER SOURCE (Figures 1-21) [0031] The present invention is a modification of a new three-stage power source for use in electric arc welding as developed by The Lincoln Electric Company, which power source is not prior art for

Of the present invention. The new three-stage power source has an input stage for converting the AC signal to the first DC output bus. This output bus has a constant voltage level and is directed to the second stage input best shown in FIGURE 16. This new second stage of the three-stage power source is an unregulated inverter that has the insulation feature and has a second DC output or a second DC bus that is proportional to the DC input bus. The level relationship is determined by the design of the unregulated inverter. The unregulated second stage inverter has a switching network in which the switches operate at a high switching frequency greater than 18 kHz and preferably around 100 kHz. The frequency of switching networks of switches in the unregulated inverter forming the second stage of the power source allows the use of small magnetic components. The isolated DC output of the unregulated inverter is directed to the third stage of the power source. This third stage can be either a contact chopper or an inverter that is regulated by a welding parameter such as current, voltage or welding power. In modification, this third stage is preferably a contact breaker. The topography of the three-stage power source has an input stage to produce the first DC signal, the second unregulated DC-DC stage to provide an isolated fixed DC voltage or DC bus, which is used by the third stage power source to regulate the current used in welding operations. Three examples of the three-stage power source to which the present invention is directed are illustrated in FIGURES 1-3. The power source PS1 in FIGURE 1 includes first stage I, second stage II and third stage III. In this embodiment, stage I includes an AC-DC converter 10 for converting the AC 12 input signal to the first DC bus 14. Input 12 is a single-phase or three-phase AC mains voltage with a voltage that can vary in the range of 200-700 Volts. The converter 10 is depicted as an unregulated device that can be in the form of a rectifier and a mains filter to produce the DC bus 14 (DC # 1). Since the AC input signal is between-wire voltage, the DC 14 bus has a substantially uniform size. The unregulated inverter A is a DC-DC converter with an isolation transformer for converting the DC 14 (DC # 1) bus to a second DC bus or a second DC 20 output (DC # 2). Output 20 creates the power input for stage III, which is inverter 30. DC voltage on line 20 at

Current suitable for welding on line B. Feedback control or control loop C detects the parameter in the welding operation and regulates the current, voltage and power on line B by adjusting the inverter 30. In practice, the inverter 30 is contact breaker, however, an inverter can be used as an alternative. Having the three-stage power supply PS1 as shown in FIGURE 1, the second stage switch network has a frequency that is normally higher than the frequency of the inverter 30 switching. In addition, DC voltage on line 20 (DC # 2) is generally less than DC voltage from stage I on line 14 (DC # 1). In practice, there is an isolation transformer in the inverter A. The transformer has an input or primary section or side with a substantially larger number of turns than the secondary section or side used to generate voltage on line 20. This ratio of the number of turns in practice is 4: 1 so that the voltage on line 20 is 1/4 of the voltage on the line 14. For DC # 1, this voltage is in practice around 400 Volts. [0032] The general topography of the three stage power source to which the present invention is directed is shown in FIGURE 1; however, FIGURE 2 shows a preferred implementation in which the PS2 power source has substantially the same stage II and stage III as the PS1 power source; however, input stage I is an AC-DC converter 40 containing a rectifier followed by an adjustable DC-DC converter as defined in the present invention. The converted signal is the DC signal on line 14 shown as the first DC bus (DC # 1). Voltage on line 14 is regulated as indicated on return line 42 according to standard technology. Thus, in the PS2 power source, the output welding converter 30 is controlled by feedback loop C. The voltage on line 14 is regulated by the feedback loop shown as line 42. Since the inverter 40 is a power factor correction converter, it detects the voltage waveform represented by line 44. Using the PS2 power source, the first DC bus 14 is a fixed DC voltage with a different voltage. one phase or three phases on input 12. Thus, output 20 is merely a transformation of the DC voltage on line 14. DC # 2 is the fixed voltage with the level determined by the isolation transformer and the constant duty cycle of the switching network in the unregulated inverter A. It is a favorable implementation of the new power source using three separate and different stages with stage II which is the unregulated inverter to transform the fixed first DC output or bus DC for a second fixed

A DC output or DC bus used to power an adjustable welding inverter, such as a contact chopper or inverter. As another alternative, a degree

And it can be adjusted by feedback from DC bus # 2 on line 20. This is shown by dotted line 46 in FIGURE 2.

[0033] The PS3 power source in FIGURE 3 is another implementation of the three stage power source. This is not a favorable implementation; however, the three stage power source of the present invention may have an input converter 50 controlled by feedback loop 52 from welding output B. With this three-stage power source utilization, the inverter 50 is controlled by the welding output and not by the voltage on line 14 as is the power source PS2. Adjustable from welding output B, the inverter 50 is both a power factor correction stage and a welding regulator. However, such an implementation of the three stage power source is presented for full technical disclosure.

[0034] As previously described, input stage I converts the signal AC 12 of either a single phase or triple phase to a fixed DC 14 (DC # 1) bus for use by the unregulated inverter A constituting the second stage II. The new three-stage power source generally uses a DC-DC stage I converter to produce the DC voltage indicated as line 14 in FIGURES 1-3. A stage I DC-DC converter can be selected to create the desired voltage on line 12. Three of these converters are shown in FIGURES 4-6 in which the input rectifier 60 provides DC voltage on Lines 60a, 60b for the DC-DC converter, which can be a boost converter 62, a lower converter 64 or a lowering converter 66, as shown in FIGURE 4, FIGURE 5 and FIGURE 6. Using these converters, the DC-DC stage I converter includes a power factor correction system, which allows the power factor to be corrected thereby reducing harmonic interference at the power source input. The use of an input DC-DC converter with power factor correction is well known in the art of welding and is used in many two-stage topographies from the prior art. Converters 62, 64 and 66 preferably include a power factor correction system; however, this is not required. The main task of stage I is to provide a DC bus (DC # 1) on line 12, which bus is indicated as lines 14a, 14b in FIGURES 4-6 to produce a fixed rail

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DC (DC # 2) on line 12 indicated by lines 20a, 20b in the same figures. Power factor correction is not required to use the new three-stage topography. The input stage without power factor correction is shown in FIGURE 7, where the output lines 60a, 60b of the rectifier 60 are connected via a large storage capacitor 68 to create an overall steady voltage on lines 14a, 14b. Stage I in FIGURE 7 does not include a power factor correction circuit or system. However, the power source still requires three stages, where the second stage is an unregulated insulated inverter A producing a substantially steady voltage on lines 20a, 20b. Another modification of the input stage I is shown in FIGURE 8, where a passive power factor correction circuit 70 is connected to the three-phase AC input L1, L2 and L3 to produce a substantially constant DC voltage on lines 14a, 14b, which lines are DC bus 14 (DC # 1) at the input of inverter A. The disclosures of the modified stage I in FIGURES 4-8 are representative only and other input stages with either a single-phase or three-phase input signal and with or without power factor correction may be used.

[0035] By providing a low steady-state voltage on the output rail 20 shown as lines 20a, 20b, the third stage of the new three-stage welding power source may be a contact breaker or other converter operating at a frequency greater than 18 kHz. The switching frequencies of the unregulated inverter and the regulated output converter can be different. Indeed, normally the switching frequency of the contact chopper is substantially less than the frequency of the unregulated inverter A. The power supply PS4 shown in FIGURE 9 illustrates the use of the present invention in which stage III is a standard adjustable converter 100 of the type used for electric arc welding. This converter is powered by a fixed DC input bus 20 and is regulated by feedback from welding operation 120 to provide welding current between output terminals 102, 104. Terminal 102 is a positive polarity terminal and terminal 104 is a negative polarity terminal. In accordance with standard output technology for power sources based on a two-stage inverter, terminals 102, 104 are directed to a standard polarity switch 110. This switch has the first position in which terminal 102 is

Directed to the welding electrode 120 such that the output of the polarity switch 110 has a positive polarity on output line 110a and a negative polarity on output line 110b. This produces a positive electrode DC welding process in welding operation 120. Reversing polarity of mains switch 110 may produce a negative DC welding process in welding operation 120. Thus, the DC welding process with either DC negative or DC positive can be performed according to the setting of the standard polarity switch 110. In a similar manner, the polarity switch 110 can be changed between a negative electrode and a positive electrode to produce an AC welding process in welding operation 120. It is a standard technology in which the polarity switch 110 powers the DC output from the adjustable converter 100 to produce either the AC welding process or the DC welding process. This process is regulated and controlled by a feedback system indicated as a line or loop 122 directed to controller 130 to adjust transducer 100 and to set the polarity of switch 110 as indicated on lines 132, 134, respectively. By regulating welding operation in stage III, the unregulated inverter in stage II can have a relatively high switching frequency to reduce the size of elements within the second stage of the power source, and can have nearly 100% duty cycle to improve efficiency. A preferred embodiment of the three-stage power source uses waveform control technology developed by The Lincoln Electric Company of Cleveland, Ohio. This type of control system is well known and is schematically shown in FIGURE 9A in which the control circuit 150 processes the waveform profile when the voltage on the line 152a is sent from the waveform generator 152. The waveform profile is controlled by feedback loop 122 as schematically represented by error amplifier 154 having output 156. Thereby, the waveform profile of the generator 152 is controlled by the feedback loop 122 and produces a signal on the output line 156. This line is routed to the corresponding circuit 160 of the pulse width modulator operating at a high frequency determined by the output of the oscillator 162. This frequency is greater than 18 kHz and is often higher than 40 kHz. The adjustable transducer 100 preferably operates at 40 kHz. The pulse width modulator output, which is normally a digital circuit within controller 130, is represented as line 132 for controlling the waveform after

55 / 95P31768EN1 by means of an adjustable inverter 100. In accordance with standard practice, the waveform of the inverter 100 may have any profile, either AC or DC. This feature is schematically represented as wave shape 152b, 152c and 152d in the right part of FIGURE 9A. Waveform 152b is the AC waveform of the type used in MIG AC welding, where higher negative ampere values ​​are provided on the electrode. Higher positive amps are also common. Wave-shaped 152c, the ampere values ​​for both the negative electrode and the positive electrode are essentially the same with the length of the negative electrode portion being greater. Of course, the AC welding process can be adjusted to provide balanced AC waveforms or unbalanced AC waveforms, either in favor of a negative electrode or a positive electrode. When polarity switch 110 is set to either DC negative or DC positive welding operation, the pulse welding waveform shown as waveform 152d is controlled by waveform generator 152. Various other waveforms, both AC and DC, can be controlled by the controller 130 so that the welding operation 120 can be adjusted to be AC ​​or DC. In addition, the welding operation may be of the TIG, MIG, submerged arc type or the like. Any process can be performed by a PS4 power source or other power sources using the present invention. The electrode can be wear-free or wear-free, such as with a metal core, flux core or solid rod. Shielding gas may or may not be used in accordance with the electrode used. A modification of the PS4 power source for performing only DC welding is shown as the PS5 power source in FIGURE 10. In this power source, the welding operation 120 only performs the DC welding operation so that the feedback loop 122 is directed to the controller 170 having an output 172. The adjustable converter 100a is preferably a contact chopper for generating DC voltage between lines 102a, 104a. The controller 170 is controlled by the waveform generator 152 as shown in FIGURE 9A. The polarization on lines 102a, 104a is either negative or positive electrode as required by the DC welding process performed in welding operation 120. The adjustable converter 100a is simpler than the welding output of the PS4 power source shown in FIGURE 9. FIGURES 9 and 10, together with the network control or circuit 150 shown in FIGURE 9A show the versatility of the new three-stage power source.

[0036] It is necessary to provide a voltage for controllers for both regulated and unregulated switching networks used in these two types of power sources. FIGURE 11 shows the architecture and diagram used to obtain the control voltages for operating various controllers of a three-stage power source, such as the PS6 power source. The use of a regulator pre-output to provide a control voltage for the regulator pre-switching controller and the second stage switching controller of a two-stage power source is well known and is disclosed in US 5,926,381 (Moriguchi). The output contact breaker for performing welding operations routinely obtains the driver control voltage from the DC input voltage for the contact breaker. These two well-known technologies are included in the PS6 power source. The three-stage power source can work with controllers having power sources designated from different locations in the power source. More precisely, the PS6 power source has a power source 180 with output 182 and inputs 184, 186 from the first DC bus on terminals 14a, 14b (DC # 1). Power source 180 includes a down converter or flyback converter, not shown, to reduce the high voltage output of the pre-regulator 40 of FIGURE 2 to the low voltage on line 182. This control voltage may be between 5 and 20 Volts. Voltage on line 182 is directed to a controller 190 having an output terminal 192 for performing operations of the pre-controller 40 in accordance with standard technology. The pre-regulator has the feedback control lines 42, 44 shown in FIGURES 2 and 3, but omitted in FIGURE 11. The unregulated inverter A does not require a controller to modulate the fill factor or the established relationship between input and output voltage. However, it requires controller 194, which receives controller operating voltage on line 196 from power source 180. This arrangement is similar to the concept disclosed in US 5,926,381 (Moriguchi), except that the second stage controller 194 is not a control controller used in the prior art two stage power source. Alternatively, the PS # 3 power source is powered through a single-phase input 12 to provide the optional power source voltage shown as dashed line 176. The adjustable 30 degree III output converter has a 200 power source marked PS # 2 with driver voltage on line 202 determined by the voltage on the DC 20 bus (DC # 2) shown as containing the terminals

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20a, 20b. Again, power source 200 includes a lowering converter or flyback converter to transform the DC bus at the output of the unregulated converter A to a lower voltage for use by the controller 210 having output 212. The signal on line 212 regulates the output of the welding converter according to the feedback signal on line C, as discussed with reference to power sources PS1, PS2 in FIGURES 1 and 2, respectively. The DC 14 (DC # 1) bus and DC 20 (DC # 2) bus provide input for 180, 200 power sources that are DC-DC converters to create low-level DC control voltage for 190, 194 and 210 controllers. Alternatively as shown using dashed line 220, power source 180 designated PS # 2 may provide control voltage for controller 210. FIGURE 11 is presented to illustrate the versatility of using a three-stage power source with controllers that can receive reduced supply voltages from various fixed DC voltage levels indicated as PS # 1 and PS # 2. Other circuits can be used to provide control voltage, such as a straightforward connection to one phase of the AC 12 input voltage through a transformer as shown in PS # 3.

[0037] The PS7 power source in FIGURE 12 is similar to the PS6 power source with elements having the same identification numbers. Output stage III is a contact chopper 230 for directing the DC current between the E electrode and the workpiece W. The current siding S provides a C feedback signal to controller 210. The high speed II switching inverter 240 has characteristics previously described with insulation provided by a transformer 250 having a primary winding 252 and a secondary winding 254. The primary side of the DC-DC 240 converter is a switching network directing alternating current to the primary winding 252. Straight output with secondary 254 is the secondary section or side of the inverter 240. The inverter 240 uses a high switching speed inverter that has a duty cycle or phase shift set by controller 194. The switching frequency is about 100kHz in the practical version of this power source. The fill factor remains the same during the welding operation by contact breaker 230; however, the fill factor or phase shift of the inverter can be adjusted as indicated by the "ADJ" circuit 260 having output 262 to adjust controller 194. The fill factor is usually close to 100%

So that the pairs of switches are conductive together during their maximum times on the primary side of the inverter 240. However, in order to change the established relationship between the first DC bus 14 and the second DC bus 20, circuit 260 can be used for adjusting the duty cycle or phase shift. Thus, the unregulated, isolation inverter 240 is changed to have a different but fixed filling ratio. However, the fill factor is normally close to 100% so that the switch pairs work essentially in accordance. The duty cycle probably varies between 80-100% in normal applications of the three-stage power source. In a preferred implementation of the new power source, the boost converter 62 shown in FIGURE 4 is used for input stage I power factor correction. This boost converter operates according to controller 190 having the control voltage 182 as previously defined. According to a slight modification, power supply 270 has a transformer connected via lines 274 between one phase of a single-phase or three-phase AC input 12. The rectifier and filter in the power source 270 generate low control voltage in the optimal dashed line 276 to be used instead of the control voltage on line 182 if it is desirable. These two alternatives do not affect the operating characteristics of the PS7 power source. Other such modifications to the three-phase power source for electric arc welding can be obtained from the previous description and well-known welding technology.

[0038] Input stage I normally includes a rectifier and a DC-DC converter with power factor correction as shown in FIGURES 4-8. These input stages can be used for both three-phase and single-phase AC signals of various sizes, represented as input 12. Some aspects of the input stage for three-phase AC power input are described with reference to the circuits in FIGURES 13-16. Each of these circuits has a three-phase DC bus input and output (DC # 1) which is obtained with a low harmonic distortion factor and a high power factor for the input stage. The disclosure in FIGURES 1-12 can generally be applied to a new three-stage power source; however, the specific stage I used is significant for both stages of the power source. In FIGURE 13, the input circuit 300 degree I comprises a three phase rectifier 302 with output terminals 302a, 302b. Boost switch 310 is in series

With a coil 312, a diode 314 and a parallel capacitor 316. The corresponding circuit 320, which is a standard power factor correction system, has an input 322 for determining the input voltage, a feedback control line 322a and an output 324 for operating the switch boost to cause current at input 12 to be in phase with input voltage in general. This system is a standard three-phase booster converter with power factor correction that can be used in the present invention and is also used for a normal two-stage power source. In a similar manner, the input circuit 330 shown in FIGURE 14 has a three-phase rectifier 302 with output terminals 302a, 302b as previously described. The booster circuit containing coil 350, diodes 352, 354 and capacitors 356, 358 is used in combination with switches 340, 342 to ensure coordination of the current at the output of the circuit 330 and the input voltage 12. To achieve this, the control system 360 provides gating pulses on lines 362, 364 according to the detected voltage at input 366 and feedback signals on lines 367, 368. It is a standard technology that provides three-phase power factor correction of the type that creates a two-stage power source input or a new three-stage power source. It has been found that active three-phase circuits 300, 330 when operating on a three-phase input provide an input power factor of 0.95. Stage I power factor when a single phase AC input is available can be corrected up to about 0.99. Since the three-phase power source can generally only be adjusted to a lower level, it has been found that the passive circuit for the input stage I two-stage or three-stage power source is somewhat commensurate with the ability of the active power factor correction circuit. The standard passive circuit 400 is shown in FIGURE 15, in which each of the three phases is rectified by a three-phase rectifier 302 which directs the DC current through the output terminals 302a, 302b to a filter circuit comprising a coil 412 and a capacitor 414. It has been found that a passive circuit such as that shown in FIGURE 15 can correct the power factor of the three-phase input to a level substantially in the range of about 0.95. This is somewhat the same as the ability of the active circuit for a three-phase input circuit. Input-lowering-boosting circuit 420 is shown in FIGURE 16. Rectified current at

Lines 302a, 302b is first lowered by switch 422 using standard power factor correction 430 having line # 32 having a voltage waveform signal from input 12, which also controls system 434 to operate the boost switch 440. Switches 422, 440 work together to control the input power factor using a circuit comprising a coil 450, a diode 454 and a capacitor 454. The 300, 330, 400 and 420 circuits are standard three-phase passive power factor correction circuits using standard technology and available switches controlled by the input voltage waveform and DC # 1 current. FIGURES 13-16 show some modifications that can be made to the first stage of the three stage power source. Of course, there is another technology to improve the power factor and reduce harmonic distortion of both DC and AC signals of the type used to direct power sources for electric arc welders.

[0039] The unregulated A stage II inverter may use different inverter circuits. A preferred circuit is shown in FIGURE 17 in which the inverter is divided between the primary section or side determined by the input to the primary winding 252 of the isolation transformer 250 and the secondary section or side determined by the output of the secondary winding 254. Referring first to the original section or side of the inverter A, a full bridge circuit 500 is used, in which the paired switches SW1-SW3 and SW2-SW4 between capacitor 548 are connected via terminals 502, 504. The switches are powered in an alternating sequence by gating pulses on lines 510, 512, 514 and 516 respectively. The controller 194 generates output gating pulses on lines 510-516 and an adjusted duty cycle as determined by logic on line 262 from circuit 260 as previously discussed. Fill factor is controlled by changing the phase shift of lines 510 and 512 and lines 514 and 516. Circuit 260 regulates the fill factor or phase shift of paired switches. This adjustment is determined during operation of the inverter A. In practice, the circuit 500 has about 100% duty cycle or phase shift, in which each pair of switches has maximum periods of overlapping conduction phases. The controller 194 has a control voltage from the corresponding power supply indicated by lines 196, as previously described. During circuit 500 operation, alternating current is directed

Through the primary winding 252. This current has a very high frequency, normally at least about 100 kHz, so that the size, weight and cost of the elements can be reduced. High switching frequency is not guided by the welding operation, but is selected for the efficiency of the unregulated stage A of the three-phase power source. The blocking capacitor is in series with the primary winding to prevent saturation with unregulated gate triggering signals. The secondary section or side of the inverter A is a rectifier 520 having synchronous rectifiers 522, 524. Synchronous rectifiers are well known in the more general electrical design art and are discussed in US 6,618,274 (Boylan). These devices are gated by signals on lines 526, 528 formed at opposite ends of the secondary winding 254 in accordance with standard technology. Terminals 530, 532 and 534 form the output terminals of rectifier 520 for creating DC voltage (DC # 2) between terminals 20a, 20b. The current is smoothed by a choke 544 and a transverse capacitor 546, according to standard welding technology. Inverter A is unregulated, which means that it is not tuned by the real-time feedback signal from the welding operation. It only converts the DC 12 (DC # 1) bus to the DC 20 (DC # 2) bus. This conversion allows for a substantial reduction of the voltage directed to the regulated third stage power source using the inverter A. The voltage reduction is first determined by the transformer ratio gear 250, whose ratio in the preferred embodiment is about 4: 1. For DC # 1, the voltage is about 400 Volts. Thus, the determined voltage on the output rail 20 is about 1/4 of the fixed voltage on the first stage output rail 12. Several advantages of an unregulated degree are included in the article entitled The incredible Shrinking (Unregulated) Power Supply by Dr. Ray Ridley presented here as a reference. The main advantage is the ability to increase the frequency to above 100 kHz in order to reduce the size and cost of the inverter stage.

[0040] Various circuits can be used for the unregulated inverter A constituting the new stage II of the invention. The specified inverter type is not controlling. Several types of inverters were used. Some are shown in FIGURES 1821. In FIGURE 18, inverter A is shown as employing a full rectification circuit 600 on the primary side of the transformer 250. The parallel diode and switch 602, 604, 606 and 608 operate according to standard

Full bridge phase shift technology as explained for inverter A in the version shown in FIGURE 17. A modification of the internal functionality for inverter A is shown in FIGURE 19 using a cascade bridge with series mounted switching circuits 610, 612 and 614, 616. These switching circuits work similarly with one-half bridge and contain input capacitors 548a, 548b providing energy for switching circuits that are in parallel with capacitor 620 and in series with diode 622, 624. Two switching circuits are in series so that there is reduced voltage at each switch when a phase shift control technique similar to the one with the full bridge inverter of FIGURE 17 is used. This type of switching network with an inverter is disclosed in US 6,349,044 (Canales-Abarca) showing an inverter using a cascade bridge, sometimes referred to as a three-level inverter. A double inverter double forward inverter is shown in FIGURE 20, in which switches 630, 632 provide pulses in section 252a of the primary winding for transformer 250a. In a similar manner, switches 634, 636 operate in unison to provide counter polarity pulses in primary section 252b. Alternating pulses generate AC on the primary winding of transformer 250a to produce an isolated DC output on the secondary winding 254. The standard half-wave bridge is shown as inverter A architecture in FIGURE 21. This one-half bridge includes switches 640, 642 alternately switched to produce AC on the primary winding 252 of transformer 250. These and other switching circuits can be used to provide the AC signal on the primary winding of transformer 250 so that the secondary isolated AC signal is rectified and sent on the output of terminals 20a, 20b as DC # 2. The description of some representative standard switching networks is not considered exhaustive but illustrative. Welding current control is not performed in the second stage. To this extent, the DC bus having a high voltage is converted to a fixed DC bus (DC # 2) having a low voltage to power the third stage, which third stage is an adjustable stage to provide a current suitable for arc welding. Electric arc welding includes and is intended to cover other welding applications such as the concept of plasma cutting.

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Different circuits used in three stages can be combined to create different architectures for basic topography, which is a three-stage power source.

[0041] In FIGURE 22, the first two stages of the improved three-stage power source include an unregulated converter A as best shown in FIGURE 17, in which the DC input signal between lines 14a, 14b is provided by a new first input stage shown as boost converter 600 having switch 602 power switched by gate signal on line 604. Switch 602 is turned on when auxiliary switch 628 is turned on. Time synchronization of gating signals on lines 192 and 192a is performed by a power factor correction controller 194. High frequency signal on line 192 produces high frequency switching signal on gate 604 of main power switch 602 with reverse diode 602a, according to standard boosting technology. Time synchronization at gate 604 is controlled as discussed above to obtain power factor correction for the power source creating the rectified signal at the input terminals 12a, 12b. The DC signal on terminals 12a, 12b is transformed by switch 602 and output rectifying diode 610 on the DC bus on terminals 14a, 14b. The invention requires the use of an active soft switching circuit 620 having a network comprising a first branch with a coil 622 and a second branch with a parasitic capacitor 624. The network is activated by a series connected auxiliary switch 628. Certain discussions identify this two-branch network as a vibration circuit or a resonant circuit. This is technically justified but it is not necessary for the soft switching function. Capacitor 624 and coil 622 form a filter circuit for soft switching 601, in which the capacitor 640 softly energizes the boost diode 610 with the diode D2. This boost diode is sometimes referred to as the output or rectifier diode. Circuit 620 is an active soft switching circuit that controls voltage and current at the power switch 602 in switching events and also at the output diode 610. Thus, the power switch 606 and the boost or output diode 610 in the boost converter 600 are switched by soft switches. This feature makes the switching technique particularly attractive for high voltage conversion applications where the boost diode suffers from

Various recovery blocking problems. For example, in a boosting circuit with power factor correction, both the power switch and the rectifier diode are subjected to high voltages. With the traditional pulse width modulation technique used, as a result of recovering blocking of the minor rectifying diode 610, fast switching losses, high EMI noise and device failure problems become clearer. Thus, the implementation of soft switching is preferred for both power switch 602 and diode 610. The voltage and current waveforms of the switches in the inverter are generally square waves beyond the periods of switching on and off of the switches when the zero voltage switching transition takes place. Both the power switch and the boost diode are subjected to minimal voltage and current loads. Auxiliary switch 628 can be very small compared to the main switch as it only supports small amounts of resonance transition energy. Because soft switching is achieved without increasing the switching voltage and current loads, there is no substantial increase in conduction losses when using active circuits 620. Basically, circuit 620 is selected to provide soft switching of both current and voltage during the pass of the power switch 602 and, optionally, at the transitions of the output diode 610. Thus, a two-stage converter is used to convert the DC signal on lines 12a, 12b to DC signal on lines 20a, 20b. The performance of this two-stage device is drastically increased by having a soft-switching circuit in the boost converter 600 and using the proper soft-switching circuit of the unregulated inverter A. As a consequence, the two-stage DC-DC converter shown in FIGURE 22 is substantially improved for the input side of a three-stage power source for welding. In operation, the high frequency switching signal on line 192, which signal exceeds 18 kHz, first powers auxiliary switch 628 through the gate signal on line 192a to activate the resonant oscillating circuit formed by coil 622 and capacitor 624. When switch 628 is turned on, is the main switch 602. This softly switches both current and voltage. At the same time, the passive part of the circuit 620 controls the voltage and current at the output rectifier diode 610. The positive polarity side of the auxiliary switch 628 is connected to the capacitor

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640 through the D1 diode. This connects the soft switching circuit to the passive output, while the circuit containing the inductance and capacitive branches does not change during operation. The circuit shown in FIGURE 22 is discussed in 1991 IEEE article High Efficiency Telecom Rectifier Using a Novel Soft-Switching Boost-Based Input Current Shaper. A similar soft switching circuit for the 602 power switch is described in the 2004 IEEE article entitled A New ZVT-ZCT-PWM DC-DC Converter. This type of similar active soft switching circuit used for the power switch 602 is shown in FIGURE 23, in which the numbers for the same elements as those shown in FIGURE 22 are the same. The active soft switching circuit 700 has resonant coils 704, 706 divided into segments and coupled with a common core 705. Current control diodes 704a, 706a, respectively. These diodes are in series with the coils, which in turn are parallel to the parasitic capacity 708. Auxiliary switch 710 has a reverse diode 712, so that switch 710 operates in accordance with the previously discussed auxiliary switch 628 of FIGURE 22. The soft switching circuit 700 includes a voltage control capacitor 720 to control the voltage at the output rectifier diode 610. To connect the positive side of the auxiliary switch 710 to the output terminal 14a, a single diode 730 is provided. This diode works as the diode D1, D2 in FIGURE 22. Soft switching circuit 700 provides soft switching of both voltage and current at power switch 602 and controls voltage and current when switching rectifier diode 610. Thus, circuit 700 essentially operates in the same manner as previously discussed soft switching circuit 600. The present invention requires an active soft switching circuit for the power switch 610 and optionally for the rectifier diode 610. The topography for soft switching circuits may vary, with two of the preferred soft switching circuits 600, 700 shown in FIGURES 22, 23, respectively. Sw1, SW2, SW2 and SW4 switches are semiconductor reverse switch diodes such as diode 602a. In addition, capacitor 506a prevents saturation of transformer core 250a.

[0042] By providing an active soft switching circuit for an upstream three-stage power source, the operation of the input stage combines with the characteristics of proper second-stage soft switching

An unregulated inverter to provide a two-stage input that improves the performance of a new three-stage power source as described in FIGURES 1-21. Circuit 700 has been found to push the voltage to near zero during fast switching 602. Circuit 600 lowers voltage, but the voltage when switching using circuit 600 is not exactly zero. In fact, it can be around 50 Volts high. Consequently, the soft switching circuit 600 is advantageous because of its lower cost and the soft switching circuit 700 is alternative because of its ability to push the current voltage near zero when switching the switch 602. These differences are the reason to illustrate two separate active soft switching circuits for use on the input stage of a new three-stage power source as described above.

[0043] The three-phase power source described in FIGURE 12 is again shown in FIGURE 24, in which the contact chopper 230 is shown as having a switch 750 controlled by high frequency gating signal on line 212 from controller 210. Feedback signal on line 762 from sensor 760 current is generated by reading the shunt S. In a similar manner, the voltage feedback signal is routed to controller 210 via line 772 from the voltage sensor 770. These two feedback signals control the operation of the pulse width modulator in the controller 210 to control the power interrupter switch 750. The input capacitor 780 controls the voltage between terminals 20a, 20b in accordance with standard practice. An optional aspect of the invention is to provide a passive soft switching circuit 800 for a contact chopper 230 in which the passive soft switching of the contact chopper is combined with active soft switching of the input stage and proper soft switching of the second stage to increase the efficiency of the three stage power source shown in FIGURE 12 and described in FIGURES 1-21. The soft switching circuit 800 is a commonly used soft switching circuit. The circuit contains an 802 coil to control the current in the power switch and the D4 diode. Capacitor 806 controls the voltage at the power switch during the switching operation. Capacitors 804 and 806 are connected via diodes D1, D2, D3 and D4. These two capacitors control the voltage across the D4 diode. Thus, the 750 power switch and the D4 diode are softly switched on both current and

Voltage during the switching operation. This circuit is shown in an article by the University of California entitled Properties and Synthesis of Passive, Lossless Soft-Switching PWM Converters. This May 1997 article explains the work of commonly used 800 soft switching passive circuits. In short, the contact breaker 230 has a power switch with a soft switching circuit to control both current and voltage during the on and off transitions. In other words, the output contact breaker 230 is provided with a soft switching circuit, which soft switching circuit controls both voltage and current in a timely manner during the switching operation.

[0044] The three stage power source described in FIGURES 1-21 is provided with an input stage having an active soft switching circuit that combines with the proper soft switching of an unregulated second stage inverter A to increase overall efficiency by reducing the switching losses and conduction losses on the input side power sources. Optionally, the output stage of the contact interrupter is equipped with a passive soft switching circuit to provide an inexpensive end stage. The contact breaker can be a separate, replaceable module that does not require circuit modification to control an auxiliary switch as required in an active soft-switching circuit. The input part of the three-stage power source contains an active power factor correction stage combined with an unregulated degree of phase shift pulse width modulation. This new combination of the first two stages is highly efficient and inexpensive as the topography of electric arc welding machines.

[0045] As shown in FIGURES 22, 25 and 26, the first stage 600 is of the DC-DC boost converter type having a coil 644 connected between the input terminal 12a and the main internal node 603, the main switching device 602 with the internal diode 602a connected between the internal node 603 and the lower input terminal 12b of the inverter. The main rectifier diode 610 is connected to its anode at node 603 and its cathode at output terminal 14a. An optional output filter capacitor 548 is connected between the output terminals 14a, 14b. As in the normal operation mode of the boost converter, the main switch 602 is activated by the pulse width modulated (PWM) control signal on its gate

To switch between a forward state (ON) in which the internal node 603 is brought substantially to voltage at the lower terminal 12b (charging phase) and a non-forward state (OFF) (discharge phase). Before each charging phase, assuming that the main switching device 602 was in a non-conduction state (OFF) for a relatively long time, the voltage at the output capacitor 548 is equal to the input voltage plus the voltage on the coil 644. Closing the main switch 602 brings the node 603 essentially to the lower voltage of terminal 12b, where the input voltage is caused on coil 644 (terminal 12a is positive with respect to terminal 603) and the main diode 610 prevents the filtering capacitor 548 from being discharged by the main switch 602. Voltage on the coil 644 causes an increase in current flow over time, with adequate energy stored in the 644 coil. Then, main switch 602 is deactivated (OFF) to start the discharge phase. Placing switch 602 in a non-conduction state causes the voltage on the main coil to change so that the voltage at node 603 increases to maintain the current flowing through the coil 644 at a constant value, while the current in the coil continues to flow and the voltage at node 603 it must be sufficient to increase polarization in the direction of conduction of the 610 boost diode (for example, about the voltage on capacitor 548 plus voltage drop on the diode), wherein the coil voltage changes polarization in the discharge phase. For high output capacity 548, the output voltage between terminals 14a and 14b remains substantially constant during the discharge phase, with the charging and discharging (switching on and off of the main switch 602) repeated with the appropriate feedback loop to adjust the pulse width modulation of the switch control signal that the output voltage at capacitor 548 can be maintained at the desired DC value.

[0046] In general, it is desirable to maximize the efficiency of each stage in a power source in which the resistance parameters in the state of the main switch 602 on, the diode forward voltage drop and the blocking recovery time for the main diode 610 are ideally minimized to combat conduction losses. It is also possible to minimize switching losses and noise generation at the inverter level 600, in which it is desirable to control the conditions under which state transitions of switch 602 and diode 610 occur. In particular, soft switching circuits can be advantageously used in

The boost converter 600 to ensure that the zero voltage switching is turned on and off, as well as the diode 610 for zero voltage or zero current are turned off. No countermeasures, switching the main switch 602 causes unwanted power loss and load for the switch 602 and / or for the main diode 610. Accordingly, the soft switching or suppression circuitry is utilized in step 600 of the boost converter to provide low current and low voltage switching of these components. In the present invention, the soft switching circuitry may be used to minimize voltage rise rate at switch 602 (e.g., dv / dt at node 603) when switch 602 is turned off, and to minimize voltage at switch 602 when switch 602 is turned on, such as also to minimize one or both of the diode 610 voltage or current while inverting, to minimize switching losses and noise emissions.

[0047] The soft start switching system shown in US 5,418,704 (Hua) can be used to increase the three-stage power source 600 as schematically shown in FIGURE 24. This patent differs from the first embodiment of the circuit shown in FIGURE 25 and the preferred embodiment of the circuit shown in FIGURE 26. The soft switching circuit described in US 5,418,704 (Hua) uses an auxiliary switch with a resonant coil and a capacitor to provide zero switching voltage of the boost converter main switch and the output diode. This is a publication relating to the two-branch network of the invention as a resonant circuit. In Hua, the auxiliary switch and resonant coil are connected in series between the inverter main switch. The auxiliary switch is turned on just before turning on the main switch so that the resonant coil is diode connected to the positive output terminal of the inverter to limit the rate of change of the main diode current. Activation of the secondary switch in Hua also discharged the internal node to zero volts, thereby ensuring that the main switch was turned on at essentially zero voltage. However, Hua suffers from a hard switching state when turning off the main transistor. In particular, the voltage at the top of the main switch in Hua must be higher than the inverter output voltage before the coil

The resonant conductor can conduct any current to the output, where the Hua resonant coil causes a very fast increase of the transistor voltage (high dv / dt) during transistor shutdown, leading to unacceptable switching losses.

[0048] As shown in FIGURES 25 and 26, exemplary booster converter stage 600 includes an active soft switching system 601 or 601a, respectively, for providing soft switching of main switch 602 and main diode 610. The exemplary soft switching system 606 in FIGURE 25, which is the original version of the invention, is a three-terminal network having a first and second terminal connected between the main switch 602 and a third terminal connected to the main cathode of the diode 610. The soft switching circuit or network comprises a coil 622, an auxiliary switching device 628 with diode 630. The first and second diodes D1 and D2, together with capacitors 624 and 640 form a three-terminal suppression circuit 606. Main and auxiliary switching devices 602 and 628 may be any suitable devices that selectively provide generally conductive and generally non-conductive states between its first and second power terminals according to a control signal in its control terminal, including, but not limited to, bipolar transistors, type devices metal oxide semiconductor (MOS), bipolar insulated gate transistors (IGBTs) and the like. Coil 622 is in the first branch in parallel with switch 602. Coil 622 has a first terminal connected to main coil 644 and a second terminal attached to the first intermediate node 607 of the circuit. Auxiliary switching device 628 is connected between node 607 and inverter terminals 12b, 14b. The diode 630 may be an internal diode of the auxiliary switching device 628 or it may be a separate element. Diode anode 630 is connected to the bottom terminals 12b, 14b of the inverter and its cathode is connected to node 607 when connecting auxiliary switch 628 and resonant coil 622. Similar to Hua circuit, one capacitor 624 is connected in circuit 606 on main switch 602. Unlike to HUA, however, soft switching circuit 606 in FIGURE 25 has a second intermediate node 609 with a second capacitor 640 connected between nodes 603 and 609. The first diode D1 of the soft switching circuit or network 601 has an anode connected to the first

Internal node 607 and cathode connected to the second internal node

609. Diode D2 has an anode connected to the second internal node 609 and a cathode connected to the cathode of the main diode 610 at the upper output terminal 14a of the inverter.

[0049] As a technical advantage over Hua with its hard switching of the auxiliary switch, the soft switching circuit 601 of FIGURE 25 provides a soft switching operation for turning on and off both the main switch 602 and the main diode 610 as well as the auxiliary switch 628. Such improvement achieves better performance, less load on components and less noise generation. Before turning on the main switch 602, the auxiliary switch 628 is turned on while the voltage at the node 603 is equal to the output voltage, where the closing of the auxiliary switch 628 causes an increase in current flowing through the resonant coil 622 initially to the main coil current level through which the main diode 610 is inverted. When the diode 610 recovers the blocking of voltage and starts blocking the current from the output, the current from the coils 644 and 622 discharges the capacitor 624, while the voltage on the diode 610 remains low during inversion in order to minimize the diode switching loss and generating noise. The main switch 602 is then turned on when the capacitor 624 is discharged (e.g., when the voltage at the switch 602 is zero) and the auxiliary switch 628 is turned off. The current flowing through the resonant coil 622 charges the first resonant capacitor 640 through the diode D1 and also charges any parasitic capacity of the auxiliary switch 628, where the voltage on node 607 and 609 increases to the inverter output level and the diode D2 begins to conduct. Any remaining energy from coil 622 is supplied to the output via diodes D1 and D2. The main switch 602 is then turned off (at a time depending on the current pulse width modulation based on the output level feedback) while the switching voltage is essentially zero. Current flowing through main coil 644 charges capacitor 624 and discharges resonant capacitor 640 through diode D2. The action on causes the voltage in node 607 to increase to the output value, after which the main diode 610 again begins to conduct current to the output.

[0050] During operation of circuit 606 of FIGURE 25, the main coil current flows through the capacitor 640 and the second diode D2 when the main switch 602 is

Initially off, where the main diode 610 begins to conduct after resonance capacitor 640 is discharged, the voltage on the first capacitor 640 is a function of its capacity, main current level and duty cycle of main switch 602 modulated by pulse width . In this way, the switching loss of the main diode 610 can be reduced or minimized by providing zero diode voltage when the diode begins to conduct current to the output capacitor 548. When the main switch 602 is in the on state, the voltage at the first resonant capacitor 640 remains substantially constant because the first diode D1 prevents charging of the capacitor, except when the auxiliary switch 628 is first turned off and the voltage at node 607 is higher than the voltage at capacitor 640. Ideally, main switch 602 has a zero voltage off condition if resonance capacitor 640 is fully discharged during boost phase with switch 602 on. However, main switch 602 will experience a non-zero off voltage if resonance capacitor 640 is not fully discharged. In addition, capacitor 640 can only provide a current bypass path for the auxiliary coil 622 when the auxiliary switch 628 is turned off, without providing sufficient bypass conductor path for parasitic inductances in the auxiliary circuit loop in soft switching circuit 606 of FIGURE 25. As a result, the transition of auxiliary switch 628 from on to off may occur at non-zero voltage, while switching losses and noise generation will be possible, along with possible overload of switch 628.

[0051] FIGURE 26 shows the preferred embodiment and preferred design of the soft switching circuit 606a, and in accordance with the invention, has a removed capacitor 624. A second capacitor 640a is connected between internal node 609 and the lower terminals 12b, 14b of the inverter, while the network capacity results from series connection of capacitors 640 and 640a, wherein this series connection is a branch parallel to the main switch 602. The lower (second) capacitor 640a is connected in parallel to the auxiliary switch 628 via diode D1. In one particular implementation, the lower capacitor 640a is substantially smaller than the upper capacitor 640. Thus, unlike the soft switching network of FIGURE 25, the circuit 601a of FIGURE 26 provides a capacitor 624 between the second internal node

EP 1 704 954 B1

609 and the lower terminals 12b, 14b of the inverter just like two capacitors 640, 640a.

This geometry helps provide soft switching for auxiliary switch 628 (for example, reduces dv / dt on switch 628).

[0052] Referring now to FIGURE 27, graph 800 illustrates various exemplary waveforms associated with the main and auxiliary switches 602 and 628, respectively, at the stage 600 of the boost converter. An example of an active soft switching circuit 601a of FIGURE 26 is also shown. Graph 800 shows the voltage waveform 810 corresponding to the auxiliary switch control voltage signal (e.g., VGS gate signal, VBE base signal, etc., depending on the type of switch), voltage waveform 820 representing the voltage at auxiliary switch 628 (e.g., voltage between internal node 607 and lower terminals 12b, 14b of the inverter), and current waveform 830 representing the current switched by auxiliary switch 628. In addition, graph 800 also provides a voltage waveform 840 showing the voltage control signal for the main switch 602 as well as a voltage waveform 850 representing the voltage at the main switch 602 (e.g., the voltage between node 603 and the lower terminals 12b, 14b of the inverter. [0053] Various discrete times are depicted in a typical switching cycle of the inverter stage 600 on graph 800, including time 870, at which time the main switch 602 is turned off (e.g., falling edge 840 voltage), time 872 when auxiliary switch 628 is activated (rising edge in the control signal 810), and time 874 when auxiliary switch 628 is turned off and main switch 602 is turned on (falling edge on waveform 810 and rising edge on waveform 840). While this is illustrated as if it were being switched simultaneously at time 874, the auxiliary switch 628 may alternatively be turned off before, simultaneously or after the time at which the main switch 602 is turned on, all such implementation variants being considered to fall within the scope of the invention and attached claims. In the embodiment of the circuit shown in FIGURE 26, the main switch 602 is turned off at 870, after which the voltages at the main and auxiliary switches 602 and 628 (e.g., voltages at nodes 603 and 607) increase as indicated in diagram 800 in part 852 and part respectively 822. It should be noted that voltage curve 850 is zero on the main switch 602 during the main switch

5570 switch while time 870, while any corresponding switching losses and / or noise emissions are mitigated. As shown in FIGURE 27, switching voltage curves 820 and 850 remain substantially constant in part 824 and part 854 with a value substantially equal to the voltage value at output filter capacitor 548 (VOUT) until time 872 at which auxiliary switch 628 is turned on (with main switch 602 remain off), with the auxiliary switch voltage dropping to zero at point 826. It should be noted that the auxiliary switch current curve 830 is essentially zero at time 872, while auxiliary switch 628 does not show a significant loss of switching. Thus, at time 874, main switch 602 is turned on again. It should be noted that between time 872 and time 874, the voltage curve 850 of the main switch drops substantially in part 856 to zero before the switch 602 is turned on, while the zero voltage on state is provided to minimize the switching loss and the noise generated by the main switch 602 . In addition, unlike the circuit or network 601 of FIGURE 25 above, the auxiliary current curve 830 initially increases in part 832 after the time 872 turns on the auxiliary switch, but is then reduced to zero in part 834 before the time 874 turns off the auxiliary switch, wherein the auxiliary turn off switch is a soft switching event with minimized (e.g., zero) switching loss and noise emission. The main switch 602 is then turned on at 874 at substantially zero voltage, and the auxiliary switch voltage 820 increases in portion 828 until the current flowing through the resonant coil 622 drops to zero. Then, the cycle continues until the next time 870, at which time the main switch 602 is turned off again, the amount of time the main switch 602 remains on in a given switching cycle can be determined by the output control conditions by pulse width modulation or other appropriate techniques . Circuit 601a of FIGURE 26 provides soft switching of auxiliary switch 628 while auxiliary switch 628 in circuit 601 has a hard shutdown. This is a separate improvement received by the preferred circuit 601a of FIGURE 26.

[0054] The soft switching system or network 601, 601a of FIGURES 25 and 26, respectively, includes two parallel branches parallel to main switch 602

55% of power. The first branch contains the inductance inductance 622 for current control for auxiliary switch 628, switch 602 and diode 610, while the second branch has voltage control capacity on switch 602. In FIGURE 26, this parallel branch is divided between two capacitors, one of which controls the voltage on auxiliary switch 628.

[0055] The capacity of the capacitors 640, 640a of FIGURE 26 is essentially equal to the capacity of the capacitor 624 of FIGURE 25. The capacitor 640 softly switches the switch 628 when it is turned off. When switch 628 is turned off, capacitor 640a is at zero voltage. It charges slowly to ensure a soft shutdown. When switch 628 is turned on, the current in the switch slowly increases through the coil 622 and the diode 610 is slowly turned off by a slow increase in the current in the coil. Thus, the network 601a softly switches the switch 628 during the on and off cycles and controls the current flowing through the boost or output diode 610. This is an improvement over the network 601 of FIGURE 25.