Signal generation method and signal generation device.
Abstract
A transmission method for simultaneously transmitting a first modulating signal and a second modulating signal at the same frequency, whereby data reception quality is improved in a receiving device by performing precoding using a fixed precoding matrix for both signals, and by regularly switching the phase of the first modulating signal and/or the second modulating signal and transmitting the signals.

Term
5.2 yearsleft in the term
Expires 1 December 2031.
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8 claims: 8 independent, 0 dependent
- 1REIVINDICACIONES Habiéndose descrito la invención como antecede, se reclama como propiedad lo contenido en las siguientes reivindicaciones:1. Un método de transmisión, caracterizado porque comprende: seleccionar un método de generación de entre una pluralidad de métodos de generación, los métodos de generación al menos incluyen un primer método de generación y un segundo método de generación;generar, para cada una de una pluralidad de ranuras, una primera señal de transmisión zl y una segunda señal de transmisión z2 a partir de datos de transmisión, la generación llevada a cabo de conformidad con el método de generación seleccionado;y transmitir la primera señal de transmisión zl y la segunda señal de transmisión z2 respectivamente desde una primera antena y una segunda antena al mismo tiempo en la misma frecuencia, la generación incluye: llevar a cabo modulación y con ello generación, para cada una de la pluralidad de ranuras, una primera señal modulada si y una segunda señal modulada s2 a partir de los datos de transmisión;177 aplicar precodificación expresada por una matriz fija F con respecto) a la primera señal modulada si y la segunda señal modulada s2;y llevar a cabo un cambio de fase con respecto a por lo menos una de una señal que resulta de precodificar la primera señal modulada si y una señal que resulta de precodificar la segunda señal modulada s2, mientras se varía regularmente un esquema de cambio de fase para cada una de la pluralidad de ranuras, la matriz F es expresada como: cuando se lleva a cabo la generación de conformidad con el primer método de generación, se aplica un primer esquema de modulación en la modulación, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, se aplica un segundo esquema de modulación en la modulación, el primer esquema de modulación y el segundo esquema de modulación difieren entre sí, y cuando se lleva a cabo la generación de conformidad con el primer método de generación, a en la matriz F toma un primer valor, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, a en la matriz F toma un segundo valor, el primer valor y el segundo valor difieren entre sí, en donde r 178 el primer valor es un número real ^?TÍeréñTé~~á m arro;........... y el segundo valor es un número real diferente a uno.
- 2Un aparato de transmisión, caracterizado porque comprende:una unidad de selección que selecciona un método de generación de entre una pluralidad de métodos de generación, los métodos de generación al menos incluyen un primer método de generación y un segundo método de generación;una unidad de generación que genera, para cada una de una pluralidad de ranuras, una primera señal de transmisión zl y una segunda señal de transmisión z2 a partir de datos de transmisión, la generación llevada a cabo de conformidad con el método de generación seleccionado;y una unidad de transmisión que transmite la primera señal de transmisión zl y la segunda señal de transmisión z2 respectivamente desde una primera antena y una segunda antena al mismo tiempo en la misma frecuencia, la unidad de generación incluye: una unidad de modulación que lleva a cabo modulación y con ello genera, para cada una de la pluralidad de ranuras, una primera señal modulada si y una segunda señal modulada s2 a partir de los datos de transmisión;una unidad de precodificación que aplica precodificación expresada por una matriz fija F con respecto a la primera señal modulada si y la segunda señal modulada 179 S2;y una unidad de cambio de fase que lleva a cabo un cambio de fase con respecto a por lo menos una de una señal que resulta de precodificar la primera señal modulada si y una señal que resulta de precodificar la segunda señal modulada s2, mientras se varía regularmente un esquema de cambio de fase para cada una de la pluralidad de ranuras, la matriz F es expresada como: cuando se lleva a cabo la generación de conformidad con el primer método de generación, se aplica un primer esquema de modulación en la modulación, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, se aplica un segundo esquema de modulación en la modulación, el primer esquema de modulación y el segundo esquema de modulación difieren entre sí, y cuando se lleva a cabo la generación de conformidad con el primer método de generación, a en la matriz F toma un primer valor, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, a en la matriz F toma un segundo valor, el primer valor y el segundo valor difieren entre sí, en donde el primer valor es un número real diferente a uno, y el segundo valor es un número real diferente a uno. 180
- 3Un método de recepción, caracterizado porque comprende:adquirir una señal de recepción obtenida al recibir una primera señal de transmisión zl y una segunda señal de transmisión z2 respectivamente transmitidas desde una primera antena y una segunda antena al mismo tiempo en la misma frecuencia, la primera señal de transmisión zl y la segunda señal de transmisión z2 han sido generadas mediante un proceso de generación predeterminado;y obtener datos de recepción al aplicar un proceso de demodulación correspondiente al proceso de generación predeterminado a la señal de recepción, el proceso de generación predeterminado incluye: seleccionar un método de generación de entre una pluralidad de métodos de generación, los métodos de generación al menos incluyen un primer método de generación y un segundo método de generación;generar, para cada una de una pluralidad de ranuras, una primera señal de transmisión zl y una segunda señal de transmisión z2 a partir de datos de transmisión, la generación llevada a cabo de conformidad con el método de generación seleccionado;y transmitir la primera señal de transmisión zl y la segunda señal de transmisión z2 respectivamente desde una primera antena y una segunda antena al mismo tiempo en la 181 misma frecuencia, la generación incluye: llevar a cabo modulación y con ello generación, para cada una de la pluralidad de ranuras, una primera señal modulada si y una segunda señal modulada s2 a partir de los datos de transmisión;aplicar precodificación expresada por una matriz fija F con respecto a la primera señal modulada si y la segunda señal modulada s2;y llevar a cabo un cambio de fase con respecto a por lo menos una de una señal que resulta de precodificar la primera señal modulada si y una señal que resulta de precodificar la segunda señal modulada s2, mientras se varía regularmente un esquema de cambio de fase para cada una de la pluralidad de ranuras, la matriz F es expresada como: cuando se lleva a cabo la generación de conformidad con el primer método de generación, se aplica un primer esquema de modulación en la modulación, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, se aplica un segundo esquema de modulación en la modulación, el primer esquema de modulación y el segundo esquema de modulación difieren entre sí, y 182 cuando se lleva a cabo la generación—«te—castormidad con el primer método de generación, a en la matriz F toma un primer valor, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, a en la matriz F toma un segundo valor, el primer valor y el segundo valor difieren entre sí, en donde el primer valor es un número real diferente a uno, y el segundo valor es un número real diferente a uno.
- 4Un aparato de recepción, caracterizado porque comprende:una unidad de adquisición que adquiere una señal de recepción obtenida al recibir una primera señal de transmisión zl y una segunda señal de transmisión z2 respectivamente transmitidas desde una primera antena y una segunda antena al mismo tiempo en la misma frecuencia, la primera señal de transmisión zl y la segunda señal de transmisión z2 han sido generadas mediante un proceso de generación predeterminado;y una unidad de demodulación que obtiene datos de recepción al aplicar un proceso de demodulación correspondiente al proceso de generación predeterminado a la señal de recepción, el proceso de generación predeterminado incluye: seleccionar un método de generación de entre una pluralidad de métodos de generación, los métodos de 183 generación al menos incluyen un primer método de geñé±á:ción«y»= un segundo método de generación;generar, para cada una de una pluralidad de ranuras, una primera señal de transmisión zl y una segunda señal de transmisión z2 a partir de datos de transmisión, la generación llevada a cabo de conformidad con el método de generación seleccionado;y transmitir la primera señal de transmisión zl y la segunda señal de transmisión z2 respectivamente desde una primera antena y una segunda antena al mismo tiempo en la misma frecuencia, la generación incluye: llevar a cabo modulación y con ello generación, para cada una de la pluralidad de ranuras, una primera señal modulada si y una segunda señal modulada s2 a partir de los datos de transmisión;aplicar precodificación expresada por una matriz fija F con respecto a la primera señal modulada si y la segunda señal modulada s2;y llevar a cabo un cambio de fase con respecto a por lo menos una de una señal que resulta de precodificar la primera señal modulada si y una señal que resulta de precodificar la segunda señal modulada s2, mientras se varía regularmente un esquema de cambio de fase para cada una de la pluralidad de ranuras, 184 la matriz F es expresada como: axe JO Va 2 +1 axe jo ,JX cuando se lleva a cabo la generación de conformidad 5 con el primer método de generación, se aplica un primer esquema de modulación en la modulación, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, se aplica un segundo esquema de modulación en la modulación, el primer esquema de modulación y el segundo 10 esquema de modulación difieren entre sí, y cuando se lleva a cabo la generación de conformidad con el primer método de generación, a en la matriz F toma un primer valor, y cuando se lleva a cabo la generación de conformidad con el segundo método de generación, a en la 15 matriz F toma un segundo valor, el primer valor y el segundo valor difieren entre sí, en donde el primer valor es un número real diferente a uno, y el segundo valor es un número real diferente a uno. 185 ,\\t : RBSÜMBN DB ΙΑ INVENCIÓN Se describe un método de transmisión que transmite simultáneamente una primera señal modulada y una segunda señal modulada en una frecuencia común realiza la precodificación en
- 55 ambas señales usando tina matriz de precodificación fija y cambia regularmente la fase de al menos una de las señales, mejorando así la calidad de señal de datos recibida en un dispositivo de recepción. 1/55 S £ 2/55 Comerte A j fo-i avos&nbolo I /a avo símbolo | to+1 avo símbolo 3/55 ‘Τ' FIG. 3 4/55 402 403 IMPI INSTITUTO MEXICANO DE LA PROPIEDAD INDUSTRIAL 5/55
- 66/55 FI6.7 i?;;·; 8/55 9/55 10/55 • · · 12/55 A? Ι201Α 13/55 FIG.13 14/55 Fecha y hora $2 15/55 17/55 FI6.17A 18/55 INSTITU L. FIG. 18A CO co ea tu í 19/55 instituto mí:DO O> O Ll- FI6.19A 20/55 FIG, 21 ''J. 22/55 FIG. 22 8. I (D 1= w •g ñ a 23/55 Antena de transmisión #1 C27 24/55 FIG.24A FIG.24B 830 r· · MPIfT INSTÍTL DE 25/55 ión relativa al método de procesamiento de señales Información relativa al método de procesamiento de señales 27/55 ’ IMP INSTITUTO * i, PE LA r £ IN ( ión relativa al método de procesamiento de señales 28/55 FIG. 28 2800 29/55 FIG. 29 2800 30/55 π o c 0) U. O co CD Ll. Tiempo O 'to Ab 'M © b ib ”s to «H to o Ίο ib to ib 'to o Ίο ib to ib •M to o 'to ib ... to % Ü” ''to ib 'to © ~to ib *^to ib ·**» to © ib 'to ib '‘to o Ί íb '‘to «b m © ib Μ íb to o ib *^to ib © A) ib •*S to ib ** to o Λ ib 'to ib “'W to © TO JS Q Φ U. S o -C >> w o x: > TO o JE N TO > TO JZ o Φ LL 31/55 FI6.31 Señal modulada z2 3102 I INSTITUTO '/ZZ'ín DEL T^feí«· 32/55 CM co Tiempo E o J= (Q s o JS > <9 (Q f ¡é ü_ § Λ > (9 X 33/55 TO O C TO O CO co co hHl Lu Tiempo ··% 43 8[® ’tü fe |«h ’ 43 o ••x 43 fe 1« 'i 4 81® «¡|ο 43 o *^3 fe|« 4i 8¡® ^43 fel» ^43 o 43 fe I * · 8|® fe|o 43 O *^3 fe|cn ·* · 4 8i® ^43 fe|o ^43 o fej«n ••S 43 «81® ·% fe|» 43 o '5. 43 fe|*n 43 «s® fe |φ •*s 43 9 -Η 43 fe{c*i ^4 8h *·*» 43 fe | h · 43 o fe) n te 8Í® *4í fe|o» · 43 o % fe|«* *n 4 81® ''m Me» o ~b
- 77—7—ΓΤ X M Λ 3 5 § _C o ra o jz ra -C ra j= Jg ra SZ y TO L_ V * N U. u IL TO «= Ί OT E dhB-U Bloque codificado (6000 bits} 35/55 íliSTl i Bloque codificado (6000 bits) 36/55 ί 37/55 FIG. 37 FIG. 38 Corriente de video (PID=0x1011 video principal) Corriente de audio (PID=Ox1100) Corriente de audio (PID=0x1101) Corriente de presentación de gráficos (PiD=0x1200) Corriente de presentación de gráficos (PID=0x1201) Corriente de gráficos interactivos (PID=0x1400) Corriente de video (PID=0xtB00 video secundario) Corriente de video (PID=0xlB01 video secundario) 39/55 o YD (0 Ό E s Ό OT U C QJ U 0) ω σ> ο o Υ3 ω σ (Λ ÍG Ε £ ω •ο « Ό ο ω 40/55 Oí TO E CD Oí TO E o <4· C5 co c o> O) TO E c Qí Oí TO E Secuencia de tramas de video c Oí Oí TO E c 0) Oí TO E TO Ε?ω to LU O IX o Ό TO N QJ -O
- 88<n c UJ ni o. 41/55 -4 - «i en » £ Ε IS CL «C £ * Q O 0 •a 45 S * o *r ro O o c LU r<o to «o CM Z CL ω (/) o O Q. £ E tfl fi CO Q <u u a 1 £ Γ « <u tr (0 CL Encabezado PMT Desoriptor #1 42/55 FIG. 42 Tipo de corriente Descriptor ffN PID (identificador) Información de corriente #1 Descriptor de corriente #1 Información de corriente #N Descriptor de corriente #N 43/55 FIG. 43 Datos multiplexados (XXX.MZTS) 44/55 45/55 O o (Ο ο 46/55 ____________________ 47155 '•i ? s u_ z < w w IW ¡w vw ss^ w Ih W IB i n / -l· \ E tfí C4 s •r □ o SI ~-W 0 - 4 Si *el«n 'v o *^ SI 'sa *l.n S|« % iW «H *í o X SI ^0» SI ''o, *cj»O s* Si ^0) Sl· •s. V w SI 'o w SH % íl 's» SI- « SI « SH A * SH »l· w;*b n í|« h w SI ksl«o Si Si SH o 1 i / \ 1*7-T T |V~T T s rxf o ·—4 ü_ N a3_ co r*· o <» •SSWfcVI*·» 48/55 41» o w % «Η 5b *te k|a •“X te \ kj n · te 4|» 'te k|o -s % k Λ 'te w k|o. ·*» te k|m 'te 41» *te k|o 'te Ί» kH ' te 4|» •u k| * *te % 0 41» · te w % k|- te 41» ‘te kl» - te % kl n -s te 4|» te teja -x te % k | n i k(o - te w kjn ^te 41» te k|o\ -x te % k|t* te 41» *te k|o\ ^te O 1» W;iW n i _| = / -ί· «< Ο LU Ια w ¿b te »Jr te 4 te rsb te k|«n te 41 ‘te *te k|v» te 0 - te 41« · te -te te ¡w £b ·* te w 41 « te 41« te 41 te kl* te 4H te 4|« te w te % ΛΗ te 41« te 4H te 41 te 4|« te 41 te 4|« te W 41 te w ¿I te s, te 4t te k|v te 4|« 4|« te 4|« te s. te S n i -ι* Τ..........Ί· \ I? / 7-Γ~\ <» ίΜ ' I§ (Β X Sis Μ· 49/55 ,λ'Λ\ 50/55 51/55 53/55 54/55 1] o 5102Α 55/55 FIG. 55
Independent claims8
1,188 paragraphs in 18 sections, as filed
(54) Title: SIGNAL GENERATION METHOD AND SIGNAL GENERATION DEVICE. (54) Title: SIGNAL GENERATION METHOD AND SIGNAL GENERATION DEVICE.
(57) Summary
A transmission method is described that simultaneously transmits a first modulated signal and a second modulated signal on a common frequency, precoding on both signals using a fixed precoding matrix and regularly changing the phase of at least one of the signals, thus improving the quality of data signal received by a receiving device.
(57) Abstract
A transmission method for simultaneously transmitting a first modulating signal and a second modulating signal at the same frequency, whereby data reception quality is improved in a receiving device by performing precoding using a fixed precoding matrix for both signáis, and by regularly switching the phase of the first modulating signal and / or the second modulating signal and transmitting the signáis.
I KNOW
SECRETARIAT M ftOWMÍA
Institute
Mexican Property
Industrial
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PATENT TITLE NO. 337079
Headlines): PANASONIC CORPORATION
Address: 1006, Oaza Kadoma, Kadoma-sh¡, Osaka, 571-8501, JAPAN
Name: SIGNAL GENERATION METHOD AND SIGNAL GENERATION DEVICE
Classification:
ii lnt.CI.8: H04B7 / 04; H04J99 / 00
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01/26/2004, 06/16/2005, 01/25/2006, 06/05/2009, 06/01/2010, 06/18/2010, 06/28/2010, 01/27/2012 and 09 / 04/2012); 1st, 3rd traction articles V subsection a). 4th and 12th fractions I and III of the Regulations of the Mexican Institute of Industrial Property (DOF 14/12/1999, amended on 07/01/2002, 07/15/2004, 07/28/2004 and 07/09/2004 / 2007); Articles 1, 3, 4, 5, section V, subsection a). 16 sections I and III and 30 of the Organic Statute of the Mexican Institute of Industrial Property (DOF 12/27/1999, amended on 10/10/2002, 07/29/2004, 08/04/2004 and 09/13/2007); 1, 3 and 5 subsection a) of the Agreement that delegates powers to the Deputy Directors General, Coordinator, Divisional Directors, Heads of Regional Offices, Divisional Sutxirectors, Departmental Coordinators and other subordinates of the Mexican Institute of Industrial Property. (DOF 12/15/1999, amended on 02/04/2000, 07/29/2004, 08/04/2004 and 09/13/2007).
Issue Date: February 11, 2016
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SIGNAL GENERATION METHOD AND GENERATION DEVICE OF
SIGNALS
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Field of the Invention
The present invention relates to a transmitting device and a receiving device for communication using multiple antennas.
Background of the Invention
A MIMO (Multiple Inputs, Multiple Outputs) system is an example of a conventional communications system that uses multiple antennas. In multi-antenna communication, of which the MIMO system is representative, each of the multiple transmission signals is modulated, and each modulated signal is transmitted at the same time from a different antenna in order to increase the transmission speed of the data.
Figure 23 illustrates an exemplary configuration of a transmit and receive device having two transmit antennas and two receive antennas, and using two modulated transmit signals (transmit currents). In the transmission apparatus, the encoded data is interpolated, the interpolated data is modulated, and frequency conversion and other similar procedures are performed to generate transmission signals, and the transmission signals are transmitted from the antennas. In that case, the method for
Ref. 250007
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simultaneously transmit different signals TR5 <3UiaGas "^ ey úe *" different transmission antennas at the same time and on the same frequency is the MIMO spatial multiplexing system.
In this context, the Patent Bibliography 1 suggests the use of a transmission device provided with a different interpolation pattern for each transmission antenna. That is, the transmission device of figure 23 must use two differentiated interpolation patterns performed by two interpolators (n<sub>to</sub> and nb). Regarding the receiving device, Non-patent Bibliography 1 and Non-patent Bibliography 2 describe how to improve reception quality by iteratively using software values for the detection scheme (by the MIMO detector in figure 2. 3).
As it happens, the models of real propagation environments in wireless communications include the NLOS (non-line-of-sight) models, of which a Rayleigh fading environment is representative, and LOS (con line of sight), of which a Rician fading environment is representative. When the transmitting device transmits a single modulated signal, and the receiving device performs the maximum ratio combination on the signals received by multiple antennas and then demodulates and decodes the resulting signals, excellent reception quality can be achieved in a LOS environment, in particular, in an environment where the Rician factor is large. The Rician factor
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I represents the received power of the direct waves in relation to the received power of the scattered waves. However, depending on the transmission system (for example, a spatial multiplexing MIMO system), a problem occurs in that the reception quality deteriorates when the Rician factor increases (see Non-patent Bibliography 3).
Figures 24A and 24B illustrate an example of the simulation results of the characteristics of the BER (bit error rate) (vertical axis: BER, horizontal axis: SNR (signal to noise ratio) for data encoded with LDPC (Density Based Parity Check) codes and transmitted by a 2x2 spatial multiplexing MIMO system (two transmit antennas, two receive antennas) in a Rayleigh fading environment and in a Rician fading environment with Rician factors of K = 3, 10 and 16 dB. Figure 24A gives the BER characteristics of the logarithmic likelihood ratio based on the Max-Log approximation (Max-log APP) without iterative detection (see Non-patent Bibliography 1 and Non-patent Bibliography 2), while Figure 24B gives the BER Max-Log APP feature with iterative detection (see Non-patent Bibliography 1 and Non-patent Bibliography 2) (number of iterations: five). Figures 24A and 24B clearly indicate that, regardless of whether iterative detection is performed or not, the reception quality is degraded in the spatial multiplexing MIMO system when
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increases the Rician factor. Therefore, the problem of reception quality degradation when the propagation environment stabilizes in the MIMO system of spatial exaction, which does not occur in a conventional single modulation signal system, is unique to the MIMO multiplexing system. space.
Broadcast or multicast communication is a service applied to various propagation environments. The radio wave propagation environment between the broadcast station and the receiving devices belonging to the users is often a LOS environment. When using a spatial multiplexing MIMO system with the above problem for broadcast or multicast communication, a situation may arise where the received power of the electric field in the receiving apparatus is high, but where the quality degradation reception makes it impossible to receive the service. In other words, in order to use a spatial multiplexing MIMO system in broadcast or multicast communication, both in an NLOS environment and in a LOS environment, there is a desire to develop a MIMO transmission system that offers a certain degree of reception quality.
Non-Patent Bibliography 8 describes a method of selecting a codebook for precoding (i.e. a precoding matrix, also called a precoding weight matrix) based on feedback information from a partner of
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communication. However, non-patent bibliography 8 does not describe at all, a method for precoding in an environment where feedback information from the communication partner cannot be acquired, as is the case in the previous broadcast or multicast communication.
Furthermore, the Non-Patent Bibliography 4 describes a method of changing the precoding matrix over time. This scheme is applicable when no feedback information is available. Non-patent Bibliography 4 describes the use of a unit matrix as the matrix for precoding and changing the unit matrix randomly, but does not describe at all a method applicable to reception quality degradation in the LOS environment. previously described. Non-patent Bibliography 4 simply mentions random jumps between precoding matrices. Obviously, the Non-Patent Bibliography 4 makes no mention at all of a precoding method or structure of a precoding matrix, to remedy degradation of reception quality in a LOS environment.
List of mentions
Patent bibliography
Patent bibliography 1
Publication of the International Patent Application
No. W02005 / 050885
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Non-patent bibliography
Non-patent bibliography 1
Achieving near-capacity on a multiple-antenna channel IEEE Transaction on Communications, vol.51, no. 3, pages 389-399, March 2003.
Non-patent bibliography 2
Performance analysis and design optimization of LDPC-coded MIMO OFDM systems IEEE Trans. Signal Processing, vol.52, no.2, pages 348-361, Feb. 2004.
Non-patent bibliography 3
BER performance evaluation in 2x2 MIMO spatial multiplexing systems under Rician fading channels IEICE Trans. Fundamentáis, vol.E91-A, no.10, pages 2798-2807, Oct. 2008.
Non-patent bibliography 4
Turbo space-time codes with time varying linear transformations IEEE Trans. Wireless Communications, vol.6, no.2, pages 486-493, Feb. 2007.
Non-patent bibliography 5
Likelihood function for QR-MLD suitable for soft-decision turbo decoding and its performance IEICE Trans. Commun., Vol.E88-B, no.l, pages 47-57, Jan. 2004.
Non-patent bibliography 6
A tutorial on 'Parallel concatenated (Turbo) coding', 'Turbo (iterative) decoding' and related topics IEICE, Technical
Report IT98-51.
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Non-patent bibliography 7
Advanced signal processing for PLCs: Wavelet-OFDM Proc. of IEEE International symposium on ISPLC 2008, pages
187-192, 2008.
Non-patent bibliography 8
DJ LoveyR. W. HeathJr., Limited feedback unitary precoding for spatial multiplexing Systems IEEE Trans. Inf. Theory, vol.51, no.8, pages 2967-1976, Aug. 2005.
Non-patent bibliography 9
DVB Document A122, Framing structure, channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2), Jun. 2008.
Non-patent bibliography 10
L. Vangelista, N. Benvenuto, and S. Tomasin Key technologies for next-generation terrestrial digital television standard DVB-T2, IEEE Commun. Magazine, vol.47, no.10, pages 146-153, Oct. 2009.
Non-patent bibliography 11
T. Ohgane, T. Nishimura, and Y. Ogawa, Application of space division multiplexing and those performance in a MIMO channel IEICE Trans. Commun., Vo. 88-B, no.5, pages 1843-1851, May 2005.
Non-patent bibliography 12
RG Gallager Low-density parity-check codes, IRE Trans. Inform. Theory, IT-8, pages 21-28, 1962.
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; P / O kj / -. L 'Oí
Non-patent bibliography 13
DJC Mackay, Good error-correcting codes based on very sparse matrices, IEEE Trans. Inform. T-héory, vol. 4 5, no. 2, pages 399-431, March 1999.
Non-patent bibliography 14 ETSI EN 302 307, Second generation framing structure, channel coding and modulation systems for broadcasting, Interactive Services, news gathering and other broadband satellite applications v.1.1.2, Jun. 2006.
Non-patent bibliography 15
Y.-L. Ueng, and C.-C. Cheng A fast-convergence decoding method and memory-efficient VLSI decoder architecture for irregular LDPC codes in the IEEE 802.16e standards IEEE VTC-2007 Fall, pages 1255-1259.
Non-patent bibliography 16
SM Alamouti Asimple transmit diversity technique for wireless Communications IEEE J. Select. Commun areas. , vol. 16, no.8, pages 1451-1458, Oct 1998.
Non-patent bibliography 17
V. Tarokh, H. Jafrkhani, and AR Calderbank Space-time block coding for wireless Communications: Performance results IEEE J. Select. Areas Commun., Vol. 17, no. 3, no.3, pages 451-460, March 1999.
Brief Description of the Invention
Technical problem
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It is an objective of the present invention to provide a MIMO system that improves reception quality in an environment
THE.
Solution to the problem
The present invention provides a signal generation scheme for generating, from multiple baseband signals, multiple signals for transmission in a common frequency band and at a common date-time, comprising the steps of: carrying out a phase change in each of a first baseband signal if generated from a first set of bits and a second baseband signal s2 generated from a second set of bits, thus generating a first baseband signal with postphase change if 'and a second baseband signal with postphase change s2'; and applying the weighting to the first baseband signal with postphase change if 'and to the second baseband signal with postphase shift s2' according to a predetermined matrix F, thus generating a first zl-weighted signal and a second z2-weighted signal as the plurality of signals for transmission in the common frequency band and in the common date-time, where the first zl-weighted signal and the second z2-weighted signal meet the relationship: (zl, z2)<sup>T </sup>= F (sl ', s2')<sup>T</sup> and the phase change is performed on the first baseband signal si and the second baseband signal s2 using a phase modification value sequentially selected from
N candidates of phase modification value, N being an integer
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, A PkGHEDAD INDUSTRIAL
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equal to or greater than two and each of the N candidates for phase modification value being selected at least once within a predetermined period.
Furthermore, the present invention provides a signal generating apparatus for generating, from multiple baseband signals, multiple signals for transmission in a common frequency band and at a common date-time, comprising a phase changer carrying performing a phase change on each of a first baseband signal if generated from a first set of bits and a second baseband signal s2 generated from a second set of bits, thus generating a first baseband signal with postphase change if 'and a second baseband signal with postphase change s2'; and a weighting unit that weights the first baseband signal with postphase change if 'and the second baseband signal with postphase shift s2' according to a predetermined matrix F, thereby generating a first signal weighted zl and a second z2-weighted signal as the plurality of signals for transmission in the common frequency band and in the common date-time, where the first zl-weighted signal and the second z2-weighted signal meet the relationship: (zl, z2)<sup>T</sup> = F (sl ', s2')<sup>T</sup> and the phase change is performed on the first baseband signal si and the second baseband signal s2 using a phase modification value sequentially selected from N candidates for phase modification value, N being an integer equal to or greater than two and each of the N fcíAjJt.
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instituto mex; cy ¿of industrial PROPERTY
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phase modification value candidates being selected at least once within a predetermined period.
Suitable effects of the invention
In accordance with the foregoing structure, the present invention provides a signal generation scheme and signal generation apparatus that remedies degradation of reception quality in a LOS environment, thereby providing high quality service to LOS users during communication by broadcast or multicast.
Brief Description of the Figures
Figure 1 illustrates an example of a transmit and receive device in a MIMO spatial multiplexing system.
Figure 2 illustrates an exemplary frame configuration.
Figure 3 illustrates an example of a transmission device that applies a phase change scheme.
Figure 4 illustrates another example of a transmission device applying a phase change scheme.
Figure 5 illustrates another exemplary plot configuration.
Figure 6 illustrates an exemplary phase change scheme.
Figure 7 illustrates an exemplary configuration 25 of a receiving device.
MEXICAN INSTITUTE
OF THE PROPERTY
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<img file="MX337079B_D0019.tif" />
Figure 8 illustrates an exemplary configuration of a signal processor of the receiving device.
Figure 9 illustrates another exemplary configuration of a signal processor of the receiving device.
Figure 10 illustrates an iterative decoding scheme.
Figure 11 illustrates eg reception conditions.
Figure 12 illustrates a further example of a transmission device applying a phase change scheme.
Figure 13 illustrates yet another example of a transmission device applying a phase change scheme.
Figures 14A and 14B illustrate a further exemplary screen configuration.
Figures 15A and 15B illustrate yet another exemplary screen configuration.
Figures 16A and 16B illustrate yet another exemplary screen configuration.
Figures 17A and 17 B illustrate yet another exemplary screen configuration.
Figures 18A and 18B even illustrate a further exemplary screen configuration.
Figures 19A and 19B illustrate examples of a schematic
<img file="MX337079B_D0020.tif" />
correlation. ......................—.
Figures 20A and 20B illustrate further examples of a correlation scheme.
Figure 21 illustrates an exemplary configuration of a weighting unit.
Figure 22 illustrates an exemplary symbol rearrangement scheme.
Figure 23 illustrates another example of a transmit and receive device in a spatial multiplexing MIMO system.
Figures 24A and 24B illustrate BER eg emplificative characteristics.
Figure 25 illustrates another example phase shift scheme.
Figure 26 illustrates yet another exemplary phase shift scheme.
Figure 27 illustrates a further exemplary phase shift scheme.
Figure 28 illustrates yet another exemplary phase shift scheme.
Figure 29 illustrates yet a further exemplary phase shift scheme.
Figure 30 illustrates an exemplary symbol arrangement of a modulated signal that provides a high quality of received signal.
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Figure 31 illustrates a cnnf i gni «wn.i¿Mn — ¿3 ^ .. exemplary frame for a modulated signal that provides a high quality of received signal.
Figure 32 illustrates another exemplary symbol arrangement for a modulated signal that provides a high quality of received signal.
Figure 33 illustrates yet another exemplary symbol arrangement for a modulated signal that provides high quality of received signal.
Figure 34 illustrates the variation in amounts of symbols and ranges required per coded block, when using block codes.
Figure 35 illustrates the variation in amounts of symbols and ranges required per coded block pair, when using block codes.
Figure 36 illustrates a global configuration of a digital broadcast system.
Figure 37 is a block diagram illustrating an exemplary receiver.
Figure 38 illustrates the multiplexed data configuration.
FIG. 39 is a schematic diagram illustrating multiplexing of current encoded data.
Figure 40 is a detailed diagram illustrating a video stream as contained in a sequence of PES packets.
<img file="MX337079B_D0022.tif" />
Figure 41 is a structural diagram of ..... pagu? Rer ~ —— -
TS and source packets of multiplexed data.
Figure 42 illustrates the PMT data configuration.
Figure 43 illustrates the information as configured in the multiplexed data.
Figure 44 illustrates the configuration of the current attribute information.
Figure 45 illustrates the configuration of a video display and audio output device.
Figure 46 illustrates an exemplary configuration of a communications system.
Figures 47A and 47B illustrate a variant of the exemplary symbol arrangement for a modulated signal that provides high quality of received signal.
Figures 48A and 48B illustrate another variant of the exemplary symbol arrangement for a modulated signal that provides high quality of received signal.
Figures 49A and 49B illustrate yet another variant of the exemplary symbol arrangement for a modulated signal that provides high quality of received signal.
Figures 50A and 50B illustrate a further variant of the exemplary symbol arrangement for a modulated signal which provides a high quality of received signal.
Figure 51 illustrates an exemplary configuration of a transmission device.
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Figure 52 illustrates another exemplary configuration of a transmission device.
Figure 53 illustrates a further exemplary configuration of a transmission device.
Figure 54 illustrates yet another exemplary configuration of a transmission device.
Figure 55 illustrates a baseband signal changer.
Detailed description of the invention
The embodiments of the present invention are described below with reference to the accompanying figures.
Mode 1
The following describes in detail a transmission scheme, a transmission device, a reception scheme and a reception device relevant to the present embodiment.
Before beginning the description itself, an outline of the transmission schemes and decoding schemes is provided in a conventional spatial multiplexing MIMO system. Figure 1 illustrates the structure of a N spatial multiplexing MIMO system.<sub>t</sub>xN<sub>r</sub>. An information vector z is encoded and interpolated. The vector of encoded bits u = (u<sub>lf</sub> ... or<sub>Nt</sub>) is obtained as the interpolation output. Here, u ± = (un, ... Ui<sub>M</sub>) (where M is the number of bits transmitted per symbol). In the case of a vector of
<img file="MX337079B_D0024.tif" />
for the #i transmit antenna. Normalizing the transmission energy, this can be expressed as E {| Yes |<sup>2</sup>} = E<sub>s</sub>/ N<sub>t</sub> (where E<sub>s</sub> is the total energy per channel). The reception vector y = (yi, - YNr)<sup>T</sup> it is expressed in Mathematics 1 (formula 1), below.
[Mathematics 1] (formula 1) y = (vi / · <sup>—</sup> HjW <sup>S + 11</sup>
Here, H<sub>NtNr</sub> is the channel matrix, n = (n<sub>x</sub>, ... n<sub>Nr</sub>) is the noise vector and the mean value of ni is zero for independently complex and identically distributed Gaussian noise (iid) of variance or<sup>2</sup>. Based on the relationship between the transmitted symbols entered into a receiver and the received symbols, the probability distribution of the received vectors can be expressed as Mathematics 2 (formula 2), below, for a multidimensional Gaussian distribution.
[Mathematics 2] (formula 2)
Here, we consider a receiver that performs iterative decoding. Such a receiver is illustrated in figure
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IMPI <sup>1NST,</sup>MEXICAN TUTO OF THE INDUSTRIAL PROPERTY
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l as constituted of an input / output decoder of<sup>1 </sup>external software and a MIMO detector. The logarithmic likelihood ratio vector (L value) of Figure 1 is given by Mathematics 3 (formula 3) to Mathematics 5 (formula 5), as follows.
[Mathematics 3] (formula 3) z (u) = (Zz (<sub>Wl</sub>), · · -, L {u <sub>Nt</sub>) J [Mathematics 4] (formula 4) <sup>L</sup>(u) = [Mathematics 5] (formula 5)
- l<sup>n</sup> = +1) p ^ j = - i) (Iterative detection scheme)
The following describes the iterative detection of MIMO signals performed by the N spatial multiplexing MIMO system.<sub>t</sub>xN<sub>r</sub>.
The logarithmic likelihood ratio of Um is defined by Mathematics 6 (formula 6).
[Mathematics 6] (formula 6) <sup>P (</sup>or<sub>m</sub>„ = <sup>+1</sup>one and) <sup>p</sup>(or<sub>mn</sub> = -<sup>1</sup>one and)
I institute
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<sup>£</sup>(wjy) =<sup>z</sup>«
Through the application of Bayes' theorem, Mathematics 6 (formula 6) can be expressed as Mathematics 7 (formula 7).
[Mathematics 7] (formula 7) ¿(<sub>w</sub>Jy) = z »= ln = In p (y \ u<sub>m</sub><sup>=+</sup>^<sup>p</sup>(or<sub>m</sub>„=+<sup>i</sup>yp (y '>
<sup>p</sup>(z / TO<sup>=+1</sup>) | ; ./ <and iM<sub>mn</sub><sup>=+1</sup>) p (yln<sub>m</sub>„= -I) p (u = + i) Σγζ p (y IW · I uj Σγζ p (y I») p (»I« J
U mn, - \
Note that Umn, <sub>±</sub>i = {u | u<sub>mn</sub> = ± l} · Using the InDaj ~ max ln aj approximation, Mathematics 7 (formula 7) can be approximated as Mathematics 8 (formula 8). The symbol ~ is used here to mean approximation.
Τ '!
_____A? ', [Mathematics 8] (formula 8) du „„ | y) »ln Ph<sup>, lm</sup> i<sup>1</sup>) + max {ln p (y | u) + P (u | «„ „)}
- max {ln p (y | u) + P (u I Umn)}
Umn, - \
In Mathematics 8 (formulas), P (u | u<sub>mn</sub>) ylnP (u | u<sub>mn</sub>) can be expressed as follows.
[Mathematics 9] (formula 9) <sup>p</sup>^ \ uJ = n<sup>p</sup>w (l »(» JH) exp = Π
0 »(rum)
<td>exp</td><td>i l <sup>2</sup> J</td><td>+ exp</td><td>F <sup>£</sup>(μ? Ί ί <sup>2</sup> J</td>
[Mathematics 10] (formula 10) <sup>lnP</sup>^ \ uJ ^ lnP (u,) -¡Pi-llJ <and J [Mathematics 11] (formula 11)
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lnP (uij) = -UijP (uy) -ln \ exp ^ + exp \ iHi / W-ylwl M> 2 ¿(// y)
Note that the logarithmic probability of the equation given in Mathematics 2 (formula 2) can be expressed as Mathematics 12 (formula 12).
[Mathematics 12] (formula 12)
Z «P (y | u) =
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y-Hs (u) | f
Therefore, given Mathematics 7 (formula 7) and Mathematics 13 (formula 13), the posterior L value for MAP or APP (posterior probability) can be expressed as follows.
[Mathematics 13] (formula 13)
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• '!
L (uJy) = ln
<td>Συ <sup>βχ</sup>ρ · LV mn, + \</td><td>one 2σ</td><td>y-Hs (u) | f + L<sup>Zni,</sup>U) J</td>
<td>Συ <sup>εχ</sup>ρ · mn, -l</td><td>• one . 2σ</td><td>y-hs (u)<sup>2</sup>+ E / «p (<sub>w</sub>j!</td>
This is hereafter referred to as iterative APP decoding. Furthermore, given Mathematics 8 (formula 8) and Mathematics 12 (formula 12), the posterior L value for the Max-log APP can be expressed as follows.
[Mathematics 14] (formula 14) <sup>Li</sup>~ u<sub>m</sub>„I y) ® max {T (u, y, £ (u))} - max {T (u, y, L (u))}
Umn, + l Umn, -l
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[Mathematics 15] (formula 15) u, y, I (u)) = - i-, ly — Hs (ll) || + Σ<sup>Ζ</sup><sup>Ρ </sup>2σ <i
This is hereafter called iterative Max-log APP decoding. In itself, the external information required by the iterative decoding system can be obtained by subtracting the previous entry from Mathematics 13 (formula 13) or Mathematics 14 (formula 14).
Ij
1.
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System Model ——— Figure 23 illustrates the basic configuration of a system related to the following explanations. The illustrated system is a 2x2 spatial multiplexing MIMO system that has an external decoder for each of two streams A and B. The two external decoders perform identical LDPC decoding (although this description considers the example of a configuration in which external encoders use LDPC codes, external encoders are not restricted to using LDPC as error correction codes). The example can also be done using other error correction codes, such as turbo codes, convolutional codes, or convolutional LDPC codes. Also, while external encoders are now described as individually configured for each transmit antenna, no limitation is provided in that regard. A single external encoder can be used for multiple transmit antennas, or the number of external encoders may be greater than the number of transmit antennas. The system also has interpolators (n<sub>to</sub>, nb) for each of currents A and B. Here, the modulation scheme is 2<sup>h</sup>-QAM (that is, h bits are transmitted per symbol).
The receiver performs iterative detection (iterative APP (or Max-log APP) decoding) of MIMO signals, as already described. LDPC codes are decoded using,
<img file="MX337079B_D0034.tif" />
for example, sum-product decoding. '
Figure 2 illustrates the frame configuration and describes the order of symbols after interpolation. Here, (i<sub>to</sub>, j<sub>to</sub>) and (ih / Jb) can be expressed as follows.
[Mathematics 16] (formula 16) ^ the 'J π a ^ a ^ LiaJa) [Mathematics 17] (formula 17) ^ lb' j ~~ 7tb (Qib, jl)
Here i<sub>to</sub> and ib represent the order of symbols after interpolation, j<sub>to</sub> y represent the bit position in the modulation scheme (where j<sub>to</sub>, jb = 1, ... h), n<sub>to</sub> y represent the interpolators of currents A and B, and Ω ia, ja and Ω lb, jb represent the order of the data of currents A and B before interpolation. Note that Figure 2 illustrates a situation where i<sub>to</sub> = ib ·
Iterative decoding
The following describes in detail the sum-product decoding used to decode the LDPC codes and the iterative detection of the MIMO signal algorithm, both used by the receiver.
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Sum-product decoding
A two-dimensional MxN matrix H = {Hmn} is used as the verification matrix of LDPC codes subject to decoding. In the case of the set [1, N] = {l, 2 ... N}, the partial sets A (m) and B (n) are defined as follows.
[Mathematics 18] (formula 18)
A (m) = {n: = 1} [Mathematics 19] (formula 19)
W <sup>Ξ</sup> : H<sub>mn</sub> = Π
Here, A (m) means the set of column indices equal to 1 for row m of the check matrix H, while B (n) means the set of row indices equal to 1 for row n of the check matrix H. The algorithm of the sum-product decoding is as follows.
Step Al (Initialization): in all the pairs (m, n) that meet Η, ^ = 1, the previous logarithmic ratio β, ηη = 1 is established. The loop variable (number of iterations) lsum = 1 is established, and the maximum number of loops is set l<sub>ITS</sub>tn, inax
Step A-2 (Processing): In all pairs (m, n) that meet Hmn = 1 in the order m = 1, 2, ... M, the logarithmic ratio of extrinsic value is updated to<sub>mn</sub> using the following
<img file="MX337079B_D0036.tif" />
upgrade formula.
[Mathematics 20] (formula 20) a,
Π '«' Ό, + ΛΑ * f Σ \ ríeA (m) \ n [Mathematics 21] (formula 21) sign (x) = x> 0 x <0 [Mathematics 22] (formula 22) / (x) Ξ ln exp (x) +1 exp (x) -1 where f is the Gallager function. λ<sub>η</sub> it can then be computed as follows.
Step A-3 (Column operations): in all the pairs (m, n) that meet = l in the order n = 1, 2, ... N, the logarithmic ratio of extrinsic value α, πη is updated using the following update formula.
[Mathematics 23] (formula 23)
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m'n
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Step A-4 (Calculation of ratio
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logarithmic): for ne [1, N], the logarithmic likelihood ratio L<sub>n</sub> it is computed as follows.
[Mathematics 24] (formula 24)
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Step A-5 (Iteration count): if l<sub>sum</sub> <1, then increase l<sub>sum</sub> and the process returns to step A-2. The sum-product decoding ends when l<sub>sum</sub> = l<sub>S</sub>um, max
The above describes an iteration of the sum-product decoding operations. Subsequently, iterative detection of MIMO signals is performed. The variables m, n, amn, βπ, η, λ<sub>η</sub>, and L<sub>n</sub> used in the preceding explanation of sum-product decoding operations are expressed as m<sub>to</sub>, na, a<sup>to</sup>m<sub>Ana</sub>.
P<sup>to</sup>mana Ana and Lh for current A and as m<sub>b</sub>, n<sub>b</sub>, to<sup>b</sup>mbnb, p<sup>b</sup>mbnbz and Lnh for stream B.
Iterative detection of MIMO signals The following describes the calculation of λ<sub>η</sub> for iterative detection of MIMO signals.
The following formula can be derived from Mathematics (formula 1).
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<img file="MX337079B_D0043.tif" />
[Mathematics 25] (formula 25) y «= (y = H<sub>!2</sub>«S (') + nW
Given the plot configuration illustrated in Figure 2, the following functions are derivable from Mathematics 16 (formula 16) and Mathematics 17 (formula 17).
[Mathematics 26] (formula 26) a
ia ha
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[Mathematics 27] (formula 27)
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<img file="MX337079B_D0046.tif" />
b ib, jb where n<sub>to</sub>, n<sub>b</sub>e [1, N]. For the iteration Je of the iterative detection of MIMO signals, the variables Ána, L<sub>na</sub>/ and Lnb are expressed as Á<sub>k> na</sub>, L<sub>k> na</sub>, TO<sub>k (nb</sub> and L<sub>k (nb</sub>.
Step Bl (Initial detection; k = 0)
For initial wave detection, Á<sub>or</sub>,<sub>n</sub>ay λ<sub>0 (Ι</sub>* are calculated as follows.
For iterative APP decoding:
[Mathematics 28]
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(formula 28) || y (zjf) “H22 (/ jr)<sup>s</sup>(<sup>or</sup>(/ A)) ||
Σ ^ θ || y (z>) - H220>) s («(z>)) ||<sup>!</sup>
For Iterative Max-log APP decoding: [Mathematics 29] (formula 29) / Lo, „=“<sup>ax</sup> M “(z \)> y (z /) F max {<sup>ψ</sup>(“(Ζ '^)> Υ (ζΛ ·))} <sup>X</sup> t? 0,<sub>n</sub>y, + l U 0
0, ηχ, -1 [Mathematics 30] (formula 30) <sup>ψ</sup>(<sup>μ</sup>(ζ \) 'Υ (ζ<sub>ύ</sub>)) = “7 ^ || y (z \ v)“ H22 (zjf)<sup>s</sup>(<sup>or</sup>G'x)) || where X = a, b. Then the count of iterations for iterative detection of MIMO signals is set to l<sub>mime</sub> = 0, with the maximum count of iterations being minimum, max ·
Step B-2 (Iterative Detection; Iteration k): When the iteration count is k, Math 11 (formula 11), the
Mathematics 13 (formula 13) to Mathematics 15 (formula 15), Mathematics 16 (formula 16) and Mathematics 17 (formula 17) can be expressed as Mathematics 31 (formula 31) to Mathematics 34 (formula 34) below. Note that (X, Y) = (a, b) (b, a).
<img file="MX337079B_D0048.tif" />
<img file="MX337079B_D0049.tif" />
For decoding APP iteratiVST [Mathematics 31] (formula 31)
Σττ <sup>former</sup>P <
^<sup>k</sup>-n<sub>x</sub> ^*-<sup>1</sup>· Ωί ·., Ν<sup>(Μ</sup>Ωχ.Α<sup>) + 1η</sup>'
-IA.,<sub>V</sub>
Στ / <sup>former</sup>P '(- / *<sub>ΛΛ</sub>·.-Ι ^ / 1 ^ 0 ^) - 1122 (/ 2: ^^ 0 ^)) 11 <sup>+ p <</sup>Uq - ..> '“/ || ϊθ%) - H22 (|>) s (uO>)) ||<sup>,+</sup> f ^ Uc¿} [Mathematics 32] (formula 32) ¡X.JX <sub>r</sub>= go * jX + Σ y = l
For Iterative Max-log APP decoding: [Math 33] (formula 33) λ *. „<sub>χ</sub> ) + max U. „<sub>x</sub><sup>Χ</sup>.,{<sup>Ψ</sup>ί<sup>U (/ x)</sup>’ <sup>Υ (Ζ</sup>'<sup>χ)</sup>’ <sup>P (w</sup>í £?)} “{<sup>Ψ</sup>(<sup>(/ χ)</sup>’<sup>and(/%)</sup>'[Mathematics 34] (formula 34)
y (ú) -H22G>) s («0T)) |<sup>2+</sup>/ '(w £ ¿j
Step B-3 (Iteration count and estimation of
<img file="MX337079B_D0050.tif" />
<img file="MX337079B_D0051.tif" />
Dt ídi I xword code) If l<sub>mime</sub> <minimum, maxz then increases l<sub>minium</sub> and the process returns to step B-2. When I<sub>m</sub>i<sub>mo</sub> = minimum, maxz finds an estimated code word, as follows.
[Mathematics 35] (formula 35) <sup>OR</sup>n<sub>x</sub>
<img file="MX337079B_D0052.tif" />
> 0 <0 where X = a, b.
Figure 3 shows an exemplary exemplary configuration of a transmission device 300 relevant to the present embodiment. An encoder 302A takes the information (data) 301A and a frame configuration signal 313 as input (which includes the error correction scheme, the encoding rate, the block length and other information used by the encoder 302A in the data error correction encoding, so that the scheme designated by frame configuration signal 313 is used. You can change the error correction scheme.) In accordance with frame configuration signal 313, encoder 302A performs error correction encoding, such as convolutional encoding, LDPC encoding, turbo encoding, or the like, and outputs encoded data 303A.
An interpolator 3 04A takes the encoded data 3 03A and frame configuration signal 313 as input, performs
<img file="MX337079B_D0053.tif" />
the interpolation, that is, it rearranges their order and then outputs the interpolated data 305A. (Depending on the frame setting signal 313, you can change the interpolation scheme.)
A correlator 306A takes the interpolated data 305A and the raster setup signal 313 as input and performs modulation on them, such as QPSK (Quadrature Phase Shift Modulation), 16-QAM. 16) Quadrature Amplitude Modulation or 64-QAM (64 State Quadrature Amplitude Modulation); it then outputs a 307A baseband signal. (Depending on the frame setting signal 313, you can change the modulation scheme.)
Figures 19A and 19B illustrate an example of a QPSK modulation mapping scheme for a baseband signal consisting of a phase I component and a quadrature Q component in the IQ plane. For example, as shown in Figure 19A, when the input data is 00, the output is I = 1.0, Q = 1.0. Similarly, when the input data is 01, the output is I = -1.0, Q = 1.0, etc. Figure 19B illustrates an example of a QPSK modulation mapping scheme in the IQ plane that differs from Figure 19A in that the signal points of Figure 19A have rotated around the origin to obtain the signal points of Figure 19B. The Non-patent Bibliography 9 and the Non-patent Bibliography 10 describe such a scheme of
<img file="MX337079B_D0054.tif" />
constellation rotation. As an alternative, caffiBl ^ fr ^ -pued »adopt the cyclical Q delay described in Non-patent Bibliography 9 and Non-patent Bibliography 10. An alternative example, other than Figures 19A and 19B, is shown in Figures 20A and 20B, illustrating a signal point design for 16-QAM in the IQ plane. The example in figure 20A corresponds to figure 19A, while the one in figure 20B corresponds to figure 19B.
An encoder 302B takes information (data) 301B and frame configuration signal 313 as input (which includes the error correction scheme, encoding rate, block length, and other information used by encoder 302A in encoding. data error correction scheme, so that the designated scheme is used by frame configuration signal 313. You can change the error correction scheme). In accordance with frame configuration signal 313, encoder 302B performs error correction encoding, such as convolutional encoding, LDPC encoding, turbo encoding, or the like, and outputs encoded data 303B.
An interpolator 304B takes the encoded data 303B and the frame configuration signal 313 as input, performs the interpolation, that is, rearranges the order thereof, and outputs the interpolated data 305B. (According to frame setting signal 313, you can change the scheme of
<img file="MX337079B_D0055.tif" />
interpolation).
A correlator 306B takes the interpolated data 305B and the frame configuration signal 313 as input and performs modulation on them, such as QPSK, 16-QAM or 64-QAM, then outputs a baseband signal 307B. (Depending on the frame setting signal 313, you can change the modulation scheme.)
An information generator for signal processing schemes 314 takes frame configuration signal 313 as input, and therefore outputs processing scheme information 315. Signal processing scheme information 315 designates the matrix of Fixed precoding to be used and includes information about the phase change pattern used to change the phase.
A weighting unit 30 8A takes the baseband signal 307A, the baseband signal 307B, and the signal processing scheme information 315 as input and, according to the signal processing scheme information 315, performs the weighting on baseband signals 307A and 307B, then outputs a weighted signal 309A. The weighting scheme is described in detail below.
A wireless unit 310A takes the 3 09A weighted signal as input and performs processing, such as quadrature modulation, band limiting, frequency conversion, amplification, etc., then outputs the transmit signal 311A. A 312A antenna then outputs the signal
<img file="MX337079B_D0056.tif" />
MEXICAN INSTITUTE OF PROPERTY
INDUSTRIAL «Μ
<img file="MX337079B_D0057.tif" />
transmission 311A as radio waves. _____
A weighting unit 3 08B takes the baseband signal 3 07A, the baseband signal 3 07B, and the signal processing scheme information 315 as input and, according to the signal processing scheme information 315, performs the weighting on the baseband signals 307A and 307B, then outputs the weighted signal 316B.
Figure 21 illustrates the configuration of the weighting units 3 08A and 308B. The area of figure 21 surrounded by dashed line represents one of the weighting units. The baseband signal 307A is multiplied by wll to obtain wll-sl (t) and multiplied by w21 to obtain w21 · si (t). Similarly, the baseband signal 3 07B is multiplied by wl2 to obtain wl2 · s2 (t) and multiplied by w22 to obtain w22 · s2 (t). Then zl (t) = wll-sl (t) + wl2-s2 (t) and z2 (t) = w21-sl (t) + w22 · s22 (t) are obtained. Here, as already explained, sl (t) and s2 (t) are the modulated baseband signals according to a modulation scheme such as BPSK (binary phase shift modulation), QPSK, 8- PSK (8-symbol phase shift modulation), 16-QAM, 32-QAM (32-channel quadrature amplitude modulation), 64-QAM, 256-QAM 16-APSK (phase shift modulation) 16 symbols), etc.
Both weighting units perform weighting using a fixed precoding matrix. The matrix of
<img file="MX337079B_D0058.tif" />
Precoding uses, for example, the scheme of Mathematics 36 (formula 36) and fulfills the conditions of Mathematics 37 (formula 37) or Mathematics 38 (formula 38), all of which are set out below. However, this is only an example. The value of oí is not restricted to Mathematics 37 (formula 37) and Mathematics 38 (formula 38) and can assume other values, for example, a = 1. Here, the precoding matrix is:
[Mathematics 36] (formula 36) vt / 22 ^
<img file="MX337079B_D0059.tif" />
e jo ax jx e J
In Mathematics 36 (formula 36) above, a can be given by:
[Mathematics 37] (formula 37) and [2 + 4
J2 + 2
As an alternative, in Mathematics 36 (formula 36) above, a can be given by:
[Mathematics 38] (formula 38)
V2 + 3 + V5 a / 2 + 3-75
The precoding matrix is not restricted to that of
<img file="MX337079B_D0060.tif" />
Mathematics 36 (formula 36), it can also be as indicated by Mathematics 39 (formula 39).
[Mathematics 39] '(formula 39) wll w! 2A (ab \ w21 w22
In Mathematics 39 (formula 39), let a = Ae<sup>j511</sup>, b = Be<sup>jS12</sup>, c = Ce<sup>j521</sup> yd = From<sup>j622</sup> . Also, one of a, b, c and d can be zero. For example, the following configurations are possible: (1) a can be zero while b, c, and d are non-zero,, (2) b can be zero while a, c, and d are non-zero, (3 ) c can be zero as long as a, b, and d are non-zero, or (4) d can be zero as long as a, b, and c are non-zero.
When any of the modulation scheme, the error correction codes and the encoding rate thereof are changed, the precoding matrix can also be set, changed and set for use.
A phase shifter 317B takes the weighted signal 316B and signal processing scheme information 315 as input, then regularly changes the phase of the signal 316B to output it. This regular change is a phase change made according to a predetermined phase change pattern that has a predetermined period (cycle) (for example, all symbols n (where n is an integer, n 1) or at a predetermined interval) .
<img file="MX337079B_D0061.tif" />
Details of the phase change pattern oe are explained below in Mode 4.
The 310B wireless unit takes the 3B postphase shift signal as input and performs processing such as quadrature modulation, band limiting, frequency conversion, amplification, etc. , then outputs the transmit signal 311B. Antenna 312B then outputs transmission signal 311B as radio waves.
FIG. 4 illustrates an exemplary exemplary configuration of a transmission device 400 that differs from that of FIG.
3. Next, the points of difference of figure 4 with respect to figure 3 are described.
An encoder 402 takes the information (data) 401 and the frame configuration signal 313 as input, and, according to the frame configuration signal 313, performs the error correction encoding and outputs the encoded data 402.
A distributor 404 takes the encoded data 403 as input, performs the distribution thereof, and outputs data 405A and data 405B. Although Figure 4 illustrates only one encoder, the number of encoders is not limited thereto. The present invention can also be carried out using m encoders (m being an integer, m 1) such that the distributor divides the encoded data created by each encoder into two groups for distribution.
Figure 5 illustrates an example of a configuration
<img file="MX337079B_D0062.tif" />
frame in the time domain for a transmission device according to the present embodiment. The symbol 500_l is to notify the receiving device about the transmission scheme. For example, symbol 500_l transmits information such as the error correction scheme used to transmit the data symbols, the encoding rate thereof, and the modulation scheme used to transmit the data symbols.
The symbol 501_l is for estimating the channel fluctuations of the modulated signal zl (t) (where t is the time) transmitted by the transmission device. The symbol 502_l is a data symbol transmitted by the modulated signal zl (t) as symbol number u (in the time domain). The symbol 503_l is a data symbol transmitted by the modulated signal zl (t) as the symbol number u + 1.
Symbol 501_2 is for estimating the channel fluctuations of the modulated signal z2 (t) (where t is time) transmitted by the transmission device. The symbol 502_2 is a data symbol transmitted by the modulated signal z2 (t) as the symbol number u (in the time domain). The symbol 503_2 is a data symbol transmitted by the modulated signal zl (t) as the symbol number u + 1.
Here, zl (t) and z2 (t) symbols that have the same date-time (identical timing) are transmitted from the transmitting antenna using the same frequency
<img file="MX337079B_D0063.tif" />
(common / shared).
The following describes the relationships between the modulated signals zl (t) and z2 (t) transmitted by the transmitting device and the received signals rl (t) and r2 (t) received by the receiving device.
In figure 5, 504 # l and 504 # 2 indicate the transmitting antennas of the transmitting device, while 505 # l and 505 # 2 indicate the receiving antennas of the receiving device. The transmitting device transmits the modulated signal zl (t) from transmitting antenna 504 # l and transmits the modulated signal z2 (t) from transmitting antenna 504 # 2. Here, it is assumed that the modulated signals zl (t) and z2 (t) occupy the same frequency (common / shared) (bandwidth). The channel fluctuations of the transmitting antenna of the transmitting device and the antennas of the receiving device are hn (t), hi<sub>2</sub>(t), h<sub>2</sub>i (t) and h<sub>22</sub>(t), respectively. Assuming that the receiving antenna 505 # l of the receiving device receives the received signal rl (t) and that the receiving antenna 505 # 2 of the receiving device receives the received signal r2 (t), the following relationship is maintained.
[Mathematics 40] (formula 40) <sub>=</sub>f / ζ, Λ) AjOY
<img file="MX337079B_D0064.tif" />
Figure 6 belongs to the scheme ^ dé ”^ bnderaeióft == - ™ ~». (precoding scheme) and the phase change scheme of the present modality. A 600 weighting unit is a combined version of the 3 08A and 3 08B weighting units in Figure
3. As shown, current sl (t) and current s2 (t) correspond to baseband signals 3 07A and 3 07B in the figure
з. That is, the currents sl (t) and s2 (t) are the baseband signals consisting of a phase I component and a quadrature Q component in accordance with the correlation carried out by a modulation scheme such as QPSK, 16- QAM and 64-QAM. As indicated by the frame configuration in Figure 6, the current sl (t) is represented as if (u) in the symbol number u, as sl (u + l) in the symbol number u + 1, etc. . Similarly, the current s2 (t) is represented as s2 (u) in symbol number и, as s2 (u + 1) in symbol number u + 1, etc. The weighting unit 600 takes the baseband signals 307A (si (t)) and 307B (s2 (t)) as well as the signal processing scheme information 315 of FIG. 3 as input, performs the weighting according with the signal processing scheme information 315, and outputs the weighted signals 309A (zl (t)) and 316B (z2 '(t)) of FIG. 3. The phase changer 317B changes the weighted signal phase 3l6B (z2 '(t)) and outputs the postphase shift signal 309B (z2 (t)).
Here, given the vector W1 = (wll, wl2) of the first row of the fixed precoding matrix F, zl (t) can be expressed
<img file="MX337079B_D0065.tif" />
like Math 41 (formula 41) [Mathematics 41] (formula 41) below.
zl (t) = Wlx (sl (0, s2 (0)<sup>T</sup>
Similarly, given the vector W2 = (w21, w22) of the second row of the fixed precoding matrix F, and letting the phase change formula applied by the phase changer be y (t), z2 (t ) can be expressed as Mathematics 42 (formula 42) below.
[Mathematics 42] „(formula 42) z2 (t) = y (t) x ΡΓ2 x (λΊ (/), s2 (í))<sup>t</sup>
Here, y (t) is a phase change formula that follows a predetermined scheme. For example, given a period (cycle) of four and date-time u, the phase change formula can be expressed as Math 43 (formula 43) below.
[Mathematics 43] (formula 43) and (u) = e<sup>J</sup>°
Similarly, the phase change formula for datetime u + 1 can be, for example, as given by Mathematics 44 (formula 44).
<img file="MX337079B_D0066.tif" />
<img file="MX337079B_D0067.tif" />
[Mathematics 44] (formula 44) and (u + i) = e<sup>2</sup>
That is, the phase change formula for the date-time u + k can be expressed as Mathematics 45 (formula
45) [Mathematics 45] (formula 45) and (u + fc) = e
<img file="MX337079B_D0068.tif" />
Note that Mathematics 43 (formula 43) through Mathematics 45 (formula 45) are given only as an example of a regular phase change.
Regular phase change is not restricted to a period (cycle) of four. Potentially, enhanced receive capabilities (error correction capabilities, to be exact) can be promoted in the receive device by increasing the amount of the period (cycle) (this does not mean that a longer period (cycle) is better, although avoiding low amounts such as two is probably ideal.)
Also, although the preceding Mathematics 43 (formula 43) to Mathematics 45 (formula 45) represent a configuration in which a phase change is carried out through rotation by predetermined consecutive phases (in the uaatvuí · ..: ..
¡72:, Á í, A; r7í: í.> A.7>
NLívSTKIaL
PT
<img file="MX337079B_D0069.tif" />
above formula, every n / 2), the change of fasenScT needs to rotate an amount, but it can be random. For example, according to the predetermined period (cycle) of y (t), the phase may change by means of sequential multiplication, as shown in Mathematics 46 (formula 46) and Mathematics 47 (formula 47). The key point of regular phase change is that the phase of the modulated signal changes regularly. The degree of the phase change is preferably as uniform as possible, such as from -n radians to π radians. However, since this describes a distribution, random changes are also possible.
[Mathematics 46] (formula 46)
Ίπ 4π e <sup>5</sup> -> e <sup>5</sup> , Ίπ, 8π
7— 7— ·> _Q -> _g, π, Ίπ e<sup>J0</sup> -> e ~<sup>¡</sup> ->
, 6π _Q 5 Q, 9π
<img file="MX337079B_D0070.tif" />
[Mathematics 47] (formula 47), π_ .3 / re '<sup>2</sup> -ye<sup>j</sup>-Ye <sup>2</sup>
J ~ A<sup>n</sup> _4 _ .5tt - »e .Ίπ
j] ϊπ __g
<img file="MX337079B_D0071.tif" />
In itself, the weighting unit 60 0 of figure w ιιιιιιι iiiagaggaMü
<img file="MX337079B_D0072.tif" />
fixed and predetermined precoding, and phase changer 317B changes the phase of the signal input as the degree of phase change varies regularly.
When using a specialized precoding matrix in a LOS environment, the quality is likely to improve tremendously. However, depending on the conditions of the direct waves, the phase and amplitude components of the direct wave may differ greatly from the specialized precoding matrix, upon reception. The LOS environment has certain rules. Therefore, the quality of data reception is tremendously improved by means of a regular change applied to a transmission signal that obeys these rules. The present invention offers a signal processing scheme for improvements to the LOS environment.
Figure 7 illustrates an exemplary exemplary configuration of a receiving device 700 belonging to the present embodiment. The wireless unit 703_X receives as input the signal 702_X that the antenna 701_X receives, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs the baseband signal 704_X.
The channel jitter estimator 705_l of the modulated signal zl transmitted by the transmission device takes the baseband signal 704 X as input, extracts the symbol of
<img file="MX337079B_D0073.tif" />
reference 501_l for the channel estimation of the f igurTTT'e'st 1 the value of hu of the Mathematics 4 0 (formula 40) and outputs the channel estimation signal 706_l.
The channel jitter estimator 7 05_2 of the modulated signal z2 transmitted by the transmission device takes the baseband signal 704_X as input, extracts the reference symbol 501_2 for the channel estimation of figure 5, estimates the value of hi<sub>2</sub> of Mathematics 4 0 (formula 40) and outputs the channel estimation signal 706__2.
The wireless unit 703_Y receives, as input, the signal 702_Y that the antenna 701_X receives, performs processing such as frequency conversion, quadrature demodulation and other similar procedures, and outputs the baseband signal 704_Y.
The channel jitter estimator 707_l of the modulated signal zl transmitted by the transmission device takes the baseband signal 704_Y as input, extracts the reference symbol 501_l for the channel estimation of figure 5, est is the value of h<sub>2i</sub> of Mathematics 40 (formula 40) and outputs the channel estimation signal 708_l.
The channel jitter estimator 707_2 of the modulated signal z2 transmitted by the transmission device takes the baseband signal 704_Y as input, extracts the reference symbol 501_2 for the channel estimation of figure 5, estimates the value of h<sub>22</sub> of Mathematics 40 (formula 40) and outputs the
<img file="MX337079B_D0074.tif" />
channel estimation signal 708_2.
A control information decoder 709 receives the baseband signal 7 04_X and the baseband signal 7 04_Y as input, detects the symbol 500_l indicating the transmission scheme of FIG. 5, and outputs an information signal of 710 transmission scheme for the transmission device.
A signal processor 711 takes the baseband signals 704_X and 704_Y, the channel estimation signals 706_1, 706_2, 708_l, and 708_2, and the transmission scheme information signal 710 as input, performs detection and decoding and then outputs the received data 712_1 and 712_2.
Next, the operations of the signal processor 711 of FIG. 7 are described in detail. Figure 8 illustrates an exemplary configuration of the signal processor 711 belonging to the present embodiment. As shown, the signal processor 711 is primarily comprised of an INTERNAL MIMO detector, software input / output decoders, and a coefficient generator. Non-patent Bibliography 2 and Non-patent Bibliography 3 describe an iterative decoding scheme using this structure. The MIMO system described in Non-patent Bibliography 2 and Non-patent Bibliography 3 is a spatial multiplexing MIMO system, while the present modality differs from Non-patent Bibliography 2 and Non-patent Bibliography.
<img file="MX337079B_D0075.tif" />
patent 3 when describing a MIMO system that changes
<img file="MX337079B_D0076.tif" />
the phase in time while using the same precoding matrix. If we take the matrix (channel) H (t) from Mathematics 36 (formula 36), letting the precoding weight matrix of figure 6 be F (here, a fixed precoding matrix that remains unchanged for a certain signal received) and letting the phase change formula used by the phase changer in figure 6 be Y (t) (here, Y (t) changes at time t), then the reception vector R (t) = (rl (t), r2 (t))<sup>T</sup> and the current vector S (t) = (if (t), s2 (t))<sup>T</sup>, the following function is derived:
[Mathematics 48] (formula 48)
R (/) = 7 / (/) xy (/) xFx5 (/) where
Here, the receiving device can use the decoding schemes of Non-patent Bibliography 2 and 3 in R (t) by computing H (t) xY (t) xF.
Accordingly, the coefficient generator 819 of FIG. 8 takes a transmission scheme information signal 818 (corresponding to 710 of FIG. 7) indicated by the transmission device (information to specify
<img file="MX337079B_D0077.tif" />
phase used when the phase changes) and outputs a signal processing scheme information signal 820.
The MIMO-INTERNAL detector 803 takes the signal from the signal processing scheme information as input and performs iterative detection and decoding using the signal and its relationship to Mathematics 48 (formula 48). Its operations are described below.
The processing unit illustrated in Figure 8 uses a processing scheme, as illustrated in Figure 10, to perform iterative decoding (iterative detection). First, detection of a modulated (current) signal code word (or frame) if and of a modulated (current) signal code word (or frame) s2 is performed. As a result, the software input / output decoder obtains the logarithmic likelihood ratio of each bit of the modulated signal codeword (or frame) if and from the modulated signal codeword (or frame) ( current) s2. The logarithmic likelihood ratio is then used to perform a second round of detection and decoding. Those operations are performed multiple times (these operations are hereafter called iterative decoding (iterative detection)). The following operations focus on the scheme of creating the logarithmic likelihood ratio of a symbol at a specific time within a plot.
<img file="MX337079B_D0078.tif" />
k;
DELA i '<sup>:</sup>M., ... M '. '·· _INDuíVRíal
In Figure 8, a memory 815 takes the baseband terminal 80IX (corresponding to the baseband signal 704_X of Figure 7), the channel estimation group signal 8 02X (corresponding to the channel estimation signals 706_l and 706_2 of FIG. 7), the baseband signal 8 01Y (corresponding to the baseband signal 704_Y of FIG. 7), and the 802Y channel estimation group signal (corresponding to the channel estimation signals 708_l and 708_2 of FIG. 7) as input, executes (computes) H (t) xY (t) xF of Math 48 (formula 48 ) in order to perform iterative decoding (iterative detection) and store the resulting matrix as a group of transformed channel signals. Memory 815 then outputs the above-described signals as necessary, specifically as the 816X baseband signal, the 817X channel estimate group transformed signal, the 816Y baseband signal transformed, and the estimate group transformed signal. channel 817Y.
Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection).
(Initial detection)
The MIMO-INTERNAL detector 8 03 takes the 801X baseband signal, the 802X channel estimation group signal, the 801Y baseband signal, and the 802Y channel estimation group signal as input. Here, the modulation scheme for the modulated signal
<img file="MX337079B_D0079.tif" />
(current) si and the modulated (current) signal s2 are taken to be 16-QAM.
The INTERNAL MIMO detector 803 first computes H (t) xY (t) xF from the 802X and 802Y channel estimation signal groups, thereby calculating a candidate signal point corresponding to the 801X baseband signal. Figure 11 represents such a calculation. In Figure 11, each black point is a candidate signal point on the IQ plane. Since the modulation scheme is 16-QAM, there are 256 candidate signal points. (However, Figure 11 is only a representation and does not indicate all 256 candidate signal points.) Letting the four bits transmitted in the modulated signal be bO, bl, b2 and b3 and the four bits transmitted in modulated signal s2 be b4, b5, b6 and b7, the candidate signal points corresponding to (bO, bl, b2 , b3, b4, b5, b6, b7) are found in figure 11. The squared Euclidean distance between each candidate signal point and each received signal point 1101 is then computed (corresponding to the baseband signal 801X). The squared Euclidean distance between each point is divided by the noise variance or<sup>2</sup>. Therefore, E is calculated<sub>x</sub>(b0, bl, b2, b3, b4, b5, b6, b7). It means<sub>x</sub> is the squared Euclidean distance between a candidate signal point corresponding to (bO, bl, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance. Here, each of the baseband signals and the modulated signals si and s2 is a complex signal.
I
<img file="MX337079B_D0080.tif" />
MEXICAN INSTITUTE OF INDUSTRIAL PROPERTY
<img file="MX337079B_D0081.tif" />
computes H (t) xY (t) xF from the channel estimation signal groups 8Ό2Χ and 802Y, computes the candidate signal points corresponding to the baseband signal 801Y, computes the Euclidean distance squared between each of the candidate signal points and the received signal points (corresponding to the 801Y baseband signal), and divide the Euclidean distance squared by the noise variance σ<sup>2</sup>. Therefore, E is calculated<sub>AND</sub>(bO, bl, b2, b3, b4, b5, b6, b7). It means<sub>AND</sub> is the squared Euclidean distance between a candidate signal point corresponding to (bO, bl, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance.
Then E is computed<sub>x</sub>(bO, bl, b2, b3, b4, b5 b6, b7) + E<sub>AND</sub>(bO, bl, b2, b3, b4, b5, b6, b7) = E (bO, bl, b2, b3 b4, b5, b6, b7).
The 803 INTERNAL MIMO detector outputs E (bO, bl, b2, b3, b4, b5, b6, b7) as an 804 signal.
The logarithmic likelihood calculator 805A takes signal 804 as input, calculates the logarithmic likelihood of bits bO, bl, b2, and b3, and outputs the logarithmic likelihood signal 806A. Note that this logarithmic likelihood calculation produces the logarithmic likelihood of a bit that is 1 and the logarithmic likelihood of a bit that is 0. The calculation scheme is as shown in Mathematics 28 (formula 28), Mathematics 29 (formula 29) and Mathematics 30 (formula
<img file="MX337079B_D0082.tif" />
30) and details are given in Non-patent Bibliography 2 and
Similarly, the logarithmic likelihood calculator 805A takes signal 804 as input, calculates the logarithmic likelihood of bits bO, bl, b2, and b3, and outputs the logarithmic likelihood signal 8 06B. An interpolator (807A) takes the logarithmic likelihood signal 806A as input, performs the deinterpolation corresponding to that of the interpolator (the interpolator (304A) in Figure 3), and outputs the deinterpolated logarithmic likelihood signal 808A.
Similarly, a deinterpolator (807B) takes the logarithmic likelihood signal 806B as input, performs the deinterpolation corresponding to that of the interpolator (the interpolator (304B) of Figure 3), and outputs the deinterpolated logarithmic likelihood signal 808B .
The logarithmic likelihood ratio calculator 809A takes the deinterpolated logarithmic likelihood signal 808A as input, calculates the logarithmic likelihood ratio of the bits encoded by encoder 302A of FIG. 3, and outputs the logarithmic likelihood ratio signal 810A. .
Similarly, the logarithmic likelihood ratio calculator 809B takes the deinterpolated logarithmic likelihood signal 808B as input, calculates the logarithmic likelihood ratio of the bits encoded by encoder 302B of FIG. 3, and outputs the ratio signal
Τλ / τ τ
<img file="MX337079B_D0083.tif" />
logarithmic likelihood 810B. ---- The 811A software input / output decoder takes the 810A logarithmic likelihood ratio signal as input, performs decoding, and outputs the decoded logarithmic likelihood ratio 812A.
Similarly, the software input / output decoder 811B takes the logarithmic likelihood ratio signal 810B as input, performs decoding, and outputs the decoded logarithmic likelihood ratio 812B.
(Iterative decoding (Iterative detection), k
Iterations)
The interpolator (813A) takes the kl<sup>ava</sup> logarithmic likelihood ratio 812A, decoded by the software input / output decoder as input, performs the interpolation and outputs the interpolated logarithmic likelihood ratio 814A. Here, the interpolation pattern used by the interpolator (813A) is identical to that of the interpolator (304A) in Figure 3.
Another interpolator (813B) takes kl<sup>ava</sup> logarithmic likelihood ratio 812B, decoded by the software input / output decoder as input, performs the interpolation and outputs the interpolated logarithmic likelihood ratio 814B. Here, the interpolation pattern used by the other interpolator (813B) is identical to that of the other interpolator (304B) in Figure 3.
<img file="MX337079B_D0084.tif" />
The INTERNAL MIMO detector 803 will take ^ a- "eñai ^ jaaaíia, base 816X, the channel estimation group transformed signal 817X, the baseband signal 816Y, the channel estimation group transformed signal 817Y, the ratio of interpolated logarithmic likelihood 814A and the interpolated logarithmic likelihood ratio 814B as input. Here, the 816X baseband signal, the 817X channel estimate group transformed signal, the 816Y baseband signal, and the 817Y channel estimate group transformed signal are used in place of the 801X baseband signal, the 802X channel estimation group signal, 801Y baseband signal and 802Y channel estimation group signal because the latter cause delays due to iterative decoding.
The iterative decoding operations of the MIMO-INTERNAL detector 803 differ from the initial detection operations thereof in that the interpolated logarithmic likelihood ratios 814A and 814B are used in signal processing for the former. The INTERNAL MIMO detector 803 first calculates E (bO, bl, b2, b3, b4, b5, b6, b7) in the same way as for the initial detection. Furthermore, the coefficients corresponding to Mathematics 11 (formula 11) and Mathematics 32 (formula 32) are computed from the interpolated log likelihood ratios 814A and 814B. The value of E (b0, bl, b2, b3, b4, b5, b6, b7) is corrected using the coefficients thus calculated to obtain E '(bO, bl, b2, b3, b4, b5, b6, b7), ΐΝ5ΤΐΤ!. · γ Ο ___ ¡• • ΒΙΛΪΚ, ΙΛΙ. ~ ca ^ - 'L ^ **' which is output as signal 804.
The logarithmic likelihood calculator TCr ^ A will take signal 804 as input, computes the logarithmic likelihood of bits b0, bl, b2, and b3, and outputs the logarithmic likelihood signal 806A. Note that this logarithmic likelihood calculation produces the logarithmic likelihood of a bit that is 1 and the logarithmic likelihood of a bit that is 0. The calculation scheme is as shown in Mathematics 31 (formula
31) to Mathematics 35 (formula 35), and the details are given in the non-patent Bibliography 2 and 3.
Similarly, the logarithmic likelihood calculator 805B takes signal 804 as input, calculates the logarithmic likelihood of bits b4, b5, b6, and b7, and outputs the logarithmic likelihood signal 806A. Operations performed by the forward deinterpolator are similar to those performed for initial detection.
Although Figure 8 illustrates the configuration of the signal processor when performing iterative detection, this structure is not absolutely necessary, as good reception enhancements can be obtained only by iterative detection. As long as the components necessary for iterative detection are present, the configuration does not need to include interpolators 813A and 813B. In such a case, the MIMO-INTERNAL detector 803 does not perform iterative detection.
The key point for this modality is the
<img file="MX337079B_D0085.tif" />
calculation of H (t) xY (t) xF. As shown in Non-patent Bibliography 5 and others, QR decomposition can also be used to perform initial detection and iterative detection.
Furthermore, as indicated in the Non-patent Bibliography 11, linear MMSE operations (minimum root mean square error) and ZF (forced zero) linear operations based on H (t) xY (t) xF can be performed when detecting initial.
Figure 9 illustrates the configuration of a signal processor, different from that of Figure 8, which serves as the signal processor for the modulated signals transmitted by the transmission device of Figure 4. The point of difference in Figure 8 is the number of software input / output decoders. A software input / output decoder 901 takes the logarithmic likelihood ratio signals 810A and 810B as input, performs decoding, and outputs a decoded logarithmic likelihood ratio 902. A distributor 903 takes the decoded logarithmic likelihood ratio 902. as input for distribution. Otherwise, the operations are identical to those explained for figure 8.
As already described, when a transmission device according to the present embodiment using a MIMO system transmits multiple modulated signals from multiple antennas, the phase changes in time while multiplying by
<img file="MX337079B_D0086.tif" />
Precoding matrix to change phase regularly results in improvements in data reception quality for a reception device in a LOS environment where direct waves are dominant, in contrast to a conventional spatial multiplexing MIMO system.
In the present embodiment and in particular in the configuration of the receiving device, the number of antennas is limited and explanations are given in this regard. However, the modality can also be applied to a larger number of antennas. In other words, the number of antennas in the receiving device does not affect the operations or the advantageous effects of the present embodiment.
Furthermore, although LDPC codes are described as a particular example, the present embodiment is not limited in that regard. Also, the decoding scheme is not limited to the given sum-product decoding example for the software input / output decoder. Other software input / output decoding schemes can also be used, such as the BCJR, SOVA algorithm and the Max-Log-Map algorithm. Details are provided in Non-Patent Bibliography 6.
Furthermore, although the present embodiment is described using a single carrier scheme, no limitation is provided in that regard. The present modality is also applicable to multi-carrier transmission. Therefore,
INSTITUTE V, 'DE LA Γ'ί'
INÜ!
<img file="MX337079B_D0087.tif" />
The present embodiment may also be effected using, for example, spread spectrum communications, OFDM (orthogonal frequency division multiplexing), SC-FDMA (single carrier frequency division multiple access), SC-OFDM (multiplexing by single carrier orthogonal frequency division), OFDM of mini-waves as described in Non-patent Bibliography 7, etc. Also, in the present embodiment, symbols other than data symbols, such as pilot symbols (preamble, single word, etc.). or symbols that transmit control information, can be arranged within the frame in any way.
The following describes an example where OFDM is used as a multi-carrier scheme.
Figure 12 illustrates the configuration of a transmission device using OFDM. In Figure 12, the components operating in the manner described for Figure 3 use identical reference numbers.
The OFDM-related processor 12OIA takes the weighted signal 309A as input, performs the OFDM-related processing on it, and outputs the transmit signal 1202A. Similarly, the OFDM-related processor 1201B takes the postphase change 309B as input, performs OFDM-related Processing on it, and outputs the transmission signal 1202A.
The figure illustrates a configuration
MPI
MEXICAN INSTITUTE Dn THE INDUSTRIAL PROPERTY
<img file="MX337079B_D0088.tif" />
example of the processors related to OFDl'T 12UÍA<sup></sup>and 1201B and forward of Figure 12. Components 1301A through 1310A belong to between 1201A and 312A of Figure 12, while components 1301B through 1310B belong to between 1201B and 312B.
Serial-to-parallel converter 1302A performs serial-to-parallel conversion on weighted signal 13OIA (corresponding to weighted signal 309A in FIG. 12) and outputs parallel signal 1303A.
The reorder 13 04A takes the parallel signal 13 03A as input, performs the reordering thereof and outputs the reordered signal 13 05A. The reordering is described in detail later.
The IFFT (Inverse Fast Fourier Transform) unit 1306A takes the rearranged signal 1305A as input, applies an IFFT to it, and outputs the post-IFFT signal
1307A.
The wireless unit 1308A takes the post-IFFT signal 1307A as input, performs processing there such as frequency conversion and amplification, and outputs the modulated signal 1309A. The antenna 1310A then outputs the modulated signal 1309A as radio waves.
Serial-to-parallel converter 1302B performs serial-to-parallel conversion on weighted signal 1301B (corresponding to post-phase shift 309B of FIG. 12) and outputs parallel signal 1303B.
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Reorder 13 04B takes Renal-¿an paral ^ .lojin ^ R as input, reorders it and outputs reordered signal 1305B. The reordering is described in detail later.
The IFFT unit 1306B takes the rearranged signal 1305B as input, applies the IFFT to it and outputs the post-IFFT signal
1307B.
The wireless unit 1308B takes the post-IFFT signal 1307B as an input, performs processing such as frequency conversion and amplification on it, and outputs the modulated signal 1309B. The antenna 1310A then outputs the modulated signal 1309B as radio waves.
The transmission device of Figure 3 does not use a multi-carrier transmission scheme. In this way, as shown in Figure 6, the phase change is performed to achieve a period (cycle) of four and the post-phase change symbols are arranged with respect to the time domain. As shown in Figure 12, when using multi-carrier transmission such as OFDM, naturally precoded post-phase shift symbols can be arranged with respect to the time domain as in Figure 3, and this holds for each (sub-) carrier. However, for multi-carrier transmission, the arrangement may also be in the frequency domain or both, the frequency domain and the time domain. The following describes those provisions.
j> <p 7
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Figures 14A and 14B indicate the
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horizontal axes and time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorders 1301A and 1301B of Figure 13. The frequency axes are made up of (sub-) carriers 0 a 9. The modulated signals zl and z2 share common date-time (timing) indicators and use a common frequency band. Figure 14A illustrates a reordering scheme for the z1 modulated signal symbols, while Figure 14B illustrates a reordering scheme for the z2 modulated signal symbols. With respect to the symbols of the weighted signal 1301A to which the serial to parallel converter is input. 1302A, the assigned order is # 0, # 1, # 2, # 3, etc. Here, since the example is about a period (cycle) of four, # 0, # 1, # 2 and # 3 are equivalent to a period (cycle). Similarly, # 4n, # 4n + l, # 4n + 2, and # 4n + 3 (where n is a nonzero positive integer) are also equivalent to a period (cycle).
As shown in Figure 14A, symbols # 0, # 1, # 2, # 3, etc. they are arranged in order, starting at carrier 0. Symbols # 0 through # 9 are given date-time $ 1, followed by symbols # 10 through # 19 which are given date-time # 2, etc. In a regular arrangement. Note that the modulated signals zl and z2 are the complex signals.
Similarly, with respect to the weighted signal symbols 1301B that are input to the converter
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serial to parallel 13 02B, assigned order - # 2 -, "# 37" etc. Here, since the example is about a period (cycle) of four, a different phase change is applied to each of # 0, # 1, # 2, and # 3, which are equivalent to one period (cycle). Similarly, a different phase change is applied to each of # 4n, # 4n + l, # 4n + 2, and # 4n + 3 (where n is a nonzero positive integer), which are also equivalent to a period ( cycle)
As shown in Figure 14B, symbols # 0, # 1, # 2, # 3, etc. they are arranged in order, starting at carrier 0. Symbols # 0 through # 9 are given date-time $ 1, followed by symbols # 10 through # 19 which are given date-time # 2, etc. In a regular arrangement.
The group of symbols 1402 shown in Figure 14B corresponds to a period (cycle) of symbols when using the phase change scheme of Figure 6. Symbol # 0 is the symbol obtained using the phase on the date-time u of figure 6, symbol # 1 is the symbol obtained using the phase on the date-time u + 1 of figure 6, symbol # 2 is the symbol obtained using the phase on the date-time u + 2 of figure 6 and symbol # 3 is the symbol obtained using the phase on the date-time u + 3 of figure
6. Therefore, for any #x symbol, the #x symbol is the symbol obtained using the phase in the date-time u of Figure 6 when x mod 4 equals 0 (i.e. when the remainder of x divided by 4 is 0, mod being the modulo operator), the symbol #x is the symbol obtained using the phase in the date-time u + 1 of the figure
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when x mod 4 equals 1, the symbol #x is the symbol obtained * using the phase in the datetime u + 2 in figure 6 when x mod 4 is equal to 2, and the symbol #x is the symbol obtained using the phase at the time-date u + 3 in figure 6 when x mod 4 equals 3.
In the present embodiment, the zl modulated signal shown in Figure 14A has not undergone a phase change.
As such, when a multi-carrier transmission scheme such as OFDM and different from single-carrier transmission is used, the symbols may be arranged with respect to the frequency domain. Of course, the symbol layout scheme is not limited to those illustrated in Figures 14A and 14B. Additional examples are shown in Figures 15A, 15B,
16A, and 16B.
Figures 15A and 15B indicate the frequency on the horizontal axes and the time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorders 1301A and 1301B of Figure 13 that differs from that of the Figures 14A and 14B. Figure 15A illustrates a reordering scheme for the z1 modulated signal symbols, while Figure 15B illustrates a reordering scheme for the z2 modulated signal symbols. Figures 15A and 15B differ from Figures 14A and 14B in that different rearrangement schemes are applied to the symbols of the modulated signal zl and to the symbols of the modulated signal z2. In Figure 15B, the
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Symbols # 0 through # 5 are arranged on carriers 4 9, Iqr # 6 through # 9 are arranged on carriers 0 through 3, and this arrangement is repeated for symbols # 10 through # 19. Here, as in Figure 14B, symbol group 1502 shown in Figure 15B corresponds to a symbol period (cycle) when using the phase shift scheme of Figure 6.
Figures 16A and 16B indicate the frequency on the horizontal axes and the time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorders 1301A and 1301B of Figure 13 that differs from that of the Figures 14A and 14B. Figure 16A illustrates a reordering scheme for the z1 modulated signal symbols, while Figure 16B illustrates a reordering scheme for the z2 modulated signal symbols. Figures 16A and 16B differ from Figures 14A and 14B in that while Figures 14A and 14B show the symbols arranged on the sequential carriers, Figures 16A and 16B do not arrange the symbols on the sequential carriers. Obviously, in the case of Figures 16A and 16B, different rearrangement schemes may be applied to the symbols of the modulated signal zl and to the symbols of the modulated signal z2 than those of Figures 15A and 15B.
Figures 17A and 17B indicate the frequency on the horizontal axes and the time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorders 1301A and 1301B of Figure 13 which
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differs from those of Figures 14A to 16B. Figure 17A illustrates a reordering scheme for the z1 modulated signal symbols and Figure 17B illustrates a reordering scheme for the z2 modulated signal symbols. While Figures 14A through 16B show the symbols arranged with respect to the frequency axis, Figures 17A and 17B use the frequency and time axes together in a single arrangement.
While Figure 6 describes an example where a phase change is made in a four interval period (cycle), the following example describes an eight interval period (cycle). In Figures 17A and 17B, symbol group 1702 is equivalent to one period (cycle) of symbols when using the phase shift scheme (i.e. eight symbols) such that symbol # 0 is the symbol obtained using the phase on the date-time u, symbol # 1 is the symbol obtained using the phase on the date-time u + 1, symbol # 2 is the symbol obtained using the phase on the date-time u + 2, symbol # 3 is the symbol obtained using the phase on the datetime u + 3, symbol # 4 is the symbol obtained using the phase on the date-time u + 4, symbol # 5 is the symbol obtained using the phase on the date-time u + 5, symbol # 6 is the symbol obtained using the phase on the date-time u + 6 and symbol # 7 is the symbol obtained using the phase on the date-time u + 7. Therefore, for any #x symbol, the #x symbol is the symbol obtained using the phase in the datetime u when x mod 8 equals 0, the #x symbol is the obtained symbol
MEXICAN INSTITUTE
OF THE PROPERTY
INDUSTRIAL using the phase on the date-time u + l when x mod 8 is equal to lp the symbol #x is the symbol obtained using the phase on the date-time u + 2 when x mod 8 equals 2, the symbol #x is the symbol obtained using the phase on the datetime u + 3 when x mod 8 equals 3, the symbol #x is the symbol obtained using the phase on the datetime u + 4 when x mod 8 is equal to 4, the symbol #x is the symbol obtained using the phase in the datetime u + 5 when x mod 8 equals 5, the #x symbol is the symbol obtained using the phase at the time date u + 6 when x mod 8 equals 6, and the #x symbol is the symbol obtained using the phase at the time date u + 7 when x mod 8 equals 7. Figures 17A and 17B use four intervals along the time axis and two intervals along the frequency axis for a total of 4x2 = 8 intervals, in which a period is provided (cycle) of symbols. Here, given mxn symbols per period (cycle) (i.e., mxn different phases are available for multiplication), then n intervals (carriers) in the frequency domain and m intervals in the time domain must be used to arrange the symbols for each period (cycle), so that m> n. This is because the phase of direct waves fluctuates slowly in the time domain with respect to the frequency domain. Accordingly, the present embodiment effects a regular phase change that reduces the influence of stable direct waves. Thus, the phase change period (cycle) should preferably reduce direct wave fluctuations. Therefore, m must be greater than n. Having
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In view of the above, the joint use of the ^ omlnióS dé 'Lltíllipu and frequency for the reordering, as shown in Figures 17A and 17B, is preferred in relation to the use of either the frequency domain or the time domain alone. , due to the strong probability that direct waves become regular
As a result, the effects of the present invention are more easily obtained. However, rearrangement in the frequency domain can lead to diversity gain due to the fact that the frequency domain fluctuations are abrupt. By itself, the joint use of frequency and time domains for reordering is not always ideal.
Figures 18A and 18B indicate the frequency on the horizontal axes and the time on the vertical axes thereof, and illustrate an example of a symbol reordering scheme used by the reorders 1301A and 1301B of Figure 13 that differs from that of Figures. 17A and 14B. Figure 18A illustrates a reordering scheme for the z1 modulated signal symbols, while Figure 18B illustrates a reordering scheme for the z2 modulated signal symbols. Similar to Figures 17A and 17B, Figures 18A and 18B illustrate the joint use of time and frequency domains. However, in contrast to Figures 17A and 17B, where the frequency domain is prioritized and the time domain is used for the arrangement of the secondary symbols, Figures 18A and 18B prioritize the time domain and the use of the domain of frequency for the arrangement of symbols
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secondary. In Figure 18B, the group of ^ symbols 1802 corresponds to a period (cycle) of symbols when the phase change scheme is used.
In Figures 17A, 17B, 18A and 18B, the reordering scheme applied to the symbols of the modulated signal zl and the symbols of the modulated signal z2 may be identical or differ as in Figures 15A and 15B. Both approaches allow good reception quality. Furthermore, in Figures 17A, 17B, 18A and 18B, the symbols can be arranged non-sequentially as in Figures 16A and 16B. Both approaches allow good reception quality.
Figure 22 indicates the frequency on the horizontal axis and the time on the vertical axis thereof, and illustrates an example of a symbol reordering scheme used by the reorders 1301A and 1301B of Figure 13 that differs from the preceding one. Figure 22 illustrates a regular phase change scheme using four intervals, similar to the date-time indicators ua u + 3 in Figure 6. The characteristic feature of Figure 22 is that, although the symbols are rearranged with respect to the frequency domain, when read along the time axis, a periodic shift of n (n = 1 is evident in the example in Figure 22) symbols. The group of symbols of the frequency domain 2210 in figure 22 indicates four symbols to which the phase change is applied in the date-time indicators ua u + 3 of figure 6.
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Here, symbol # 0 is obtained by pof'mgtTTtrte ^ UTT ^^ eai ^ i © ™ ».. phase on the date-time u, symbol # 1 is obtained by means of a phase change on the date-time u + 1; symbol # 2 is obtained by means of a phase change on the date-time u + 2 and symbol # 3 is obtained by means of a phase change on the date-time u + 3.
Similarly, for the group of symbols in the frequency domain 2220, symbol # 4 is obtained by means of a phase change in the time-date u, symbol # 5 is obtained by means of a phase change in the date-time u + 1, symbol # 6 is obtained by a phase change on the date-time u + 2 and symbol # 7 is obtained by a phase change on the date-time u + 3 .
The phase change described above applies to the symbol on the $ 1 date-time. However, in order to apply periodic offset in the time domain, the following phase shifts are applied to symbol groups 2201, 2202, 2203, and 2204.
For the symbol group of time domain 2201, symbol # 0 is obtained by means of a phase change on the date-time u, symbol # 9 is obtained by means of a phase change on the date-time u +1, symbol # 18 is obtained by a phase change on the date-time u + 2 and symbol # 27 is obtained by a phase change on the date-time u + 3.
For the symbol group of time domain 2202, symbol # 28 is obtained by means of a phase change in the
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date-time u, symbol # 1 is obtained by medTS ^ e ^ un ”oambd.o..jie phase on time-date u + 1, symbol # 10 is obtained by phase change on date -time u + 2 and the symbol # 19 is obtained by means of a phase change in the datetime u + 3.
For the time domain 2203 symbol group, symbol # 20 is obtained by phase change on the date-time u, symbol # 29 is obtained by phase-change on the date-time u +1, symbol # 2 is obtained by a phase change on the date-time u + 2 and symbol # 11 is obtained by a phase change on the date-time u + 3.
For the symbol group of time domain 2204, symbol # 12 is obtained by means of a p hase change on the date-time u, symbol # 21 is obtained by means of a phase change on the date-time u +1, symbol # 30 is obtained by a phase change on the date-time u + 2 and symbol # 3 is obtained by a phase change on the date-time u + 3.
The characteristic feature of figure 22 is seen in that, taking symbol # 11 as an example, the two neighboring symbols of it that have the same date-time in the frequency domain (# 10 and # 12) are both symbols that are change using a different phase from symbol # 11, and the two neighboring symbols of it that have the same carrier in the time domain (# 2 and # 20) are both symbols that are changed using a different phase from the symbol #eleven. This is true not only for symbol # 11, but also for any symbol that has two neighboring symbols in
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the frequency domain and the time domain. Accordingly, the phase change is effectively carried out. This is highly likely to improve the quality of data reception as the influence of direct wave regularization is less prone to reception.
Although Figure 22 illustrates an example where n = 1, the invention is not limited in that regard. The same can be applied to a case where n = 3. Likewise, although Figure 22 illustrates the modality of the effects previously described arranging the symbols in the frequency domain and advancing in the time domain as to achieve the characteristic effect of imparting a periodic displacement to the order of the arrangement of symbols, for the The same effect symbols can also be arranged randomly (or regularly).
Mode 2
In Mode 1 described above, the phase change is applied to a weighted signal (precoded with a fixed precoding matrix) z (t). The following modalities describe various phase change schemes by which the effects of Mode 1 can be obtained.
In the previously described embodiment, as shown in Figures 3 and 6, the phase changer 317B is configured to perform a phase change on only one of the signals output by the weighting unit 600.
However, the phase change can also be applied
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before precoding the unit-'de-peftde «rekeieia- £ ÍLÍl<sub>-</sub>"In addition to the components illustrated in FIG. 6, the transmission device may also have weighting unit 600 before phase changer 317B, as shown in FIG. 25.
In such circumstances, the following configuration is possible. The phase changer 317B performs a regular phase change with respect to the baseband signal s2 (t), in which the correlation has been performed according to a selected modulation scheme, and outputs s2 '(t) = s2 (t) y (t) (where y (t) varies in time t). The weighting unit 600 executes the premodification at s2't, outputs z2 (t) = W2s2 '(t) (see Math 42 (formula 42)), and then outputs the result.
As an alternative, the phase change can be performed on both modulated signals si (t) and s2 (t). The transmission device itself is configured to include a phase changer that takes both signals output by the weighting unit 600, as shown in Figure 26.
Like phase shifter 317B, phase shifter 317A performs a regular phase shift at the signal input, and as such changes the signal phase zl '(t) precoded by the weighting unit. A transmitter then outputs the post-phase shift signal zl (t).
However, the phase change rate applied by phase changers 317A and 317B varies simultaneously in order
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to perform the phase change shown in ± a<sup>,</sup>figüFá'2D ”<sup>=</sup>TLÓ ”<sup>=</sup>The following describes a non-limiting example of the phase change scheme). For the date-time u, the phase shifter 317A of Fig. 26 performs the phase change such that zl (t) = yxítjzl '(t), while the phase shifter 317B performs the phase change so that z2 (t) = y2 (t) z2 '(t). For example, as shown in Figure 26, for the date-time u, yx (u) = e<sup>j0</sup> y y2 (u) = e<sup>jLj / 2</sup>, for the date-time u + 1, yi (u + l) = e<sup>jD / 4</sup>.y y2 (u + l) = e<sup>_j3D / 4</sup>, and for the date-time u + k, yi (u + k) = e<sup>JkD / 4</sup> y y2 (u + k) = <sub>β</sub><sup>5ίΛ3α / 4</sup><sup>D / 2></sup> . Here, the period of regular phase change (cycle) may be the same for both phase changers 317A and 317B or may vary for each.
In addition, as already described, a phase change can be performed before the weighting unit is precoded. In such a case, the transmission device should be configured as illustrated in Figure 27.
When a phase change is performed on both modulated signals, each of the transmission signals for example, is control information that includes information about the phase change pattern. Obtaining the control information, the receiving device knows the phase change scheme by which the transmission device regularly varies the change, that is, the phase change pattern and, in this way, it can demodulate (decode) the signs correctly.
Variants of the
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Exemplary configurations shown in Figures 6 and 25 with reference to Figures 28 and 29. Figure 28 differs from Figure 6 in the inclusion of the phase change on / off information 2800 and in that the phase change is performed in only one of zl '(t) and z2' (t) (that is, it is performed on one of zl '(t) and z2' (t), which have identical date-time indicators or a common frequency). Accordingly, in order to perform the phase change on one of zl '(t) and z2' (t), the phase changers 317A and 317B shown in Figure 28 can each be activated and perform the change of phase or deactivated and do not make the phase change. The ON / OFF information of the phase change 2800 is the control information for this. An information generator for signal processing schemes 314 shown in Figure 3 outputs the ON / OFF information of phase change 2800.
The phase shifter 317A in Fig. 28 changes the phase to produce zl (t) = y<sub>x</sub>(t) zl '(t), while the phase changer 317B changes the phase to produce z2 (t) = y<sub>2</sub>(t) z2 '(t).
Here, a phase change having a period (cycle) of four is applied, for example, to zl '(t). (Meanwhile, the phase of z2 '(t) does not change.) Therefore, for the date-time u, and<sub>x</sub>(u) = <sup>eg</sup>° Y Y2 (u) = 1, for the date-time u + 1, yx (u + l) = e<sup>JD / 2</sup> y y2 (u + l) = 1, for the time date u + 2, and<sub>x</sub>(u + 2) = e<sup>í0</sup> y y2 (u + 2) = 1 and for the date-time u + 3 yx (u + 3) = e<sup>J3D / 2</sup> y y2 (u + 3) = 1.
Next, a phase change having a period (cycle) of four is applied, for example, to z2 '(t). (Meanwhile,
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does not change the phase of zl '(t)). Therefore, for the datetime u + 4, yi (u + 4) = 1 yy<sub>2</sub>(u + 4) = e<sup>J</sup>°, for the date-time u + 5, yx (u + 5) = 1 and y2 (u + 5) = e<sup>3</sup>'<sup>D / 2</sup>, for the date-time u + 6, yi (u + 6) = 1 and y2 (u + 6) = e<sup>JLj</sup> and for the date-time u + 7 and<sub>x</sub>(u + 7) = 1 yy<sub>2</sub>(u + 7) = e<sup>j3n / 2</sup>.
Therefore, given the preceding examples, for any 8k date-time, and<sub>x</sub>(8k) = e<sup>J</sup>° yy<sub>2</sub>(8k) =
1,
<td></td><td>for</td><td>any</td><td>date hour</td><td>8k + l,</td><td>and<sub>x</sub>(8k + l) = e<sup>jQ / 2</sup></td><td>and</td>
<td>and<sub>2</sub>(8k + i)</td><td>= 1, for</td><td>any</td><td>date hour</td><td>8k + 2,</td><td>yi (8k + 2) = e<sup>JD</sup></td><td>and</td>
<td>and<sub>2</sub> (8k + 2)</td><td>= 1, for</td><td>any</td><td>date hour</td><td>8k + 3,</td><td>and<sub>x</sub>(8k + 3) = e<sup>j3Q / 2</sup></td><td>and</td>
= e<sup>j0</sup>, = e<sup>JD / 2</sup>, = e<sup>J</sup>’<sup>D</sup>, and<sub>2</sub>(8k + 3) = 1, for any 8k + 4 datetime, and<sub>2</sub> (8k + 4) = 1 yy<sub>2</sub> (8k + 4) for any time date 8k + 5, yi (8k + 3) = lyy<sub>2</sub>(8k + 5) for any 8k + 6 date-time, and<sub>x</sub>(8k + 6) = lyy<sub>2</sub>(8k + 6) for any datetime 8k + 7, yi (8k + 7) = lyy<sub>2</sub>(8k + 7)
As already described, there are two intervals, one where the phase change is made only in zl '(t), and another where the phase change is made only in z2' (t). Also, the two intervals form a phase change period (cycle). While j3D / 2
I iüdví / íkí;.!.
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the preceding explanation describes as equals eT'inüér Vello where 'the phase change is made only in zl' (t) and the interval where the phase change is made only in z2 '(t), no limitation is foreseen in that sense. The two intervals may also differ. Furthermore, although the preceding explanation describes the mode of a phase change with a period (cycle) of four only in zl '(t) and then the mode of a phase change with a period (cycle) of four only in z2 '(t), no limitation is provided in this regard. Phase changes can be performed at zl '(t) and z2' (t) in any order (for example, phase change can alternate between being performed at zl '(t) and z2' (t), or can be done in random order).
The phase shifter 317A in Fig. 29 changes the phase to produce if '(t) = yi (t) sl (t), while the phase shifter 317B changes the phase to produce s2' (t) = y<sub>2</sub>(t) s2 (t).
Here, a phase change with a period (cycle) of four is applied, for example, to sl (t). (Meanwhile, s2 (t) remains unchanged.) Therefore, for the date-time u, yi (u) = e ^<sup>0</sup> y y2 (u) = 1, for the datetime u + 1, ya. (u + 1) = e<sup>jD / 2</sup> y y2 (u + l) = 1, for the date-time u + 2, ya (u + 2) = yy<sub>2</sub>(u + 2) = 1 and for the date-time u + 3, and<sub>x</sub>(u + 3) = e<sup>j3D / 2</sup> yy<sub>2</sub>(u + 3) = 1.
Then a phase change with a period (cycle) of four is applied, for example, to s2 (t). (Meanwhile, sl (t) remains unchanged.) Therefore, for the datetime u + 4, ya (u + 4) = 1 yy<sub>2</sub>(u + 4) = e<sup>J</sup>°, for the date-time
Txí.r.
u + 5, and<sub>x</sub>(u + 5) = 1 yy<sub>2</sub>(u + 5) = e<sup>ha / 2</sup>, for the date ^ óTShü + S7 '<sup>r</sup>Vi fu + §T '= 1 yy<sub>2</sub>(u + 6) = e<sup>jD</sup> and for the datetime u + 7, and<sub>x</sub>(u + 7) = 1 yy<sub>2</sub>(u + 7) _ θJ3Z1 / 2.
Therefore, given the preceding examples, for any 8k date-time, and<sub>x</sub>(8k) = e<sup>J</sup>° yy<sub>2</sub>(8k) =
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1,
<td></td><td>For any</td><td>date hour</td><td>8k + l,</td><td>and<sub>x</sub>(8k + l) = e<sup>jn / 2</sup></td><td>and</td>
<td>and<sub>2</sub> (8k + l)</td><td>= 1, For any</td><td>date hour</td><td>8k + 2,</td><td>and<sub>x</sub>(8k + 2) = e<sup>J</sup>°</td><td>and</td>
<td>and<sub>2</sub> (8k + 2)</td><td>= 1, For any</td><td>date hour</td><td>8k + 3,</td><td>and<sub>x</sub>(8k + 3) = e<sup>j3O / 2</sup></td><td>and</td>
= e<sup>W2</sup>, and<sub>2</sub> (8k + 3) = 1, for any 8k + 4 datetime, and<sub>x</sub> (8k + 4) = 1 yy<sub>2</sub> (8k + 4) for any 8k + 5 date-time, and<sub>x</sub>(8k + 5) = lyy<sub>2</sub>(8k + 5) for any 8k + 6 date-time, and<sub>x</sub>(8k + 6) = lyy<sub>2</sub>(8k + 6) for any 8k + 7 date-time, and<sub>x</sub> (8k + 7) = 1 yy<sub>2</sub> (8k + 7)
As already described, there are two intervals, one where the phase change is made only in sl (t) and the other where the phase change is made only in s2 (t). Also, the two intervals form a phase change period (cycle). Although the preceding explanation describes as equal the interval where
J3CI / 2 | -ΊΙΙ | · 1, II II, ι. U.1,.
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the phase change is made only in sl (t) and the interval where the phase change is only made in s2 (t), no limitation is provided in this sense. The two intervals may also differ. Furthermore, although the preceding explanation describes the phase change mode with a period (cycle) of four only in sl (t) and then the phase change mode with a period (cycle) of four only in s2 (t) , no limitation is provided in this regard. Phase changes can be made in sl (t) and s2 (t) in any order (for example, they can alternate between being made in sl (t) and s2 (t) or they can be made in random order).
Accordingly, the reception conditions are equalized under which the reception device receives each transmission signal zl (t) and z2 (t). By periodically changing the phase of symbols on the received signals zl (t) and z2 (t), you can increase the chance that error correction codes will correct errors, thereby improving signal quality.
received in the LOS environment.
Therefore, Mode 2 as already described can produce the same results as Mode 1 described above.
Although the present embodiment uses a single carrier scheme, that is, the phase change of the time domain, as an example, no limitation is provided in this regard. The same effects can also be achieved using multi-carrier transmission. Accordingly, it should also be performed using, for example, spread spectrum communications, OFDM, SC-FDMA (single carrier frequency division multiple access), SC-OFDM, OFDM of mini-waves, as described in the Non-patent bibliography 7, etc. As previously described, although the present modality explains the phase change as the phase change with respect to the time domain t, as an alternative the phase can change with respect to the frequency domain, as explained in Mode 1. That is, considering the phase change scheme in the time domain t described in the present modality and replacing t with f (where f is the frequency ((sub-) carrier)) leads to a phase change applicable to the frequency domain . Furthermore, as already explained in relation to Mode 1, the phase change scheme of the present modality is also applicable to the phase change with respect to both the time domain and the frequency domain.
Accordingly, although Figures 6, 25, 26, and 27 illustrate phase changes in the time domain, the replacement of time t by carrier f in each of Figures 6, 25, 26, and 27 corresponds to a change phase in the frequency domain. In other words, replacing (t) with (t, f) where t is time and f is frequency corresponds to making the phase change in the time-frequency blocks.
Likewise, in this modality,
<img file="MX337079B_D0109.tif" />
<img file="MX337079B_D0110.tif" />
within the frame in any way symbols other than data symbols, such as pilot symbols (preamble, single word, etc.) or symbols that transmit control information.
Mode 3
Modalities 1 and 2, described above, explain the regular phase changes. Mode 3 describes an admission scheme of the receiving device to obtain good quality of the signal received from the data, regardless of the arrangement of the receiving device, considering the location of the receiving device with respect to the transmitting device.
Mode 3 refers to the arrangement of the symbols within the signals obtained by means of a phase change.
Figure 31 illustrates an example frame configuration for a portion of the symbols wi thin a signal in the time-frequency domain, given a transmission scheme where a regular phase change is performed for a multi-carrier scheme, such as OFDM .
First, an example is explained in which the phase change is performed on one of two baseband signals, precoded in the way explained in Mode 1 (see Figure 6).
(Although Figure 6 illustrates a phase change in the time domain, the change in time t by carrier f in
<img file="MX337079B_D0111.tif" />
Figure 6 corresponds to a phase change erf ~ el ~ domi ηιδ of frequency. In other words, replacing (t) with (t, f) where t is the time and f is the frequency, corresponds to making phase changes in the time-frequency blocks).
Figure 31 illustrates the frame configuration of the modulated signal z2 ', which is input to the phase shifter 317B of Figure 12. Each square represents a symbol (although both si and s2 signals are included for the purpose of the precoding, according to the precoding matrix, only one of signals can be used if and s2).
Consider the symbol 3100 on Carrier 2 and the datetime $ 2 in Fig. 31. The carrier described here can be referred to as a subcarrier alternative.
Within carrier 2, there is a very strong correlation between the channel conditions for the 3100 symbol on carrier 2, the time-date $ 2, and the channel conditions for the time domain symbols closest to the time-date $ 2 , that is, symbol 3013 on date-time $ 1 and symbol 3101 on date-time $ 3 inside carrier 2.
Similarly, for datetime $ 2, there is a very strong correlation between channel conditions for symbol 3100 on carrier 2, datetime $ 2, and channel conditions for symbols in the frequency domain closest to carrier 2, that is, symbol 3104 on carrier 1, datetime $ 2, and symbol 3104 on datetime $ 2, carrier 3.
<img file="MX337079B_D0112.tif" />
As already described, there is a very strong correlation between channel conditions for symbol 3100 and channel conditions for symbols 3101, 3102, 3103, and 3104.
The present description considers N different phases (N being an integer, N 2) for multiplication in a transmission scheme where the phase changes regularly. The symbols illustrated in Figure 31 are indicated as e<sup>J</sup>°, for example. This means that this symbol is the signal z2 'of figure 6 with phase change through multiplication by e<sup>J</sup>°. That is, the values indicated in figure 31 for each of the symbols are the values of y (t) of Mathematics 42 (formula 42), which are also the values of z2 (t) = y<sub>2</sub>(t) z2 '(t) described in Mode 2.
The present embodiment takes advantage of the high correlation in the channel conditions existing between the neighboring symbols in the frequency domain and / or the neighboring symbols in the time domain, in a symbol arrangement that enables the receiving device that receives the symbols with the phase changed obtain a high quality of data reception.
In order to achieve this high quality of data reception, conditions # 1 and # 2 are necessary.
(Condition # 1)
As shown in Figure 6, for a transmission scheme involving a regular phase change performed on the precoded baseband signal z2 'using transmission
<img file="MX337079B_D0113.tif" />
multi-carrier, such as OFDM, time X, carrier Y is a symbol for transmitting the data (hereinafter the data symbol), the neighboring symbols in the time domain, i.e. at time Xl, the carrier Y and at time X + l, carrier Y are also data symbols, and a different phase change must be made to the precoded baseband signal z2 'corresponding to each of the three data symbols, i.e. at the precoded baseband signal z2 'at time X, carrier Y, at time Xl, carrier Y, and at time X + l, carrier Y.
(Condition # 2)
As shown in Figure 6, for a transmission scheme involving a regular phase change performed on the precoded baseband signal z2 'using multi-carrier transmission such as OFDM, time X, carrier Y is a symbol data, neighboring symbols in the frequency domain, i.e. at time X, carrier Yl and at time X, carrier Y + l are also data symbols, and a different phase change must be made in the precoded baseband signal z2 'corresponding to each of the three data symbols, that is, in the precoded baseband signal z2' at time X, carrier Y, in at time X, carrier Yl and at time X, carrier Y + l.
Ideally, data symbols that meet Condition # 1 should be present. Similarly, they must be
<img file="MX337079B_D0114.tif" />
present the data symbols that meet l ^ - ^ ondreión # 2i „
The reasons for Conditions # 1 and # 2 are the following.
There is a very strong correlation between the channel conditions of the given symbol of a transmit signal (hereafter, symbol A) and the channel conditions of symbols neighboring symbol A in the time domain, as already described.
Consequently, when three neighboring symbols in the time domain each have different phases, despite degradation of reception quality in the LOS environment (poor signal quality caused by degradation of conditions due to phase relationships of direct waves, despite the high quality of if in terms of SNR) for symbol A, the remaining two symbols to symbol A are very likely to provide good reception quality. As a result, good received signal quality can be achieved after error correction and decoding.
Similarly, there is a very strong correlation between the channel conditions of the given symbol of a transmission signal (hereafter, symbol A) and the channel conditions of symbols neighboring symbol A in the frequency domain, as already described.
Therefore, when three neighboring symbols in the frequency domain each have different phases, despite
<img file="MX337079B_D0115.tif" />
of the degradation of reception quality in the éntorao ^ CDS ^ ° ~ ““ “(poor signal quality caused by the degradation of the conditions due to the phase relationships of direct waves despite the high signal quality in terms of SNR) for symbol A, the two remaining symbols neighboring symbol A are highly likely to provide good reception quality.
As a result, you can achieve good received signal quality after error correction and decoding.
By combining Conditions # 1 and # 2, it is likely that even higher data reception quality can be achieved for the receiving device. Therefore, it can be derived to next Condition # 3.
(Condition # 3)
As shown in Figure 6, for a transmission scheme involving a regular phase change performed on the precoded baseband signal z2 'using multi-carrier transmission such as OFDM, time X, carrier Y is a symbol data, the neighboring symbols in the time domain, that is, at time Xl, carrier Y and at time X + l, carrier Y are also data symbols, and neighboring symbols in the frequency domain, that is, at time X, carrier Yl and at time X, carrier Y + l are also data symbols, and a different phase change must be made to the precoded baseband signal z2 'corresponding to each of these five data symbols, i.e. , at the sign of
<img file="MX337079B_D0116.tif" />
<img file="MX337079B_D0117.tif" />
precoded baseband z2 'at time X, carrier Y, at time X, carrier Yl, at time X, carrier Y + l, at time Xl, carrier Y, and at time X + l , the carrier
AND.
Here, the different phase changes are as follows. Phase changes are defined from 0 radians to 2Ü radians. For example, for time X, carrier Y, a phase change of is applied to the precoded baseband signal z2 'of FIG. 6, for time Xl, carrier Y, a phase change of e is applied.<sup>j0x</sup><sup>1, Y</sup> to the precoded baseband signal z2 'of Figure 6, for time X + l, carrier Y, a phase change of e is applied<sup>j6x + 1, Y</sup> to the precoded baseband signal z2 'in Figure 6, so that 0 θχ, γ <20, 0 ^ θχ-ι, γ <2Π, and 0 θχ + ι, γ <2Ü, all units being radians . Therefore, for Condition # 1, it follows that θχ, γ / θχ-ι, γ, θχ, γ 4 θχ + ι, γ, and that θχ-ι, γ / θχ + ι, γ · Similarly, for Condition # 2 follows that θχ, γ / θχ, γ-ι, θχ, γ θχ, γ + ι, and that θχ, γ-ι 4 θχ, γ + ι · and, for Condition # 3, follows that θχ, γ θχ-ι, γ, θχ, γ Ψ θχ + ι, γ, θχ, γ θχ, γ-ι, θχ, γ θχ, γ-l, θχ-ι, γ 4 θχ + ι, γ, θχ -1, Υ 4 θχ, Υ-1, θχ-1, Υ θχ + 1, Υ, θχ + Ι, Υ 4 θχ-Ι, Υ, θχ + Ι, Υ θχ, γ + ι, and that θχ, γ -ι ί θχ, γ + ι.
Ideally, a data symbol should meet the
Condition # 3.
Figure 31 illustrates an example of Condition # 3 where symbol A corresponds to symbol 3100. The symbols are arranged so that the phase by which the signal is multiplied
<img file="MX337079B_D0118.tif" />
precoded baseband z2 'of Figure 6 differs for symbol 3100, for both of its neighboring symbols in time domain 3101 and 3102, and for both of its neighboring symbols in frequency domain 3102 and 3104. Accordingly, despite the degradation of the received signal quality of symbol 3100 to the receiver, a good signal quality in neighboring signals is highly likely, thus ensuring good signal quality after error correction.
Figure 32 illustrates a symbol arrangement obtained by phase changes under these conditions.
As is evident from Figure 32, with respect to any data symbol, a different phase shift is applied to each neighboring symbol in the time domain and in the frequency domain. By itself, it can improve the ability of the receiving device to correct errors.
In other words, in Figure 32, when all neighboring symbols in the time domain are data symbols, Condition # 1 is met on all X and all Y.
Similarly, in Figure 32, when all of the neighboring symbols in the frequency domain are data symbols, Condition # 2 is met on all X's and all Y's.
Similarly, in Figure 32, when all of the neighboring symbols in the frequency domain are data symbols and all of the neighboring symbols in the time domain are data symbols, Condition # 3 is met on all X's and all the Y.
The following describes an example in which a phase change is performed on two precoded baseband signals, communication or explained in Mode 2 (see Figure 26).
When a phase change is made to the precoded baseband signal zl 'and the precoded baseband signal z2' as shown in Figure 26, various phase change schemes are possible. The details thereof are explained below.
Scheme 1 involves a phase change performed on the precoded baseband signal z2 'as already described, to achieve the phase change illustrated in Figure 32. In Figure 32, a phase change having a period (cycle) of 10 to the precoded baseband signal z2 '. However, as already described, in order to meet Conditions # 1, # 2 and # 3, the phase change applied to the precoded baseband signal z2 'in each (sub-) carrier varies over time. (Although such changes are applied in Figure 32 with a period (cycle) of ten, other phase change schemes are also possible.) Then, as shown in FIG. 33, the phase change performed on the precoded baseband signal zl 'produces a constant value ie one tenth of the value of the phase change performed on the precoded baseband signal z2'. In Figure 33, for a period (cycle) (of phase change performed on the precoded baseband signal z2 ') that includes the time-date $ 1, the value of the phase change performed on the band signal
I
I lL
I).
not:·
<img file="MX337079B_D0119.tif" />
precoded base zl 'is e ^<sup>0</sup>. Then, for the next period (cycle) (of the phase change performed on the precoded baseband signal z2 ') that includes the time-date $ 2, the value of the phase change performed on the precoded baseband signal zl' is and<sup>jD / 9</sup>, etc.
The symbols illustrated in figure 33 are indicated as e<sup>J</sup>°, for example. This means that this symbol is the signal zl 'in figure 26 in which a phase change has been applied through multiplication by e<sup>J</sup>°. That is, the values indicated in figure 33 for each of the symbols are the values of zl '(t) = y<sub>2</sub>(t) zl '(t) described in Mode 2 for yi (t).
As shown in figure 33, the phase change made in the precoded baseband signal zl 'produces a constant value, i.e. one tenth of the value of the phase change made in the precoded baseband signal z2', so that the value of the post-phase change varies with the quantity of each period (cycle). (As already described, in figure 33, the value is e<sup>J</sup>° for the first period (cycle), e ^<sup>9</sup> for the second period (cycle), etc.).
As already described, the phase change made in the precoded baseband signal z2 'has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten taking into account the phase change applied to the precoded baseband signal zl 'and the signal «JM
<img file="MX337079B_D0120.tif" />
z2 'precoded baseband. Accordingly, the reception quality of the data for the receiving device can change.
Scheme 2 involves a phase change of the precoded baseband signal z2 'as already described, to achieve the phase change illustrated in Figure 32. In Figure 32, a phase change with a period (cycle) is applied. ) of ten to the precoded baseband signal z2 '. However, as already described, in order to meet Conditions # 1, # 2 and # 3, the phase change applied to the precoded baseband signal z2 'in each (sub-) carrier varies over time. (Although such changes are applied in Figure 32 with a period (cycle) of ten, other phase change schemes are also possible.) Then, as shown in Figure 30, the phase change made in the precoded baseband signal zl 'differs from that made in the precoded baseband signal z2', as it has a period (cycle) of three instead of ten.
The symbols illustrated in Figure 30 are indicated as e ^ °, for example. This means that this symbol is the signal zl 'in figure 26 to which a phase change has been applied through multiplication by e ^ °. That is, the values indicated in figure 3 0 for each of the symbols are the values of zl (t) = yi (t) zl '(t) described in Mode 2 for yi (t).
As already described, the phase change made
<img file="MX337079B_D0121.tif" />
in the precoded baseband signal z2 'has ~ AUi ~ pexid £ L (cycle) of ten, but taking into account the phase changes applied to the precoded baseband signal zl' and the precoded baseband signal z2 ', the period (cycle) can be effectively equivalent to 30 for both precoded baseband signals zl 'and z2'. Accordingly, the quality of data reception can improve for the receiving device. An effective way to apply scheme 2 is to perform a phase change on the precoded baseband signal zl 'with a period (cycle) of N and to perform a phase change on the precoded baseband signal z2' with a period ( cycle) of M such that N and M are co-primes. By itself, taking into account both precoded baseband signals zl 'and z2', a period (cycle) of NxM can easily be achieved, effectively making the period (cycle) longer when N and M are co-primes.
The foregoing describes an example of the phase change scheme relevant to Mode 3. The present invention is not limited in that regard. As explained in relation to Modalities 1 and 2, a phase change can be made with respect to the frequency domain or the time domain or, in the time-frequency blocks. A similar improvement can be obtained in the quality of reception of the data for the receiving device in all cases.
The same is true for frames that have a different configuration from that described above, where
<img file="MX337079B_D0122.tif" />
They insert pilot symbols (SP (English acronym ™ '^ a ^ a- ~ pii £ itxa. scattered)) and symbols that transmit control information between data symbols. The phase change details in such circumstances are as follows.
Figures 47A and 47B illustrate the frame configuration of the modulated signals (the precoded baseband signals) zlozl 'and z2' in the time-frequency domain. Figure 47A illustrates the frame configuration of the modulated signal (the precoded baseband signals) zl or zl 'while Figure 47B illustrates the frame configuration of the modulated signal (the precoded baseband signals) z2'. In Figures 4 7A and 47B, 4701 marks the pilot symbols while 4702 marks the data symbols. Data symbols 4702 are symbols on which pre-modifying or pre-modifying and a phase change have been performed.
Figures 47A and 47B, like Figure 6, indicate the arrangement of symbols when a phase change is applied to the precoded baseband signal z2 '(although no phase change is made to the precoded baseband signal zl ). (Although Figure 6 illustrates a phase change with respect to the time domain, changing time t by carrier f in Figure 6 corresponds to a phase change with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is time and f is frequency corresponds to making a phase change in the time-frequency blocks). Therefore,
<img file="MX337079B_D0123.tif" />
the numerical values indicated in Figures 4 7A and 4 7B for each of the symbols are the values of the precoded baseband signal z2 'after the phase change. No value is given for the precoded baseband signal symbols zl '(zl), since no changes are made to them.
The key point of Figures 47A and 4 7B is that the phase change is performed on the data symbols of the precoded baseband signal z2 ', that is, on the precoded symbols. (The symbols under discussion, being precoded, actually include both symbols si and s2). Consequently, no phase change is made to the pilot symbols inserted in z2 '.
Figures 48A and 48B illustrate the frame configuration of the modulated signals (the precoded baseband signals) zlo zl 'and z2' in the time-frequency domain. Figure 48A illustrates the frame configuration of the modulated signal (the precoded baseband signals) zl or zl 'while Figure 47B illustrates the frame configuration of the modulated signal (the precoded baseband signals) z2'. In Figures 48A and 48B, 4701 marks the pilot symbols while 4702 marks the data symbols. Data symbols 4702 are symbols on which precoding or precoding and a phase change have been performed.
Figures 48A and 48B, like Figure 26, indicate the arrangement of symbols when a phase change is applied to the precoded baseband signal zl 'and the baseband signal
<img file="MX337079B_D0124.tif" />
precoded z2 '. (Although Figure 26 illustrates ^ n-aamb-io-de-fas®. With respect to the time domain, the change in time t by carrier f in Figure 26 corresponds to a phase change with respect to the frequency domain In other words, replacing (t) with (t, f) where t is time and f is frequency corresponds to making a phase change in the time-frequency blocks). Accordingly, the numerical values indicated in Figures 48A and 48B for each of the symbols are the values of the precoded baseband signal zl 'and z2' after the phase change.
The key point of Fig. 47 is that a phase change is made in the data symbols of the precoded baseband signal zl ', i.e. in the precoded symbols thereof, and in the signal data symbols precoded baseband z2 ', that is, in the precoded symbols thereof. (The symbols under discussion, being precoded, actually include both symbols si and s2). Accordingly, no phase change is made to the pilot symbols inserted in zl ', nor to the pilot symbols inserted in z2'.
Figures 4 9A and 49B illustrate the frame configuration of the modulated signals (the precoded baseband signals) zlo zl 'and z2' in the time-frequency domain. Figure 49A illustrates the frame configuration of the modulated signal (the precoded baseband signals) zl or zl ', while Figure 4 9B illustrates the frame configuration of the modulated signal (the precoded baseband signal) z2 '. In the ri 'i' ;?
Figures 49A and 49B, 4701 marks the pilot symbols, 4702 marks the data symbols, and 4901 marks the null symbols for which the in-phase component of the baseband signal I = 0 and the quadrature component Q = 0. As As such, data symbols 4702 are symbols on which precoding or precoding and phase change have been performed. Figures 49A and 49B differ from Figures 47A and 47B in the configurable scheme for symbols other than data symbols. The times and carriers at which the pilot symbols are inserted into the modulated signal zl 'are null symbols in the modulated signal z2'.
Conversely, the times and carriers at which the pilot symbols are inserted into the modulated signal z2 'are null symbols in the modulated signal zl ·'.
Figures 4 9A and 4 9B, like Figure 6, indicate the arrangement of symbols when a phase change is applied to the precoded baseband signal z2 '(although no phase change is made to the baseband signal precoded zl). (Although Figure 6 illustrates a phase change with respect to the time domain, the change in time t by carrier f in Figure 6 corresponds to a phase change with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is the time and f is the frequency corresponds to making a phase change in the time-frequency blocks). Therefore, the numerical values indicated in Figures 49A and 49B for each of the symbols are the baseband signal values.
<img file="MX337079B_D0125.tif" />
z2 'precoded after a phase change. No value is given for the precoded baseband signal symbols zl '(zl), since no phase change is performed on them.
The key point of Figures 49A and 49B is that a phase change is performed on the data symbols of the precoded baseband signal z2 ', that is, on the precoded symbols. (The symbols under discussion, being precoded, actually include both symbols si and s2). Consequently, no phase change is made to the pilot symbols inserted in z2 '.
Figures 50A and 50B illustrate the frame configuration of the modulated signals (the precoded baseband signals) zl or zl 'and z2' in the time-frequency domain. Figure 50A illustrates the frame configuration of the modulated signal (the precoded baseband signal) zl or zl ', while Figure 50B illustrates the frame configuration of the modulated signal (the precoded baseband signal) z2' . In Figures 50A and 50B, 4701 marks the pilot symbols, 4702 marks the data symbols, and 4901 marks the null symbols for which the in-phase component of the baseband signal I = 0 and the quadrature component Q = 0 As such, data symbols 4702 are symbols on which precoding or precoding and a phase change have been performed. Figures 50A and 50B differ from Figures 48A and 48B in the configurable scheme for
<img file="MX337079B_D0126.tif" />
symbols different from data symbols. Ws ”temples ~ Y ~ Tas' '™ ~'<sup>TO</sup> carriers in which the pilot symbols are inserted into the modulated signal zl 'are null symbols in the modulated signal z2'.
Conversely, the times and carriers when the pilot symbols are inserted into the modulated signal z2 'are null symbols in the modulated signal zl'.
Figures 50A and 50B, like Figure 26, indicate the arrangement of symbols when a phase change is applied to the precoded baseband signal zl 'and the precoded baseband signal z2'. (Although Figure 26 illustrates a phase change with respect to the time domain, the change in time t by carrier f in Figure 26 corresponds to a phase change with respect to the frequency domain. In other words, replacing (t) with (t, f) where t is the time and f is the frequency corresponds to making a phase change in the time-frequency blocks).
Accordingly, the numerical values indicated in Figures 50A and 50B for each of the symbols are the values of the precoded baseband signal zl 'and z2' after a phase change.
The key point of Figures 50A and 50B is that a phase change is made in the data symbols of the precoded baseband signal zl ', that is, in the precoded symbols thereof, and in the data symbols of the precoded baseband signal z2 ', that is, in the precoded symbols thereof. (The symbols under discussion, being precoded, actually include both symbols si and s2). Therefore, no
<img file="MX337079B_D0127.tif" />
no phase change is made to the pilot symbols grafted to zl ', nor to the pilot symbols inserted to z2'.
Figure 51 illustrates an exemplary configuration of a transmission device that generates and transmits the modulated signal having the frame configuration of Figures 4 7A, 47B, 49A and 49B. The components thereof perform the same operations as those in Figure 4 and use the same reference symbols.
In Figure 51, the weighting units 308A and 308B and the phase changer 317B only operate at the times indicated by the frame setting signal 313 as corresponding to the data symbols.
In FIG. 51, a pilot symbol generator 5101 (which also generates null symbols) outputs baseband signals 5102A and 5102B for a pilot symbol each time frame configuration signal 313 indicates a pilot symbol (or a symbol null).
Although not indicated in the frame configurations of Figures 47A to 50B, when precoding (or phase rotation) is not performed, e.g. ex. when a modulated signal is transmitted using only one antenna (so that the other antenna does not transmit any signal) or when a space-time coding transmission scheme (in particular, space-time block coding) is used to transmit control information symbols, frame configuration signal 313] Ν
100
<img file="MX337079B_D0128.tif" />
takes the symbols from control information 5104 and control information 5103 as input. When frame configuration signal 313 indicates a control information symbol, baseband signals 5102A and 5102B are output therefrom.
Wireless units 310A and 310B of FIG. 51 take multiple baseband signals as input and select a desired baseband signal according to frame configuration signal 313. Wireless units 310A and 310B then apply OFDM signal processing and output modulated signals 311A and 311B in accordance with the frame configuration.
Figure 52 illustrates an exemplary configuration of a transmission device that generates and transmits the modulated signal having the frame configuration of Figures 48A, 48B, 50A, and 50B. The components thereof perform the same operations as in Figures 4 and 51 and use the same reference symbols. FIG. 51 presents an additional phase changer 317A that only operates when frame configuration signal 313 indicates a data symbol. At all other times, the operations are identical to those explained with respect to figure 51.
Figure 53 illustrates an exemplary configuration of a transmission device that differs from that of Figure 51. The following describes the points of difference.
101
<img file="MX337079B_D0129.tif" />
As shown in figure 53, the change<sub>1</sub>QX..de-'f'ase-<sup>-</sup>5T7B ^ ™ ”takes multiple baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 317B performs a phase change on the precoded baseband signal 316B. When frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, phase changer 317B pauses phase change operations, so that the output symbols are output. the baseband signal as is. (This can be interpreted as performing the forced rotation corresponding to e<sup>5</sup>°) .
A selector 53 01 takes the multiple baseband signals as the input and selects a baseband signal that has a symbol indicated by the frame configuration signal 313 for the output.
Figure 54 illustrates an exemplary configuration of a transmission device that differs from that of Figure 52. The following describes the points of difference. As shown in Figure 54, the phase shifter 317B takes multiple baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 317B performs a phase change on the precoded baseband signal 316B. When the frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, the phase changer 317B pauses iíi nninm
<img file="MX337079B_D0130.tif" />
phase change operations so that the baseband signal symbols are output as they are. (This can be interpreted as performing the forced rotation corresponding to e<sup>j0</sup>) .
Similarly, as shown in Figure 54, phase changer 5201 takes multiple baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 5201 performs a phase change on the precoded baseband signal 3 09A. When frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, phase changer 52 01 pauses phase change operations such that output symbols are output. the baseband signal as is. (This can be interpreted as performing the forced rotation corresponding to e ^<sup>0</sup>) .
The foregoing explanations are given using pilot symbols, control symbols, and data symbols as examples. However, the present invention is not limited in that regard. When symbols are transmitted using different precoding schemes, such as single antenna transmission or transmission using space-time block coding, it is important not to perform a phase change. Conversely, performing a phase change on symbols that have been precoded is the key point of the present invention.
Therefore, a characteristic function of the
103
<img file="MX337079B_D0131.tif" />
The present invention is that phase change is not performed on all symbols within the frame configuration in the time-frequency domain, but is only performed on signals that have been precoded.
Mode 4
Modalities 1 and 2, described above, explain a regular phase change. Mode 3, however, describes the mode of a different phase change in neighboring symbols.
The present embodiment describes a phase change scheme that varies according to the modulation scheme and the coding rate of the error correction codes used by the transmission device.
Table 1 below is a list of phase shift scheme settings corresponding to the transmission device settings and parameters.
104
Table i
<img file="MX337079B_D0132.tif" />
<td>Number of signals modulated from transmission</td><td>Schedule of modulation</td><td>Cup of coding</td><td>Change pattern phase</td>
<td> 2</td><td># 1: QPSK, # 2: QPSK</td><td> #1: 1/2, #2 2/3</td><td># 1: # 2: A</td>
<td> 2</td><td># 1: QPSK, # 2: QPSK</td><td> #1: 1/2, #2: 3/4</td><td># 1: A, # 2: B</td>
<td> 2</td><td># 1: QPSK, # 2: QPSK</td><td> #1: 2/3, #2: 3/5</td><td># 1: A, # 2: C</td>
<td> 2</td><td># 1: QPSK, # 2: QPSK</td><td> #1: 2/3, #2: 2/3</td><td># 1: C, # 2: -</td>
<td> 2</td><td># 1: QPSK, # 2: QPSK</td><td> #1: 3/3, #2: 2/3</td><td># 1: D, # 2: E</td>
<td> 2</td><td># 1: QPSK, # 2: 16-QAM</td><td> #1: 1/2, #2: 2/3</td><td># 1: B, # 2: A</td>
<td> 2</td><td># 1: QPSK, # 2: 16-QAM</td><td> #1: 1/2, #2: 3/4</td><td># 1: A, # 2: C</td>
<td> 2</td><td># 1: QPSK, # 2: 16-QAM</td><td> #1: 1/2, #2: 3/5</td><td># 1: -, # 2: E</td>
<td> 2</td><td># 1: QPSK, # 2: 16-QAM</td><td> #1: 2/3, #2: 3/4</td><td># 1: D, # 2: -</td>
<img file="MX337079B_D0133.tif" />
105
<img file="MX337079B_D0134.tif" />
<td> 2</td><td># 1: QPSK, # 2: 16-QAM</td><td> #1: 2/3, #2: 5/6</td><td># 1: D, # 2: B</td>
<td> 2</td><td># 1: 16-QAM, # 2: 16-QAM</td><td> #1: 1/2, #2: 2/3</td><td># 1: -, # 2: E</td>
<td> •</td><td></td><td> •</td><td> •</td>
In Table 1, # 1 denotes the modulated si signal of Mode 1 described above (the baseband signal si modulated with the modulation scheme established by the transmission device) and # 2 denotes the modulated signal s2 (the s2 baseband modulated with the modulation scheme established by the transmission device). The encoding rate column in Table 1 indicates the encoding rate of the error correction codes for modulation schemes # 1 and # 2. The phase shift pattern column in Table 1 indicates the phase shift scheme applied to the precoded baseband signals zl (zl ') and z2 (z2<sup>r</sup>), as explained in Modalities 1 to 3. Although the phase change patterns are labeled A, B, C, D, E, etc., this refers to the applied degree of phase change, for example in a phase change pattern given by the preceding Mathematics 46 (formula 46) and Mathematics 47 (formula 47). In the phase change pattern column of Table 1, the hyphen means that it does not apply
106
<img file="MX337079B_D0135.tif" />
no phase change.
The modulation scheme and encoding rate combinations listed in Table 1 are examples. Other modulation schemes (such as 128-QAM and 256-QAM) and encoding rates (such as 7/8) not listed in Table 1 may also be included. In addition, as described in Mode 1, the error correction codes used for si and s2 may differ (Table 1 is given for cases where only one type of error correction code is used, as in Figure 4). Also, the same modulation scheme and coding rate can be used with different phase shift patterns. The transmitting device transmits information indicative of phase change patterns to the receiving device. The receiving device specifies the phase shift pattern by cross-referencing the information and Table 1, then performs demodulation and decoding. When the modulation scheme and error correction scheme determine a single phase shift pattern, while the transmitting device transmits the modulation scheme and the information regarding the error correction scheme, the receiving device knows the pattern phase change obtaining that information. In itself, the information relevant to the phase change pattern is not strictly necessary.
In modes 1 to 3, the phase shift is applied to the precoded baseband signals. However, also
107 the amplitude can be modified together with the phaseja._f periodic and regular changes. Accordingly, an amplification modification pattern can also be made that regularly modifies the amplitude of the modulated signals to conform to Table 1. In such circumstances, the transmitting device must include an amplification modifier that modifies the amplification after the weighting unit unit 308A or the weighting unit 308B of Figure 3 or 4. Furthermore, the amplification modification can be performed only on one or both of the precoded baseband signals zl (t) and z2 (t) (in the first case, the amplification modifier is only necessary after the weighting unit 308A and 308B).
Also, although not indicated in Table 1 above, also the correlation scheme can be regularly modified by the correlator, without a regular phase change.
That is, when the correlation scheme for the modulated signal sl (t) is 16-QAM and the correlation scheme for the modulated signal s2 (t) is also 16-QAM, the correlation scheme applied to the modulated signal s2 ( t) can change regularly as follows: from 16-QAM to 16-APSK, to 16-QAM in the IQ plane, to a first correlation scheme that produces a different signal point design from 16-APSK, to 16- QAM in the IQ plane, to a second mapping scheme that produces a point design
108
<img file="MX337079B_D0136.tif" />
signal different from 16-APSK, etc. In this way, you can improve the quality of data reception for the receiving device, much like the results obtained by a regular phase change described above.
Furthermore, the present invention can use any combination of schemes for a regular phase change, correlation scheme, and amplitude, and the transmit signal can transmit with all of this in mind.
The present embodiment can be carried out using single carrier schemes as well as multi-carrier schemes. Accordingly, the present embodiment may also be effected using, for example, spread spectrum communications, OFDM, SC-FDM, SC-OFDM, OFDM of mini-waves as described in Non-Patent Bibliography 7, etc. As already described, the present modality sets out to change the phase, amplitude and correlation schemes making modifications of phase, amplitude and correlation scheme with respect to the time domain t. However, much like Mode 1, the same changes can be made with respect to the frequency domain. That is, considering the modification of the phase, amplitude and correlation scheme in the time domain t that is described in the present modality and replacing t with f (where f is the frequency ((sub-) carrier)) leads to the modification. phase, amplitude and correlation scheme applicable to the frequency domain. In addition, the modification of
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<img file="MX337079B_D0137.tif" />
The phase, amplitude, correlation scheme of the present embodiment is also applicable to the modification of the phase, amplitude, correlation scheme both in the time domain and in the frequency domain.
Also, in the present embodiment, symbols other than data symbols, such as pilot symbols (preamble, single word, etc.) or symbols that transmit control information, may be arranged within the frame in any way.
Al Mode
The present embodiment describes a scheme for regularly changing the phase when encoding is performed using block codes as described in Non-patent Bibliography 12 15, such as LDPC QC (quasi-cyclical) codes (may be used not only QC-LDPC codes but also LDPC codes), concatenated LDPC and BCH codes (Bose-Chaudhuri-Hocquenghem), turbo codes or duo-binary turbo codes using tail bits, etc. The following example considers a case where two currents si and s2 are transmitted. However, when the encoding has been performed using block codes and no control information or the like is required, the number of bits that make up each coded block matches the number of bits that make up each block code (may be included anyway the control information etc. described below). When the coding has been done using block codes or the like and information
110
<img file="MX337079B_D0138.tif" />
MEXICAN INSTITUTE. FROM PROPERTY V IND '.' S Control TS.IAL or the like (for example, the CRC (cyclic redundancy check) transmission paxáxrxxs ^ ---. Is required), then the number of bits that make up each encoded block it is the sum of the number of bits that make up the block codes and the number of bits that make up the information.
Figure 34 illustrates the varying amounts of symbols and ranges required in each coded block when using block codes. Figure 34 illustrates the varying amounts of symbols and ranges required in each coded block when using block codes if, for example, two currents si and s2 are transmitted as indicated by the transmission device of Figure 4, and the device Transmission has only one encoder. (Here, the transmission scheme can be any single carrier scheme or multi-carrier scheme, such as OFDM).
As shown in Figure 34, when using block codes, there are 6000 bits that make up a single coded block. In order to transmit those 6000 bits, the number of symbols required depends on the modulation scheme, which is 3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for
64-QAM.
Then, since the transmitting device of Figure 4 transmits two currents simultaneously, 1500 of the above mentioned 3000 needed symbols are assigned if the modulation scheme is QPSK and assigned
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<img file="MX337079B_D0139.tif" />
to s2 the other 1500 symbols. Thus, 1500 slots are required to transmit the 1500 symbols (hereinafter, slots) for each of si and s2.
With the same reasoning, when the modulation scheme is 16-QAM, it takes 750 slots to transmit all the bits that make up a single coded block, and when the modulation scheme is 64-QAM, it takes 500 slots to transmit all the bits. bits that make up a single coded block.
The following describes the relationship between the previously defined intervals and the multiplication phase, as pertinent to the schemes for a regular phase change.
Here, five different phase shift values (or phase shift sets) are assumed as prepared for use in the scheme for a regular phase shift. That is, five different phase change values (or phase change sets) have been prepared for the phase changer of the transmission device of figure 4 (equivalent to the period (cycle) of Modes 1 to 4) (as In Figure 6, five phase shift values are needed to perform a phase shift with a period (cycle) of five on the precoded baseband signal z2 'only. Furthermore, as in Figure 26, two phase shift values are needed for each interval in order to perform phase shift on both precoded baseband signals zl 'and z2'. Those two phase shift values are called the set
112 ι
<img file="MX337079B_D0140.tif" />
phase change. Therefore, five phase change sets should ideally be prepared in order to perform the phase change with a period (cycle) of five in such circumstances. Those five phase shift values (or phase shift sets) are expressed as PHASE [0], PHASE [1], PHASE [2], PHASE [3], and PHASE [4].
As for the previously described 1500 intervals necessary to transmit the 6000 bits that constitute a single encoded block when the modulation scheme is QPSK, PHASE [0] is used in the 300 intervals, it is used
PHASE [1] in the 300 intervals, PHASE [2] is used in the 300 intervals, PHASE [3] is used in the 300 intervals, and is used
PHASE [4] at all 300 intervals. This is due to the fact that any deviation in the use of phases causes a great influence that the most frequently used phase must exert, since the receiving device depends on such influence in relation to the quality of data reception.
Similarly, for the above described 700 intervals necessary to transmit the 6000 bits that constitute a single coded block when the modulation scheme is 16-QAM, PHASE [0] is used in all 150 intervals, PHASE [1] is used in At 150 intervals, PHASE [2] is used at all 150 intervals, PHASE [3] is used at all 150 intervals, and PHASE [4] is used at all 150 intervals.
Likewise, for the previously described 500
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<img file="MX337079B_D0141.tif" />
intervals necessary to transmit the 6000 bits that constitute a single encoded block when the modulation scheme is 64-QAM, PHASE [0] is used in the 100 intervals, PHASE [1] is used in the 100 intervals, PHASE [2 is used ] in the 100 intervals, PHASE [3] is used in the 100 intervals, and
PHASE [4] in the 100 intervals.
As already described, a scheme for a regular phase change requires the preparation of N phase change values (or phase change sets) (where the N different phases are expressed as PHASE [0], PHASE [1] , PHASE [2] ...
PHASE [N-2], PHASE [Nl]). Thus, in order to transmit all the bits that constitute a single encoded block, PHASE [0] is used in the K<sub>or</sub> intervals, PHASE [1] is used in the Κχ intervals, PHASE [i] is used in the Κχ intervals (where i = 0, 1, 2 ... N-1; that is, 0íi ^ Nl, i being an integer ), and PHASE [N-1] is used in the K<sub>N</sub>_i intervals, so that Condition # A01 is met.
(Condition # A01)
K<sub>or</sub> = Κχ ... = Κχ = ... Kjj-i. I mean, K<sub>to</sub> = K<sub>b</sub> (M. and where a, b, = 0, 1, 2 ... Nl, a / b).
Then, when a communications system that supports multiple modulation schemes selects such a supported modulation scheme to use, Condition # A01 is preferably met for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses
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<img file="MX337079B_D0142.tif" />
τ. ' > '-'RCñ'ÍEDAD' / ír'al symbols that transmit a different amount dabí-Uspor-s-íj (although it may happen that some use the same amount), Condition # A01 may not be met in some modulation schemes. In such a case, the following condition applies in place of Condition # A01.
(Condition # A02)
The difference between K<sub>to</sub> and K<sub>b</sub> meets 0 or 1. That is, | K<sub>to</sub> - K<sub>b</sub>| meets 0 or 1 (Va, Md, where a, b = 0, 1, 2 ... Nl, a / b)
Figure 35 illustrates the varying amounts of symbols and ranges required in two coded blocks when using block codes. Figure 35 illustrates the varying amounts of symbols and ranges required in each coded block when using block codes if, for example, two currents si and s2 are transmitted as indicated by the transmission device of Figure 3 and Figure 12, and the transmission device has two encoders. (Here, the transmission scheme can be any single carrier scheme or multi-carrier scheme, such as OFDM).
As shown in Figure 35, when using block codes, there are 6000 bits that make up a single coded block. In order to transmit those 6000 bits, the number of symbols required depends on the modulation scheme, which is 3000 symbols for QPSK, 1500 symbols for 16-QAM, and 1000 symbols for 64-QAM.
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<img file="MX337079B_D0143.tif" />
The transmitting device of the figure and the transmitting device of Figure 12 each transmit two streams at a time, and have two encoders. In themselves, the two streams each transmit different code blocks. Therefore, when the modulation scheme is QPSK, two coded blocks extracted from si and s2 are transmitted within the same interval, for example, first coded block extracted from si is transmitted, then a second coded block extracted from s2 is transmitted. So, 3,000 intervals are needed to transmit the first and second coded blocks.
With the same reasoning, when the 16-QAM modulation scheme, 1500 slots are needed to transmit all the bits that make up the two coded blocks and, when the modulation scheme is 64-QAM, 1000 slots are needed to transmit all the bits that make up the two coded blocks.
The following describes the relationship between the previously defined intervals and the multiplication phase, as pertinent to the schemes for a regular phase change.
Here, five different phase shift values (or phase shift sets) are assumed as prepared for use in the scheme for a regular phase shift. That is, five different phase change values (or phase change sets) have been prepared for the phase changers of the
116
<img file="MX337079B_D0144.tif" />
transmitting devices of Figures 3 and'TSMequivalent to the period (cycle) of Modalities 1 to 4) (as in Figure 6, five phase change values are needed to perform a phase change with a period (cycle) of five on the precoded baseband signal z2 'only In addition, as in Figure 26, two phase shift values are needed for each interval in order to perform the phase shift on both precoded baseband signals zl' and z2 '. Those two phase shift values are called the phase shift set. Therefore, five phase change sets should ideally prepare to perform the phase change with a period (cycle) of five under such circumstances. Those five phase shift values (or phase shift sets) are expressed as PHASE [0], PHASE [1], PHASE [2], PHASE [3], and PHASE [4].
For the previously described 3000 intervals necessary to transmit the 6000x2 bits that constitute a single encoded block when the modulation scheme is QPSK, PHASE [0] is used in the 600 intervals, PHASE [1] is used in the 600 intervals, it is used PHASE [2] in the 600 intervals, PHASE [3] is used in the 600 intervals, and PHASE [4] is used in the 600 intervals. This is due to the fact that any deviation in the use of phases causes a great influence that the most frequently used phase must exert, since the receiving device depends on such influence for the quality of reception of the data.
Also, in order to transmit the first block
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<img file="MX337079B_D0145.tif" />
PHASE [1] is used in 600 times intervals, PHASE [2] is used in 600 times intervals, PHASE [3] is used in 600 times intervals, and PHASE [4] is used in 600 intervals. times. Also, in order to transmit the second encoded block, PHASE [0] is used in the 600 times intervals, PHASE [1] is used in the 600 times intervals, PHASE [2] is used in the 600 times intervals, PHASE [3] is used in the 600 times intervals, and PHASE [4] is used in the 600 times intervals.
Similarly, for the above-described 1500 intervals necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation scheme is 16-QAM, PHASE [0] is used in all 300 intervals, PHASE [1] is used. in the 300 intervals, PHASE [2] is used in the 300 intervals, PHASE [3] is used in the 300 intervals, and PHASE [4] is used in the 300 intervals.
Also, in order to transmit the first encoded block, PHASE [0] is used in 300 times intervals, PHASE [1] is used in 300 times intervals, PHASE [2] is used in 300 times intervals, PHASE [3] is used in 300 times intervals, and PHASE [4] is used in 300 times intervals. Also, in order to transmit the second encoded block, PHASE [0] is used in the 300 times intervals, PHASE [1] is used in the 300 times intervals, PHASE [2] is used in the 300 times intervals. , PHASE [3] is used at intervals of 300 times, and ¿a »
<img file="MX337079B_D0146.tif" />
i 'YES'. 'INDUSTRIAL PROPERTY PHASE [4] is used at intervals of 3 00 times ... ™. .._
Similarly, for the above described 1000 intervals necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation scheme is 64-QAM, PHASE [0] is used in all 200 intervals, PHASE [1] is used in at 200 intervals, PHASE [2] is used at 200 intervals, PHASE [3] is used at 200 intervals, and PHASE [4] is used at 200 intervals.
Also, in order to transmit the first encoded block, PHASE [0] is used in the 200 times intervals, PHASE [1] is used in the 200 times intervals, PHASE [2] is used in the 200 times intervals, PHASE [3] is used in the 200 times intervals, and PHASE [4] is used in the 200 times intervals. Also, in order to transmit the second encoded block, PHASE [0] is used in the 200 times intervals, PHASE [1] is used in the 200 times intervals, PHASE [2] is used in the 200 times intervals, PHASE [3] is used in the 200 times intervals, and PHASE [4] is used in the 200 times intervals.
As already described, a scheme for regularly changing the phase requires the preparation of phase change values (or phase change sets) expressed as PHASE [0], PHASE [1], PHASE [2] ... PHASE [N-2], PHASE [Nl]. By itself, in order to transmit all the bits that constitute two encoded blocks, PHASE [0] is used in the K<sub>or</sub> intervals, PHASE [1] is used in the Ki intervals, PHASE [i] is used in the Κχ intervals (where i = 0,
<img file="MX337079B_D0147.tif" />
119
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1, 2..JSF — 1), and PHASE [N-1] is used in the K<sub>N</sub>_i intervals, so that Condition # A03 is met.
(Condition # A03)
K<sub>or</sub> = Ki ... = Ki = ... K<sub>N</sub>-i. I mean, K<sub>to</sub> = K<sub>b</sub> (Vá y \ fc> where a, b, = 0,1,2 ... N-1, a + b).
Also, in order to transmit all the bits that make up the first encoded block, PHASE [0] of K is used<sub>0</sub>, i times, PHASE [1] of Κι is used, ι times, PHASE [i] of Ki is used,<sub>x</sub> times (where i = 0, 1, 2 ... N-1), and PHASE [N-1] of K is used<sub>N</sub>_i, i times, so that Condition # A04 is met.
(Condition # A04)
Κο, ι <sup>=</sup> Κι, ι <sup>=</sup> ··· Ki, i = ··· Kn-i, i. I mean, K<sub>to</sub>, i = K<sub>b</sub>, i (and Mo where a, b, = 0, 1, 2 ... N-1, a / b).
Also, in order to transmit all the bits that make up the second encoded block, PHASE [0] of K is used<sub>0(2 </sub>times, PHASE [1] of Ki is used,<sub>2</sub> times, PHASE [i] of K ± is used,<sub>2</sub> times (where i = 0, 1, 2 ... N-1), and PHASE [N-1] of K is used<sub>n</sub>-i,<sub>2</sub> times, so Condition # A05 is met.
(Condition # A05)
Ko,<sub>2</sub> = Ki,<sub>2</sub> = ... K<sub>i (2</sub> = ... Kn-i, 2 · That is, K<sub>to</sub>,<sub>2</sub> = K<sub>b) 2</sub> (7á and Mo where a, b, = 0, 1, 2 ... N-1, a Ψ b).
Then, when a communications system that supports multiple modulation schemes selects such a supported modulation scheme to use, they must preferably meet Condition # A03, # A04, and # A05 for the modulation scheme
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<img file="MX337079B_D0149.tif" />
admitted.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols that transmit a different number of bits per symbol (although some may use the same amount), Conditions # A03, # A04 may not be met and # A05 in some modulation schemes. In such case, the following conditions apply in place of Condition # A03, # A04 and # A05.
(Condition # A06)
The difference between K<sub>to</sub> and K<sub>b</sub> meets 0 or 1. That is, | K<sub>to</sub> - K<sub>b</sub>| meets 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... Nl, a Ψ b) (Condition # A07)
The difference between K<sub>ajl</sub> and K<sub>b</sub>, i meets 0 or 1. That is, | K<sub>to</sub>, i - K<sub>b</sub>, i | meets 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... Nl, a * b) (Condition # A08)
The difference between K<sub>to</sub>,<sub>2</sub> and K<sub>b</sub>,<sub>2</sub> meets 0 or 1. That is, | Ka<sub>2</sub> - Kb, 2 | meets 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... Nl, a Ψ b)
As already described, the offset between the phases used to transmit the encoded blocks is eliminated creating a relationship between the encoded block and the multiplication phase. In this way, you can improve the quality of data reception for the receiving device.
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<img file="MX337079B_D0150.tif" />
In the present embodiment, N phase change values (or phase change sets) are needed to perform a phase change with a period (cycle) of N with the scheme for a regular phase change. So the N phase change values (or phase change sets) are prepared PHASE [0], PHASE [1], PHASE [2] ... PHASE [N-2], and PHASE [N-1 ]. However, there are schemes for reordering the phases in the order established with respect to the frequency domain. In this sense, no limitation is foreseen. The N phase change values (or phase change sets) can also change the phases of the blocks in the time domain or time-frequency domain to obtain a symbol arrangement as described in Mode 1. Although the preceding examples explain a phase change scheme with a period (cycle) of N, the same effects can be obtained using N phase change values (or phase change sets) in a random fashion. That is, the N phase change values (or phase change sets) are not always needed for a regular period (cycle). As long as the conditions described above are met, great improvements can be made in the quality of data reception for the receiving device.
Also, given the existence of modes for spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, space-time block encoding schemes, single stream transmission, and
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<img file="MX337079B_D0151.tif" />
schemes using a regular phase shift transmission described in Modes 1 to 4), the transmission device (broadcast station, base station) can select any of those transmission schemes.
As described in Non-Patent Bibliography 3, spatial multiplexing MIMO schemes involve the transmission of the si and s2 signals, which are correlated using a selected modulation scheme, on each of two different antennas. As described in Modes 1 through 4, MIMO schemes that use a fixed precoding matrix involve performing precoding only (without any phase change). In addition, space-time block coding schemes are described in Non-Patent Bibliography 9, 16, and 17. Single current transmission schemes involve transmitting the signal si, correlated with a selected modulation scheme, from an antenna after performing the predetermined processing.
Schemes using multi-carrier transmission such as OFDM involve a first group of carriers made up of multiple carriers and a second group of carriers made up of multiple different carriers from the first group of carriers, etc., so that multi-carrier transmission is performed with multiple groups of carriers. For each group of carriers, you can use any of the spatial multiplexing MIMO schemes, the MIMO schemes that
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<img file="MX337079B_D0152.tif" />
they use a fixed precoding matrix, space-time block coding schemes, single stream transmission, and schemes that use a regular phase shift. In particular, schemes using a regular phase change are preferably used in a selected group of (sub-) carriers to perform the present embodiment.
When a phase change is performed, then for example a phase change value is made for PHASE [i] of X radians on only a precoded baseband signal, the phase changers of Figures 3, 4, 5, 12 , 25, 29, 51 and 53 multiply the precoded baseband signal z2 'by e<sup>jX</sup>. Then, a phase change, for example, performed by a phase change set for PHASE [i] of X radians and Y radians, which is performed on both precoded baseband signals, the phase changers of Figures 26, 27, 28, 52 and 54 multiply the precoded baseband signal z2 'by e<sup>jX</sup> and multiply the precoded baseband signal zl 'by e<sup>jY</sup>.
B1 mode
The following describes an exemplary configuration of an application of the transmission and reception schemes explained in the modalities and a system using the application.
Figure 36 illustrates the configuration of a system that includes devices that execute the transmission and reception schemes described in the modalities
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precedents. As shown in figure TT7Tos ”devices that execute the transmission schemes and the reception schemes described in the preceding embodiments include various receivers such as a broadcast station, a television 3611, a DVD recorder 3612, an STB (decoder) 3613, a 3620 computer, a 3641 vehicle-installed television, a 3630 mobile phone, etc., within a 3600 digital broadcast system. Specifically, broadcast station 3 601 uses a transmission scheme explained in the previously described modes for transmitting multiplexed data, in which video, audio and other data is multiplexed by a predetermined transmission band.
The signals transmitted by the broadcast station 3601 are received by an antenna (such as the 3660 or 3640 antenna) incorporated within externally connected to each of the receiving devices. Each receiver obtains the multiplexed data using the reception schemes explained in the previously described modalities to remodulate the signals received by the antenna. Accordingly, the 3600 digital broadcast system can achieve the effects of the present invention, as explained in the previously described embodiments.
The video data included in the multiplexed data is encoded with a video encoding method in accordance with a standard such as MPEG-2 (moving picture expert group), MPEG4-AVC (acronym
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<img file="MX337079B_D0154.tif" />
in English for advanced video encoding), VC-1 or similar. The audio data included in the multiplexed data is encoded with an audio encoding method in accordance with a standard such as Dolby AC-3 (Audio Encoding), Dolby Digital Plus, MLP (Meridian Lossless Packing), DTS (Digital Theater Systems), DTS-HD, PCM (pulse code modulation) or the like.
Figure 37 illustrates the configuration of a 7900 receiver that executes a reception scheme set forth in the previously described modes. The receiver 3700 corresponds to a receiver included in one of the television 3611, the DVD recorder 3612, the STB 3613, the computer 3620, the vehicle-installed television 3641, the mobile phone 3630, etc., of figure 36. The receiver 3700 includes a tuner 3701 that converts a high frequency signal received by an antenna 3760 into a baseband signal, and a demodulator 37 02 that remodulates the baseband signal thus converted to obtain the multiplexed data. The demodulator 3702 executes a reception scheme explained in the previously described modalities and, in this way, achieves the effects of the present invention as already explained.
The receiver 3700 further includes a current interface 3720 that demultiplexes the audio and video data from the multiplexed data obtained by the 3702 demodulator, a 3704 signal processor that decodes the video data.
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obtained from the demultiplexed video data into a video signal by applying a corresponding video decoding method and decoding the audio data obtained from the demultiplexed audio data into an audio signal by applying a corresponding audio decoding method, a unit of audio output 3706 that outputs the decoded audio signal through a speaker or other similar device, and a 3707 video output unit that outputs the decoded video signals on a screen or similar device.
When, for example, a user uses a 3750 remote control, the information for a selected channel (selected program (television) or audio broadcast) is transmitted to an operation input unit 3710. Thereafter, the receiver 3700 performs the processing in the received signal, which the antenna 3760 receives, which includes the demodulation of the signal corresponding to the selected channel, the error correction decoding, etc., in order to obtain the received data. At this point, the receiver 3700 obtains the control symbol information including the information about the transmission scheme (the transmission scheme, the modulation scheme, the error correction scheme, etc. from the previously described modalities) ( as described using Figures 5 and 41) of the control symbols included in the signal corresponding to the selected channel. By itself, the 3700 receiver can correctly set the receive operations, the
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demodulation scheme, the correction scheme of errors etc., thus making it possible to obtain the data included in the data symbols transmitted by the broadcasting station (base station). Although the preceding description is given for an example where the user uses the 3750 remote, the same operations apply when the user presses a select key embedded in the 3700 receiver to select a channel.
In accordance with this configuration, the user can view the programs received by the 3700 receiver.
The receiver 3700 pertinent to the present embodiment further includes a unit 3708 which may be a magnetic disk, an optical disk, a non-volatile semiconductor memory, or a similar recording medium. The receiver 3700 stores the data included in the demultiplexed data obtained through demodulation by demodulator 3702 and error correction decoding (in some circumstances, data obtained through demodulation by demodulator 3702 may not be subject Error correction In addition, the 3700 receiver can perform additional processing after error correction. The same is valid from now on for similar statements concerning other components), the data corresponding to such data (for example, the data obtained through the compression of such data), the data obtained through audio and video processing , etc., in
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drive 3708. Here, an optical disc is a recording medium, such as DVD (Digital Versatile Disc) or BD (Blu-ray Disc), that is, readable and recordable using a laser beam. A magnetic disk is a floppy disk, a hard disk or similar recording medium on which information can be stored using magnetic flux to magnetize a magnetic body. A nonvolatile semiconductor memory is a recording medium, such as flash memory or ferroelectric random access memory, ka of an element or semiconductor elements. Specific examples of nonvolatile semiconductor memory include an SD card that uses flash memory and a Flash SSD (Solid State Drive). Naturally, the specific types of recording media mentioned here are merely examples. They can also use other types of recording media.
In accordance with this structure, the user can record and store the programs received by the receiver 3700 and thereby view the programs at a certain time after broadcast by reading the data recorded thereon from the medium.
Although the foregoing explanations describe receiver 3700 that stores multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding in unit 3708, a portion of the data included in the data may instead be extracted and recorded. multiplexed data. When data broadcast services or similar content are included along with audio data
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and video in the multiplexed data obtained through the demodulation performed by the 3702 demodulator and the error correction decoding, the audio and video data can be extracted from the multiplexed data remodulated by the 3702 demodulator and stored as new multiplexed data. Also, unit 3 708 can store the audio data or the video data included in the multiplexed data obtained through the demodulation carried out by the demodulator 3702 and the error correction decoding as new multiplexed data. The content of the aforementioned data broadcast service, included in the multiplexed data, can also be stored in unit 3708.
Also, when a television, the recording device (for example, a DVD recorder, BD recorder, HDD recorder, SD card, or the like) or the mobile phone incorporating the 3700 receiver of the present invention receive the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding that includes data to correct software defects used to operate the television or recording device, To correct software defects to prevent leaking of personal information and recorded data, etc., such software defects can be corrected by installing the data on the television or recording device. In this way, the defects of the 3700 receptor are corrected through the inclusion of the naitiáAiief
<img file="MX337079B_D0159.tif" />
Data to correct defects in the 3700 receiver software. Accordingly, the television, recording device, or mobile phone incorporating the 3700 receiver may be made to operate more reliably.
Here, for example, current interface 3703 performs the process to extract a portion of the data included in the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding. Specifically, stream interface 3703 demultiplexes the various data included in the multiplexed data remodulated by demodulator 3702, such as audio data, video data, content from the data broadcast service, etc., as instructed to do so by a non-diagrammed controller such as a CPU. The current interface 3703 then extracts and multiplexes only the indicated demultiplexed data, thereby generating new multiplexed data. The user can determine the data to be extracted from the demultiplexed data or it can be determined in advance according to the type of recording medium.
In accordance with such a structure, the receiver 37 00 can extract and record only the necessary data in order to view the recorded program. In this way, the amount of data to be recorded can be reduced.
Although the preceding explanation describes unit 3708 as storing the multiplexed data obtained through
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_ .... ...,. ί Γ Τ Ί.-ΐ-'Λΐϊ - ^ / Τ 'of the demodulation carried out by the demodulator 3 7 02 and the error correction decoding, the video data included in the Multiplexed data thus obtained can be converted using a different video encoding method from the original video encoding method applied to it, thereby reducing the amount of data or the bit rate thereof. Unit 3708 can then store the converted video data as new multiplexed data. Here, the video encoding method used to generate the new video data may be in accordance with a different standard than the one used to generate the original video data. Alternatively, the same video encoding method can be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding can be converted using an audio encoding method different from the original audio encoding method applied to the same . The 3708 unit can also store the converted audio data as the new multiplexed data.
Here, for example, the current interface 3703 or the signal processor 3704 performs the process by which the audio or video data included in the multiplexed data obtained through the demodulation performed by the 3702 demodulator and the correction decoding of mistakes, they become
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as to reduce the amount of data or the bit rate of the same. Specifically, stream interface 3703 demultiplexes the various data included in the demodulator multiplexed data by demodulator 3702, such as audio data, video data, the content of the data broadcast service, etc. , as instructed to do so by a diagrammed controller such as a CPU. Signal processor 3704 then performs the processing to convert the thus demultiplexed video data using a different video encoding method from the original video encoding method applied thereto, and performs the processing to convert the thus demultiplexed .audio data using a different video encoding method than the original audio encoding method applied to them. As instructed to do so by the controller, the current interface 3703 then multiplexes the converted audio and video data, thereby generating new multiplexed data. Signal processor 3704, in accordance with the instructions of the controller, can perform conversion processing on either the video data or the audio data alone, or it can perform conversion processing on both types of data. Furthermore, the user can specify or determine in advance the amounts of video and audio data or the bit rate thereof to be obtained by the conversion, according to the type of recording medium.
In accordance with such a structure, the 3700 receiver can
<img file="MX337079B_D0162.tif" />
modify the amount of data or the speed ^ é ~ BTC ^ de'Ttors' "<sup>1</sup>audio and video dartoe 'for storage according to the data storage capacity of the recording medium, or according to the data reading or writing speed of the 3708 unit. Therefore, the programs can be stored in the unit even though the storage capacity of the recording medium is less than the amount of multiplexed data obtained through the demodulation carried out by the demodulator 3702 and the error correction decoding or that the read or write speed of the data is less than the bit rate of the demultiplexed data obtained through the demodulation carried out by the 3702 demodulator. By itself, the user can watch the programs at any given time after broadcast by reading the recorded data.
Receiver 3700 further includes a current output interface 3709 that transmits multiplexed data, demultiplexed by demodulator 3702 to external devices through a 3730 communication medium. The current output interface 3709 may be, for example, a wireless communications device that transmits the modulated multiplexed data to an external device using a wireless transmission scheme in accordance with a wireless communications standard such as Wi-Fi ™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.llg, IEEE 802.11η, etc.)., WiGig, WirelessHD, Bluetooth, ZigBee, etc., via wireless medium
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(corresponding to the communications medium. 3709 power output can also be a wired communications device that transmits the modulated multiplexed data to an external device using a communications scheme in accordance with a wired communications standard such as Ethernet ™, USB (Universal Serial Bus). ), PLC (Power Line Communication), HDMI (Random Definition Multimedia Interface), etc. , through a wired transmission path (corresponding to the communication medium 3730) connected to the current output interface 3709.
According to this configuration, the user can use an external device with the multiplexed data received by the receiver 3700 using the reception scheme exposed in the previously described modalities. The use of the multiplexed data by the user here includes the use of the multiplexed data to view in real time on an external device, record the multiplexed data with a recording unit included in an external device, and transmit the multiplexed data from an external device. to even another external device.
Although the preceding explanations describe the receiver 3700 that outputs the multiplexed data obtained through the demodulation performed by the 3702 demodulator and the error correction decoding by means of the interface
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output signal 3709, instead, a portion of the data included in the multiplexed data can be output and output. For example, when content from the data broadcast service or the like is included together with the audio and video data in the multiplexed data obtained through the demodulation carried out by the 3702 demodulator and the error correction decoding, the audio data and video can be extracted from the multiplexed data obtained through the demodulation carried out by the 3702 demodulator and the error correction decoding, and 3709 current output interface can multiplex them and output them as new multiplexed data. Furthermore, the current output interface 3709 can store any of the audio data or video data included in the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding as new multiplexed data.
Here, for example, current interface 3703 performs the process to extract a portion of the data included in the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding. Specifically, stream interface 3703 demultiplexes the various data included in the multiplexed data demodulated by demodulator 3702, such as audio data, video data, content of the
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data dissemination, etc. , as instructed to do so by a diagrammed controller such as a CPU. The current interface 3 703 then extracts and multiplexes only the indicated demultiplexed data, thereby generating new multiplexed data. The user can determine the data to be extracted from the demultiplexed data or it can be determined in advance according to the type of current output interface 3709.
In accordance with this structure, the receiver 37 00 can extract and output only the required data to an external device. In this way, less multiplexed data is output using less communication bandwidth.
Although the preceding explanation describes the current output interface 3709 as outputting the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding, the video data included in the multiplexed data thus obtained may converted using a different video encoding method than the original video encoding method applied to them, as to reduce the amount of data or the bit rate of the same. The current output interface 3709 can then output the converted video data, such as new multiplexed data. Here, the video encoding method used to generate the new video data may be in accordance with a different standard than the one used to generate the original video data. As an alternative,
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The same video encoding method can be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding can be converted using an audio encoding method different from the original audio encoding method applied to the same, as to reduce the amount of data or the bit rate of the same. The current output interface 3709 can then output the converted audio data as new multiplexed data.
Here, for example, the current interface 3703 or the signal processor 3704 carry out the process by which the audio or video data included in the multiplexed data obtained through the demodulation carried out by the demodulator 3702 and the decoding are converted. error correction to reduce the amount of data or the bit rate of the same. Specifically, stream interface 3703 demultiplexes the various data included in the multiplexed data demodulated by demodulator 3702, such as audio data, video data, the content of the data broadcast service, etc., as instructed to make it a diagrammed controller. Signal processor 3704 then performs the processing to convert the thus demultiplexed video data using a different video encoding method from the original applied video encoding method.
<img file="MX337079B_D0168.tif" />
to them, and performs the processing for audio co-V'éT't'ii 'so demultiplexed using a video encoding method different from the original audio encoding method applied thereto. As instructed to do so by the controller, the current interface 3703 multiplexes the converted audio and video data, thereby generating new multiplexed data. Signal processor 3704, in accordance with the instructions of the controller, can perform conversion processing on either the video data or the audio data alone, or it can perform conversion processing on both types of data. Furthermore, the amounts of video and audio data or the bit rate thereof to be obtained by conversion can be specified by the user or determined in advance according to the type of current output interface 3709.
In accordance with this structure, the receiver 3700 can modify the bit rate of the video and audio data for the output according to the communication rate with the external device. In this way, despite the fact that the communication speed with an external device is lower than the bit rate of the multiplexed data obtained through the demodulation carried out by the demodulator 3 702 and the error correction decoding, giving rise to the new multiplexed data from the current output interface to the external device, the user can use the new data
<img file="MX337079B_D0169.tif" />
multiplexed with other communication devices.
The receiver 3 700 further includes an audiovisual output interface 3711 that outputs the audio and video signals decoded by the signal processor 3704 to the external device through an external communications medium. Audiovisual output interface 3711 may be, for example, a wireless communications device that transmits modulated audiovisual data to an external device using a wireless transmission scheme in accordance with a wireless communications standard such as Wi-Fi ™ (IEEE 802.11a , IEEE 802.11b, IEEE 802.llg, IEEE 802.11η, etc.), WiGig, WirelessHD, Bluetooth, ZigBee, etc. through a wireless medium. The 3709 power output interface can also be a wired communications device that transmits the modulated audiovisual data to an external device using a communications scheme in accordance with a wired communications standard such as Ethernet ™, USB, PLC, HDMI, etc. . via a wired transmission path connected to the 3709 current output interface. Also, the current output interface 3709 may be a terminal for connecting a cable that outputs analog audio and video signals as they are.
In accordance with such a structure, the user can use the audio signals and the video signals decoded by the signal processor 3704 with an external device.
In addition, the 3700 receiver includes an input unit
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code 3710 that receives user operations as input. The receiver 3 700 performs in accordance with the control signals input by the operation input unit 3710 in accordance with user operations, for example connecting or disconnecting the power supply, changing the channel received, activating or disabling the subtitle display, switching between languages, changing the volume output by the 3706 audio output unit and various other operations, including modifying the settings of channels that can be received and other such procedures.
The 3700 receiver may further include functionality to display an antenna level that represents the received signal quality while the 3700 receiver receives a signal. The antenna level can be, for example, an index showing the received signal quality, calculated according to the RSSI (received signal strength indicator), the magnetic field strength of the received signal, the C / N ratio (carrier for noise), BER, packet error rate, frame error rate, channel status information, etc. received by the 3700 receiver and indicating the level and quality of a received signal. In such circumstances, the demodulator 37 02 includes a signal quality calibrator that measures the RSSI, the received signal's magnetic field strength, the C / N ratio, the BER, the packet error rate, the error rate of frames, channel status information, etc. In
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In response to user operations, the receiver 3700 displays the antenna level (signal level, signal quality) in a user-recognizable format on the video display unit 3707. The display format of the antenna level (signal level, signal quality) can be a numerical value displayed according to the RSSI, the magnetic field strength of the received signal, the C / N ratio, the BER, the rate packet error, frame error rate, channel status information, etc. , or it can be an image display that varies according to the RSSI, the magnetic field strength of the received signal, the C / N ratio, the BER, the packet error rate, the frame error rate, the channel status information etc. The 3700 receiver can display multiple antenna levels (signal level, signal quality), calculated for each current si, s2, etc., that are demultiplexed using the reception scheme explained in the previously described modes, or it can be a single antenna level (signal level, signal quality) calculated for all such currents. When the video data and audio data that make up a program are transmitted hierarchically, the signal level (signal quality) can also be displayed for each hierarchical level.
In accordance with the preceding structure, the user is given knowledge of the antenna level (signal level, signal quality) numerically or visually during reception using the reception schemes explained above.
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Although the preceding example describes the receiver 37 00 as including the audio output unit 3706, the video display unit 3707, the unit 3708, the current output interface 3709, and the audiovisual output interface 3711, all of these components are not they are strictly necessary. As long as the receiver 3700 includes at least one of the previously described components, the user will be able to use the multiplexed data obtained through the demodulation carried out by the demodulator 3702 and the error correction decoding. Any receiver can be freely combined with the components previously described according to the use of the scheme.
Multiplexed data
The following is a detailed description of an exemplary configuration of multiplexed data. The
The data configuration typically used in broadcasting is an MPEG-2 Transport Stream (TS). Therefore, the following description exposes an example related to MPEG2-TS. However, the data configuration of the multiplexed data transmitted by the transmission and reception schemes explained in the previously described modalities is not limited to MPEG2-TS. The advantageous effects of the previously described modalities can also be achieved using any other data structure.
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Figure 38 illustrates an exemplary exemplary configuration for multiplexed data. As shown, multiplexed data are elements that constitute programs (or events that are a portion thereof) currently provided by various services. For example, one or more video streams, audio streams, presentation graphics streams (PGs), interactive graphics streams (IGs), and other streaming streams. such elements are multiplexed to obtain the multiplexed data. When a broadcast program provided by the multiplexed data is a movie, the video streams represent the main video and the sub-video of the movie, the audio streams represent the main audio of the movie and the sub-audio to be mixed with the main audio , and the presentation graphics streams represent the subtitles of the movie. The main video refers to the video images normally presented on a screen, where the sub-video refers to the video images (for example, text images that explain the film review) that must be presented on a small screen inserted inside of the video images. Interactive graphics streams represent an interactive presentation made up of GUI (graphical user interface) components presented on a screen.
Each stream included in the multiplexed data is identified by an identifier called PID
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exclusively assigned to the stream. For example, the video stream carrying the main video images of a movie is assigned 0x1011, each audio stream is assigned a different one from 0x1100 to OxlllF, each PG stream is assigned a different one from 0x1200 to 0xl21F, each IG stream is assigned a different one from 0x1400 to 0xl41F, each video stream that carries sub-video images from the movie is assigned a different one from OxlBOO to OxlBlF, Each subaudio audio stream to be mixed with the main audio is assigned a different OxlAOO to OxlAlF.
FIG. 39 is a schematic diagram illustrating an example of multiplexed data being multiplexed. First, a video stream 3901, consisting of multiple video frames, and an audio stream 3904, consisting of multiple audio frames, are respectively converted to PES 3902 and 3905 packet sequences, then further converted to TS packets 3903 and 3906. Similarly, a presentation graphics stream 3911 and an interactive graphics stream 3914 are respectively converted to PES packet sequences 3912 and 3915, then further converted to TS packets 3913 and 3916. Multiplexed data 3917 consists of TS packets 3903, 3906, 3913 and 3916 multiplexed into a single stream.
Figure 40 illustrates more details of a sequence of PES packets as contained in the video stream.
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The first row of Figure 4 0 shows a sequence of video frames from the video stream. The second row shows a sequence of PES packets. The arrows yyl, yy2, yy3, and yy4 indicate the multiple video display units, which are I images, B images, and P images, in the video stream as split and individually stored as the payload of a PES packet. Each PES packet has a PES header. A PES header contains a PTS (display time) in which the image should be presented, a DTS (decode time) in which the image should be decoded, etc.
Figure 41 illustrates the structure of a TS packet as it is ultimately written to multiplexed data. A TS packet is a 188-byte fixed-length packet consisting of a 4-byte PID that identifies the stream and a 184-byte TS payload that contains the data. The PES packets described above are divided and stored individually as the TS payload. For a BD-ROM, each TS packet has a 4-byte TP_Extra_Header set to build a 192-byte source packet, which must be written as the multiplexed data. The TP_Extra_Header contains information such as an Arrival_Time_Stamp arrival time (ATS). The ATS indicates a time to start transferring the TS packet to the PID filter of a decoder. The multiplexed data is made up of source packets arranged in communication, as indicated in the bottom row of Figure 41. For each packet, an SPN (packet number of
<img file="MX337079B_D0175.tif" />
source), starting at the start of the multiplexed data.
In addition to video streams, audio streams, presentation graphics streams, and the like, the TS packets included in the multiplexed data also comprise a PAT (Program Association Table), a PMT (acronym for Program Association Table). in English for program map table), a PCR (program time reference), etc. The PAT indicates the PID of a PMT used in the multiplexed data, and the PID of the PAT itself is recorded as 0. The PMT includes the PIDs that identify the respective streams, such as video, audio, and subtitles, contained in the multiplexed data and attribute information (frame rate, aspect ratio, and the like) of the streams identified by the respective PIDs. In addition, the PMT includes various types of descriptors related to multiplexed data. One such descriptor may be copying control information indicative of whether or not copying of multiplexed data is permitted. The PCR includes information to synchronize the ATC (arrival time) that serves as the ATS timeline to the STC (system time) that serves as the PTS and DTS timeline. Each PCR packet includes an STC time corresponding to the ATS at which the packet must be transferred to the decoder.
Figure 42 illustrates the detailed data configuration of a PMT. PMT starts with a PMT header indicating
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the length of the data contained in the ”'TTTTT ^ Ce” apüés' '”Hel'” ”PMT header, the descriptors relevant to the multiplexed data are arranged. An example of a descriptor included in the PMT is the copy control information described above. After the descriptors, the current information relevant to the respective currents included in the multiplexed data is arranged. Each stream information element is made up of stream descriptors that indicate a stream type that identifies a compression codec used for a corresponding stream, a PID for the stream, and attribute information (frame rate, aspect ratio, and others by the style) of the stream. The PMT includes the same number of current descriptors as the number of currents included in the multiplexed data.
When recorded on a recording medium or the like, the multiplexed data is recorded along with a multiplexed data information file.
Figure 43 illustrates an exemplary configuration for the multiplexed data information file. As shown, the multiplexed data information file is management information relating to the multiplexed data, is provided in a one-to-one correspondence with the multiplexed data, and consists of the information on multiplexed data, the attribute information of stream and a ticket map.
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The multiplexed data information is made up of a system speed, an initial hour of reproduction and an final hour of reproduction. The system rate indicates the maximum data transfer rate of the multiplexed data to the PID filter of a target system decoder described below. The multiplexed data includes the ATS in a set interval so that it does not exceed the speed of the system. The start playback time is set to the time specified by the PTS of the first video frame in the multiplexed data, where the end playback time is set to the calculated time by adding the playback duration of one frame to the PTS of the latest video frame in multiplexed data.
Figure 44 illustrates an exemplary configuration for the current attribute information included in the multiplexed data information file. As shown, the current attribute information is the attribute information of each current included in the multiplexed data, recorded for each PID. That is, different attribute information elements are provided for the different streams, ie, for the video streams, the audio streams, the presentation graphics streams, and the interactive graphic streams. The video stream attribute information indicates the compression codec used to compress the video stream, the resolution of
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the individual images that make up the video stream, aspect ratio, frame rate, etc. The audio stream attribute information indicates the compression codec used to compress the audio stream, the number of channels included in the audio stream, the language of the audio stream, the sample rate, etc. This information is used to initialize the decoder prior to playback by a player.
In the present embodiment, the type of stream included in the PMT is used among the information included in the multiplexed data. When the multiplexed data is recorded on a recording medium, the video stream attribute information included in the multiplexed data information file is used. Specifically, the video encoding method and device described in any of the preceding modalities can be modified to further include a step or unit for adjusting a specific information element in the type of stream included in the PMT or in the attribute information of video stream. The specific information element is to indicate that the video data is generated by the video encoding method and the device described in the modality. In accordance with such a structure, the video data generated by the video encoding method and the device described in any of the preceding modalities are distinguishable from video data that is in accordance with other standards.
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Figure 45 illustrates an exemplary configuration of an audiovisual output device 4500 including a receiving device 4504 that receives a modulated signal that includes the audio and video data transmitted by a broadcast station (base station) or the data intended for diffusion. The configuration of the receiving device 4504 corresponds to the receiving device 3700 of FIG. 37. The 4500 AV output device incorporates, for example, an OS (operating system acronym in English) or incorporates a communications device 4506 for connecting to the Internet (for example, a communications device intended for a LAN (network acronym) local area) wireless or Ethernet ™). By itself, a video display unit 4501 can simultaneously display audio and video data, or video data video for broadcast 4502 and hypertext 4503 (from the World Wide Web network) provided by the Internet. By operating a 4507 remote control (as an alternative to a mobile phone or keyboard), any of the 4502 video broadcast video data and 4503 hypertext provided over the Internet can be selected to change operations. For example, when hypertext 4503 provided by the Internet is selected, the displayed website can be changed by remote control operations. When the audio and video data is selected, or the video of the video data for broadcast 4502, the remote control 4507 can transmit the information of a selected channel (program (television) or
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selected audio broadcast). In this way, an interface 4505 obtains the information transmitted by the remote control. The receiving device 4504 performs the processing, such as demodulation and error correction, corresponding to the selected channel, thereby obtaining the received data. At this point, the receiving device 4504 obtains the control symbol information including the transmission scheme information (as described using Figure 5) from the control symbols included in the signal corresponding to the selected channel. By itself, the reception device 4 504 can correctly establish the reception operations, the demodulation scheme, the error correction scheme, etc., thus making it possible to obtain the data included in the data symbols transmitted by the broadcast station (base station). Although the foregoing description is given for an example of the user using remote control 4507, the same operations apply when the user presses a select key embedded in audiovisual output device 4500 to select a channel.
Also, the 4500 audiovisual output device can be operated using the Internet. For example, audiovisual output device 4500 can be made to record (store) a program through another terminal connected to the Internet. (Accordingly, audiovisual output device 4500 must include unit 3708 of FIG. 37). Channel e selects
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before starting recording. In this way, reception 4504 '' ei “d ± ^ spos ± 'tivo-de *”' performs the processing, such as demodulation and error correction, corresponding to the selected channel, thus obtaining the received data. At this point, the receiving device 4504 obtains the control symbol information that includes the information about the transmission scheme (the transmission scheme, the modulation scheme, the error correction scheme, etc.) from those previously described. modalities) (as described using Figure 5) of the control symbols included in the signal corresponding to the selected channel. In itself, the reception device 4504 can correctly establish the reception operations, the demodulation scheme, the error correction scheme, etc., thus making it possible to obtain the data included in the data symbols transmitted by the station. diffuser (base station).
Supplement
The present disclosure considers a communication / broadcast device such as a broadcast station, base station, access point, terminal, mobile phone, or the like provided with the transmission device, and a communication device such such as a television, radio, terminal, personal computer, mobile phone, access point, base station, or the like provided with the receiving device. The device
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Transmission and receiving device relevant to the present invention are communication devices in a form suitable for executing applications, such as a television, radio, personal computer, mobile phone or the like, through connection to some kind of interface ( for example, USB).
Also, in the present embodiment, symbols other than data symbols, such as pilot symbols (i.e. preamble, single word, postamble, reference symbols, scattered pilot symbols, etc.), symbols may be freely arranged within the frame, the symbols intended for control information, etc. Although these are currently called pilot symbols and symbols intended for control information, such symbols may be freely named differently, as their function remains and is considered important.
As long as a pilot symbol, for example, is a known symbol modulated with PSK modulation on the transmitter and receiver (alternatively, the receiver can be synchronized so that the receiver knows the symbols transmitted by the transmitter), the receiver can use that symbol for frequency synchronization, time synchronization, channel estimate (the CSI estimate for each modulated signal), signal detection and other such procedures.
Symbols intended for control information are symbols that transmit information (such as the
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modulation, the error correction coding scheme, the error correction code coding rate, and the setting information for the upper layer used in communications) that is transmitted to the receiving party in order to execute the transmission of no data (i.e. applications).
The present invention is not limited to the embodiments, but can also be carried out in various other ways. For example, although the preceding embodiments describe communication devices, the present invention is not limited to such devices and can be implemented as software for the corresponding communication scheme.
Although the previously described modalities describe phase change schemes for the transmission schemes of two modulated signals from two antennas, no limitation is provided in this regard. Precoding and a phase change can be performed on four signals that have been correlated to generate four modulated signals transmitted using four antennas. That is, the present invention is applicable to perform a phase change on N signals that have been correlated and precoded to generate N modulated signals transmitted using N antennas.
Although the previously described modalities describe examples of systems where two modulated signals are transmitted from two antennas and are received by two respective antennas in a MIMO communication system, the present
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The invention is not limited in this regard and is also applicable to MISO (multiple input, single output) communication systems. In a MISO system, the receiving device does not include antenna 701_Y, wireless unit 703_Y, channel jitter estimator 707_l for the modulated signal zl, and channel jitter estimator 707_2 for the modulated signal z2 of FIG. 7 . However, the processing described in Mode 1 can still be performed to estimate rl and r2. The technology for receiving and decoding multiple signals transmitted simultaneously on a common frequency and received by a single antenna is widely known. The present invention is additional processing that supplements conventional technology for a signal processor that inverts a phase changed by the transmitter.
Although the present invention describes examples of systems where two modulated signals are transmitted from two antennas and received by two respective antennas in a MIMO communication system, the present invention is not limited in this regard and is also applicable to MISO systems. In a MISO system, the transmitting device performs precoding and phase change so that the points described so far are applicable. However, the receiving device does not include the antenna 701_Y, the wireless unit 703_Y, the channel jitter estimator 707_l for the modulated signal
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zl, and the channel jitter estimator 707_2 for the modulated signal z2 of FIG. 7. However, the processing set forth in the present description can still be performed to estimate the data transmitted by the transmission device. The technology for receiving and decoding multiple signals transmitted simultaneously on a common frequency, and received by a single antenna is widely known (a single antenna receiver can apply ML operations (Max-log APP or the like)). The present invention can cause the signal processor 711 of FIG. 7 to perform demodulation (detection) taking into account the precoding and phase change applied by the transmitter.
The present description uses terms such as precoding, precoding weights, precoding matrix, etc. The terminology itself may be different (for example, a codebook may be referred to as an alternative), since the key point of the present invention is the signal processing itself.
Also, although the present description explains the examples mainly using OFDM as the transmission scheme, the invention is not limited in that sense. Other multi-carrier schemes other than OFDM and single carrier schemes can be used to achieve similar modalities. Here, spread spectrum communications can also be employed. When using single carrier schemes, a change is made
157
ΓΜΡ
<img file="MX337079B_D0185.tif" />
'Ί phase with respect to the time domain. - - Also, although this description explains the use of ML, APP, Max-log APP, ZF, MMSE etc. operations. By the receiving device, these operations can all be generalized as wave detection, demodulation, detection, estimation and demultiplexing as the qualitative results (logarithmic likelihood and logarithmic likelihood ratio) and the quantitative results (zeros and ones) thus obtained are the bits of individual data transmitted by the transmission device.
Different data can be transmitted for each current sl (t) and s2 (t) (si (i), s2 (i)) or identical data of the same.
The baseband signals of the two currents si (i) and s2 (i) (where i indicates sequence (with respect to time or frequency (carrier))) go through precoding and a regular phase change (the order of the operations can be freely reversed) to generate two post-processing baseband signals zl (i) and z2 (i). In the post-processing baseband signal zl (i), the phase component I is Ii (i) while the quadrature component is Qi (i), and in the post-processing baseband signal z2 (i ), the phase component is Ii (i) while the quadrature component is Q<sub>2</sub> (i). Baseband components may change, as long as the following are maintained.
Let be the component in phase and the component of
158
<img file="MX337079B_D0186.tif" />
quadrature of the changed baseband signal rl (i) Ιχ (i) and Q<sub>2</sub> (i), and let be the phase component and the quadrature component of the changed baseband signal r2 (i) I<sub>2</sub> (i) and Qi (i>. The modulated signal corresponding to the changed baseband signal rl (i) is transmitted by transmit antenna 1 and the modulated signal corresponding to the changed baseband signal r2 (i) is transmitted from transmitting antenna 2, simultaneously on a common frequency. Thus, the modulated signal corresponding to the changed baseband signal rl (i) and the modulated signal corresponding to the changed baseband signal r2 (i) are transmitted from different antennas, simultaneously on a common frequency. As an alternative,
For the changed baseband signal rl (i), the in-phase component can be Ιχ (ΐ) while the quadrature component can be I<sub>2</sub> (i), and for the changed baseband signal r2 (i), the in-phase component can be Qx (i) while the quadrature component can be Q<sub>2</sub>(i).
For the changed baseband signal rl (i), the in-phase component can be I<sub>2</sub>(i) insofar as the quadrature component can be Ιχ (ϊ), and for the changed baseband signal r2 (i), the phase component can be Qx (i) while the quadrature component can be Q<sub>2</sub>(i).
For the changed baseband signal rl (i), the in-phase component can be Ix (i) while the quadrature component can be I<sub>2</sub> (i), and for the changed baseband signal
<img file="MX337079B_D0187.tif" />
r2 (i), the phase component can be Q<sub>2</sub>(i) while the quadrature component can be Qi (i).
For the changed baseband signal rl (i), the in-phase component can be I<sub>2</sub>(i) insofar as the quadrature component can be I<sub>x</sub>(i), and for the changed baseband signal r2 (i), the in-phase component can be Q<sub>2</sub>(i) insofar as the quadrature component can be Q<sub>x</sub>(i).
* For the changed baseband signal rl (i), the in-phase component can be Ii (i) while the quadrature component can be Q<sub>2</sub> (i), and for the changed baseband signal r2 (i), the in-phase component can be Qi (i) while the quadrature component can be I<sub>2</sub>(i).
For the changed baseband signal rl (i), the in-phase component can be Q<sub>2</sub>(i) while the quadrature component can be Ii (i), and for the changed baseband signal r2 (i), the phase component can be I<sub>2</sub>(i) while the quadrature component can be Qi (i).
For the changed baseband signal rl (i), the in-phase component can be Q<sub>2</sub>(i) while the quadrature component can be Ii (i), and for the changed baseband signal r2 (i), the phase component can be Q<sub>x</sub>(i) insofar as the quadrature component can be I<sub>2</sub>(i).
"For the changed baseband signal r2 (i), the in-phase component may be I<sub>x</sub>(i) insofar as the quadrature component can be I<sub>2</sub> (i), and for the changed baseband signal
160
<img file="MX337079B_D0188.tif" />
rl (i), the phase component can be Qi (i) while '' the '' quadrature component can be Q<sub>2</sub>(i).
For the changed baseband signal r2 (i), the in-phase component can be I<sub>2</sub>(i) insofar as the quadrature component can be I<sub>x</sub> (i), and for the changed baseband signal rl (i), the in-phase component can be Qi (i) while the quadrature component can be Q<sub>2</sub> (i).
For the changed baseband signal r2 (i), the in-phase component can be Ii (i) while the quadrature component can be I<sub>2</sub> (i), and for the changed baseband signal rl (i), the phase component can be Q<sub>2</sub> (i) while the quadrature component can be Qi (i).
For the changed baseband signal r2 (i), the in-phase component can be I<sub>2</sub>(i) while the quadrature component can be Ii (i), and for the changed baseband signal rl (i), the in-phase component can be Q<sub>2</sub>(i) while the quadrature component can be Qi (i).
»For the changed baseband signal r2 (i), the in-phase component can be I<sub>x</sub>(i) insofar as the quadrature component can be Q<sub>2</sub> (i), and for the changed baseband signal rl (i), the phase component can be I<sub>2</sub>(i) while the quadrature component can be Qi (i).
• For the changed baseband signal r2 (i), the in-phase component can be Ii (i) while the quadrature component can be Q<sub>2</sub> (i), and for the changed baseband signal
161
<img file="MX337079B_D0189.tif" />
rl (i), the phase component can be Qi (i) while the quadrature component can be I<sub>2</sub>(i).
»For the changed baseband signal r2 (i), the in-phase component can be Q<sub>2</sub>(i) while the quadrature component can be (i), and for the changed baseband signal rl (i), the phase component can be I<sub>2</sub>(i) while the quadrature component can be Qi (i).
• For the changed baseband signal r2 (i), the in-phase component can be Q<sub>2</sub>(i) insofar as the quadrature component can be Ii (i), and for the changed baseband signal rl (i), the phase component can be Qi (i) while the quadrature component can be I<sub>2</sub> (i). As an alternative, although the preceding description explains the modality of two types of signal processing in the signals of both currents as to change the component in phase the quadrature component of the two signals, the invention is not limited in that sense. The two types of signal processing can be performed on more than two currents, such as changing the phase component and the quadrature component thereof.
Alternatively, although the preceding examples describe changing baseband signals that have a common date-time (common (sub-) frequency), the changing baseband signals need not necessarily have a common datetime. . For example, any of the following are possible.
162
<img file="MX337079B_D0190.tif" />
For the changed baseband signal rl (i), the in-phase component can be Ii (i + v) while the quadrature component can be Q<sub>2</sub>(i + w), and for the changed baseband signal r2 (i), the in-phase component can be l<sub>2</sub>(i + w) while the quadrature component can be QMi + v).
* For the changed baseband signal rl (i), the in-phase component can be Ii (i + v) while the quadrature component can be I<sub>2</sub>(i + w), and for the changed baseband signal r2 (i), the in-phase component can be Qi (i + v) while the quadrature component can be Q<sub>2</sub>(i + w).
* For the changed baseband signal rl (i), the in-phase component can be I<sub>2</sub>(i + w) insofar as the quadrature component can be Ii (i + v), and for the changed baseband signal r2 (i), the phase component can be Qi (i + v) while the quadrature component can be Q<sub>2</sub>(i + w).
For the changed baseband signal rl (i), the in-phase component can be I<sub>x</sub>(i + v) while the quadrature component can be I<sub>2</sub>(i + w), and for the changed baseband signal r2 (i), the in-phase component can be Q<sub>2</sub>(i + w) while the quadrature component can be Qi (i + v).
* For the changed baseband signal rl (i), the in-phase component can be I<sub>2</sub>(i + w) while the quadrature component can be Ii (i + v), and for the changed baseband signal r2 (i), the phase component can be Q<sub>2</sub>(i + w) while the quadrature component can be Qi (i + v).
163
I '. <- ·>
ΙΝ37ΙΪ * For the changed baseband signal γΓ (ι7'7 “^ 'Ϊ<sup>1</sup>* '' · Phase component can be Ii (i + v) while quadrature component can be Q<sub>2</sub>(i + w), and for the signal of. changed baseband r2 (i), the in-phase component can be Qi (i + v) while the quadrature component can be I<sub>2</sub>(i + w).
For the changed baseband signal rl (i), the in-phase component can be Q<sub>2</sub>(i + w) while the quadrature component can be Ii (i + v), and for the changed baseband signal r2 (i), the phase component can be I<sub>2</sub>(i + w) while the quadrature component can be Qi (i + v).
For the changed baseband signal rl (i), the in-phase component can be Q<sub>2</sub>(i + w) insofar as the quadrature component can be Ii (i + v), and for the changed baseband signal r2 (i), the phase component can be Qi (i + v) while the quadrature component can be I<sub>2</sub>(i + w).
• For the changed baseband signal r2 (i), the in-phase component can be Ii (i + v) while the quadrature component can be I<sub>2</sub>(i + w), and for the changed baseband signal rl (i), the in-phase component can be Qi (i + v) while the quadrature component can be Q<sub>2</sub>(i + w).
"For the changed baseband signal r2 (i), the in-phase component may be I<sub>2</sub>(i + w) insofar as the quadrature component can be Ii (i + v), and for the changed baseband signal rl (i), the phase component can be Qi (i + v) while the quadrature component can be Q<sub>2</sub>(i + w).
164
<img file="MX337079B_D0191.tif" />
* For the changed baseband signal r2 (i), the in-phase component can be I<sub>x</sub>(i + v) while the quadrature component can be I<sub>2</sub>(i + w), and for the changed baseband signal rl (i), the in-phase component can be Q<sub>2</sub>(i + w) while the quadrature component can be Q<sub>x</sub>(i + v).
For the changed baseband signal r2 (i), the in-phase component can be I<sub>2</sub>(i + w) while the quadrature component can be Ii (i + v), and for the changed baseband signal rl (i), the phase component can be Q<sub>2</sub>(i + w) while the quadrature component can be Q<sub>x</sub>(i + v).
For the switched baseband signal r2 (i), the in-phase component can be l<sub>x</sub>(i + v) while the quadrature component can be Q<sub>2</sub>(i + w), and for the changed baseband signal rl (i), the phase component can be I<sub>2</sub>(i + w) while the quadrature component can be Q<sub>x</sub>(i + v).
• For the changed baseband signal r2 (i), the in-phase component can be I<sub>x</sub>(i + v) while the quadrature component can be Q<sub>2</sub>(i + w), and for the changed baseband signal rl (i), the in-phase component can be Q<sub>x</sub>(i + v) while the quadrature component can be I<sub>2</sub>(i + w).
"For the changed baseband signal r2 (i), the in-phase component may be Q<sub>2</sub>(i + w) while the quadrature component can be Ii (i + v), and for the changed baseband signal rl (i), the phase component can be I<sub>2</sub>(i + w) while the quadrature component can be Qi (i + v).
165
<img file="MX337079B_D0192.tif" />
For the phase component baseband signal it can be Q<sub>2</sub>(i + w) insofar as the quadrature component can be Ii (i + v), and for the changed baseband signal rl (i), the phase component can be Qi (i + v) while the quadrature component can be I<sub>2</sub>(i + w).
Figure 55 illustrates a baseband signal changer 5502 that explains the above. As shown, of the two processed baseband signals zl (i) 5501_l and z2 (i) 5501_2, the processed baseband signal zl (i) 5501_l has the phase component Ii (i) and the quadrature component Qi (i), while the processed baseband signal z2 (i) 5501_2 has the phase component I<sub>2</sub>(i) and the quadrature component Q<sub>2</sub>(i). Then, after the change, the changed baseband signal rl (i) 5503_l has the phase component I<sub>r</sub>i (i) and the quadrature component Q<sub>ri</sub>(i), whereas the changed baseband signal r2 (i) 5503_2 has the phase component I<sub>r</sub>2 (i) and the quadrature component Q<sub>r</sub>2 (i) · The component in phase I<sub>r</sub>i (i) and the quadrature component Q<sub>r</sub>i (i) of the changed baseband signal rl (i) 5503_l and the in-phase component Ir2 (i) and the quadrature component Q<sub>r2</sub>(i) of the changed baseband signal r2 (i) 5503_2 can be expressed as any of the preceding. Although this example describes the change made to baseband signals that have a common date-time (common ((sub-) carrier) frequency) and that have gone through two types of signal processing, it can be applied to signals baseband that have gone through two types
166
<img file="MX337079B_D0193.tif" />
signal processing but have different date-time indicators (different frequencies ((sub-) carrier)).
Each of the transmitting antennas of the transmitting device and each of the receiving antennas of the receiving device shown in the figures may be made up of multiple antennas.
The present description uses the symbol V, which is the universal quantifier, and the symbol Ξ, which is the existential quantifier.
Also, the present description uses the radian as the phase unit in the complex plane, for example, for its argument.
When dealing with the complex plane, the coordinates of complex numbers can be expressed as polar coordinates. For a complex number z = a + jb (where a and b are real numbers and j is the imaginary unit), the corresponding point (a, b) on the complex plane is expressed with the polar coordinates [r, Θ], converted as follows :
a = rx sew b = rx sene [Mathematics 49] (formula 49) r = y¡a<sup>2</sup> + b<sup>2</sup> where r is the absolute value of z (r = | z |), and Θ is
<img file="MX337079B_D0194.tif" />
its argument. Thus, z = a + jb can be expressed as re<sup>J</sup>®.
In the present invention, the baseband signals si, s2, zl, and z2 are described as complex signals. A complex signal consisting of the phase I signal and the quadrature signal Q can also be expressed as the complex signal I + j'Q. Here, either I and Q can be equal to zero.
Figure 46 illustrates an emplicative eg diffusion system using the phase change scheme set forth in the present disclosure. As shown, a 4601 video encoder takes video as input, performs video encoding, and outputs encoded 4602 video data. An audio encoder takes audio as input, performs audio encoding, and outputs 4604 encoded audio data. A 4605 data encoder takes data as input, performs data encoding (for example, data compression), and outputs the 4606 encoded data. Taken together, these components form a source information encoder.
4600.
A 4607 transmitter takes the encoded video data
4602, encoded audio data 4604, and encoded data 4606 as input, performs error correction encoding, modulation, precoding, and phase shifting (for example, signal processing by the transmission device of the figure 3) in a subset or in its entirety
168
<img file="MX337079B_D0195.tif" />
of them, and gives exit to the signs of ι - * »« · η «ιηι · πι · ί nn..A.goa__l at 4608_N. Transmission signals 4608_l through 4608_N are then transmitted by antennas 4609_l through 4609_N as radio waves.
A receiver 4612 takes the received signals 4611_1 to 4611_M that the antennas 4610_l to 4610_M receive as input, performs processing such as frequency conversion, phase change, decoding the precoding, calculating the logarithmic likelihood ratio, and error correction decoding (eg, processing by the receiving device of Figure 7), and outputs the received data 4613, 4615, and 4617. A source information decoder 4 619 takes the received data 4613, 4615 and 4617 as input. A video decoder 4614 takes received data 4613 as input, performs video decoding, and outputs a video signal. The video is then presented on a television screen. An audio decoder 4616 takes the received data 4615 as input. The audio decoder 4616 performs the audio decoding and outputs an audio signal. The audio is then played through the speakers. A data decoder 4618 takes the received data 4617 as input, performs data decoding, and outputs information.
In the previously described embodiments relevant to the present invention, the number of encoders in the transmission device using a transmission scheme
<img file="MX337079B_D0196.tif" />
169
<img file="MX337079B_D0197.tif" />
of multicarrier such as OFDM can be cuaT¿¿lél'cl., LU.1 cuiuu already described. Therefore, as in Figure 4, for example, the transmission device may have only one encoder and apply a scheme to distribute the output to the multi-carrier transmission scheme, such as OFDM. In such circumstances, the wireless units 310A and 310B in Figure 4 should replace the OFDM 13OIA and 1301B related processors in Figure 12. The description of OFDM related processors is as given in Mode 1.
Although Mode 1 gives Math 36 (formula 36) as an example of a precoding matrix, another precoding matrix can also be used, when the following scheme is applied.
[Mathematics 50] (formula 50) <sup>r</sup>wll <sub>k</sub>w21
34^22^
<img file="MX337079B_D0198.tif" />
and
JO {ax jF \ ax
JO e)
In the precoding matrices of Mathematics 36 (formula 36) and Mathematics 50 (formula 50), the value of □ is established as given by Mathematics 37 (formula 37) and Mathematics 38 (formula 38). However, no limitation is provided in this regard. A simple precoding matrix can be obtained by setting a = 1, which is also a valid value.
170
<img file="MX337079B_D0199.tif" />
In Modality Al, the phase changers of Figures 3, 4, 6, 12, 25, 29, 51 and 53 are indicated as having a phase change value of PHASE [i] (where i = 0, 1, 2 ... N-2, Nl, that is, O ^ i ^ N-1, i being an integer) to achieve a period (cycle) of N (value reached in the case that Figures 3, 4, 6 , 12, 25, 29, 51 and 53 perform a phase change on only one baseband signal). The present description explains performing a phase change on a precoded baseband signal (i.e. in Figures 3, 4, 6, 12, 25, 29 and 51 ), i.e. on the precoded baseband signal z2 '. Here, PHASE [k] is calculated as follows.
[Mathematics 51] (formula 51)
Á 77
PHASE [£] = radians TV where k = 0, 1, 2 ... N-2, Nl. When N = 5, 7, 9, 11 or 15, the receiving device can obtain a good quality of data reception.
Although the present description explains the details of phase change schemes involving two modulated signals transmitted by multiple antennas, no limitation is provided in that regard. Precoding and phase change can be performed on three or more baseband signals on which correlation has been performed according to a modulation scheme, followed by processing
171
<img file="MX337079B_D0200.tif" />
default on baseband signals with '· depbLf L'f as tí change and transmission using multiple antennas, to perform the same results.
Programs to execute the above transmission scheme can be stored, eg, in advance in ROM (read-only memory) and read for operation by a CPU.
Also, programs for executing the above transmission scheme may be stored on a computer-readable recording medium, programs stored on the recording medium may be loaded into the computer's RAM (Random Access Memory), and the computer can work according to the programs.
The components of the previously described embodiments can typically be assembled as an LSI (Large Scale Integration), a type of integrated circuit. The individual components can be made respectively on discrete chips, or a subset or all of the components can be made on a single chip. Although an LSI has been mentioned before, the terms IC (integrated circuit), LSI system, super LSI, or ultra LSI may also apply, depending on the degree of integration. Also, the method for assembling the integrated circuit is not limited to LSI. A dedicated circuit or general purpose processor can be used. After mounting the LSI, either an FPGA (programmable gate array) or a reconfigurable processor can be used.
~~ * Τ * ι * Τ V η—
172
<img file="MX337079B_D0201.tif" />
Also, should progress in the semiconductor field or the emergence of technologies lead to the replacement of the LSI by other integrated circuit methods, such technology can of course be used to integrate the functional blocks. The applications of biotechnology are also plausible.
Industrial Applicability
The present invention is widely applicable to wireless systems that transmit multiple different modulated signals from multiple antennas, such as an OFDM-MIMO system. In addition, in a wired communication system with multiple transmission locations (such as a PLC (power line communication) system, an optical communication system, or a DSL (digital subscriber line) system, The present invention can be adapted to a MIMO system, where multiple transmission locations are used to transmit multiple modulated signals as described in the present invention. Modulated signals can also be transmitted from multiple transmission locations.
173
<img file="MX337079B_D0202.tif" />
List of reference signs
<td>302A,</td><td>302B</td><td>Encoders</td>
<td>304A,</td><td>304B</td><td>Interpolators</td>
<td>306A,</td><td>306B</td><td>Correlators</td>
<td> 314</td><td></td><td>Information generator</td>
<td>308A,</td><td>308B</td><td>processing scheme signs Weighting units</td>
<td>310A,</td><td>310B</td><td>Wireless units</td>
<td>312A,</td><td>312B</td><td>Antennas</td>
<td>317A,</td><td>317B</td><td>Phase changers</td>
<td>402 404 504 # l,</td><td> 504#2</td><td>Encoder Distributor Transmission antennas</td>
<td>505 # l,</td><td> 505#2</td><td>Receiving antennas</td>
<td>600 701_X,</td><td>701_Y</td><td>Weighting unit Antennas</td>
<td>703_X,</td><td>703_Y</td><td>Wireless units</td>
<td>705_l</td><td></td><td>Fluctuation estimator</td>
<td> 705_2</td><td></td><td>channel Fluctuation estimator</td>
<td> 707 1</td><td></td><td>channel Fluctuation estimator</td>
channel
174
<img file="MX337079B_D0203.tif" />
07_2
709
711
803
805Α, 805Β
807Α, 807Β
809Α, 809Β
811Α, 811Β
813Α, 813Β
815
819
901
903
1201Α, 1201Β
1302Α, 1302 '
Channel jitter estimator
Control information decoder
Signal processor
INNER MIMO detector
Log-Likelihood Calculators Deinterpolators
Logarithmic Likelihood Ratio Calculators
Software input / output decoders Interpolators
Memory
Coefficient generator
Software input / output decoder
Distributor
OFDM related processors
Serial to parallel converters
1304A, 1304B
Reordered tes
175
1306Α, 1306Β
1308Α, 1308Β
IFFT units
Wireless units
<img file="MX337079B_D0204.tif" />
It is noted that in relation to this date, the best method known to the applicant for practicing the present invention is the one that is clear from the present description of the invention.
<img file="MX337079B_D0205.tif" />
176
<img file="MX337079B_D0206.tif" />
Contents18
261 sheets
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70 members in 8 offices
Priority claims3
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| KR20130142987A | Republic of Korea | A | |
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Numbers
- Publication
- 337079
- Application
- 2014008920
Titles2
- Spanish
- METODO DE GENERACION DE SEÑALES Y DISPOSITVO DE GENERACION DE SEÑALES.
- English
- SIGNAL GENERATION METHOD AND SIGNAL GENERATION DEVICE.
Classification
- CPC, 4
- H04B7/0456
- H04L25/0391
- H04B7/0617
- H04B7/0413
- IPC, 2
- H04J99 00
- H04B7 04