An ultrawide-band communications system and method

Background of the Invention Field of Invention The present invention relates to the field of communication, and more specifically, the present invention relates to an ultra-wideband impulse communication system using a subcarrier and a method thereof. Related Technologies Radio technology designers of personal communications equipment, medical and military equipment, etc. are currently facing some development challenges. Low power consumption, reuse of available spectra, channelization and cost are the four main issues. These problems are partially solved by the latest revolutionary technology called impulse radio communication (hereinafter referred to as impulse radio in the present specification). Impulse Radio has U.S. Patent Application Nos. 4,641,317 (issued February 3, 1987), 4,813,057 (issued March 14, 1989), and 4,979,186 to Larry W. Fullerton. It is first fully described in a series of patents, including 07 / 368,831 (accepted December 18, 1989). These patent specifications are incorporated herein by reference. A basic impulse radio transmitter is an average pulse-to-pulse It emits short Gaussian monocycle pulses with tightly controlled intervals. Pulse position modulation is a form of time modulation in which the value of each instantaneous sample of a modulated signal modulates the temporal position of the pulse. In impulse radio communication, the pulse interval has two components, an information element and a pseudo-random code. It changes for each pulse depending on the component). Diffuse spectrum systems use pseudo-random codes to spread a normally narrowband information signal over a relatively wide frequency band. The diffusion spectrum receiver correlates these signals to extract the original information signal. Unlike diffuse spectrum systems, pseudo-random codes for impulse radio communications do not need to be energy diffused because the monocycle pulse itself has an inherently wide information bandwidth (bands below). The information bandwidth, called width, is the frequency range in which the performance of some features falls within the range of the characteristics). Instead, pseudo-random codes are channelized, energy smoothing in the frequency domain, and jamming. Used for resistance). The impulse radio receiver is a homodyne receiver with a cross-correlation front end. The front end coherently converts a monocycle pulse electromagnetic pulse train into a baseband signal (baseband signal is the basic information channel of a basic impulse radio communication system, also known as the information hand width). The data rate of impulse radio transmission is a fraction of the periodic timing signal used as the time base. Each data bit<u style="single">To</u>Multiple pulses of periodic timing signal<u style="single">Time position</u>Modulate. This results in a modulated coded timing signal containing the same pulse train for each single data bit. The cross-correlator of the impulse radio receiver integrates a large number of pulses to restore the transmitted information. As with all aspects of the electronics field, what is desired is a smaller, lower power and more flexible system. However, the generally accepted principles of continuous wave (CW) radio technology do not easily apply to time domain systems such as impulse radio. An explanation of some of the basic concepts described below can be found in numerous references, Robert C. Dixon's Spread Spectrum Systems (John Wiley &amp; Sons, Inc., New York, 1984, 2nd ed.) And Don Torrieri's Principles of Military Communication Systems (Artech House, Inc.) ., Dedham Massachusetts, 1982, 3rd ). Abstract of the Invention The impulse radio communication system according to the invention uses one or more subcarriers to communicate information from an impulse radio transmitter to an impulse radio receiver. Examples of three types of impulse radio communication systems including a one-channel system, a two-channel system, and a system of three or more channels will be described. Applications of typical radio frequency impulse radio communication systems include cellular phone telephones, radio telephones, wireless PBX / local area networks, and others. The impulse radio communication system is an ultra wideband time domain system. The operation in the time domain follows the general impulse radio logic described in Part 2 below. The use of subcarriers provides channelization, smoothing and fidelity with impulse radio transmission. Subcarriers with different frequencies or waveforms can be used (simultaneously) to add channelization of impulse radio signals. That is, the impulse radio link can communicate simultaneously on many independent channels by using different subcarriers on each channel. Examples of three types of impulse radio transmitters will be described. First and second transmitter embodiments include subcarrier generators and modulators that modulate a periodic timing signal with one or more information signals. According to the first embodiment, the (coding) of the impulse radio signal is realized by a periodic timing signal before it is time modulated by the modulated subcarrier signal. According to the second embodiment, the coding of the impulse radio signal is achieved by coding the modulated subcarrier signal before using it for time modulation of the periodic timing signal. A third transmitter embodiment uses one or more information signals to provide a subcarrier generator and modulator that modulates a periodic timing signal in combination with direct digital modulation of a digital data signal. Including. In this example The modulated subcarrier signal is used to time-modulate the directly digitally modulated signal. Impulse radio transmitters generally include a time base that produces a periodic timing signal. The time base includes voltage controlled oscillators or similar circuits with timing requirements of nanoseconds or less. The periodic timing signal is supplied to the code source and the code time modulator. The code source is pseudo-random noise (PN:) that is almost orthogonal. noise) A storage device for storing a code and a means for outputting a PN code as a code signal are included. The code source can monitor the periodic timing signal and synchronize the code signal with the code time modulator. In one embodiment, the code time modulator uses the code signal to modulate the sync timing signal for channelization and smoothing of the finally emitted impulse radio signal. The output of the code time modulator is called a coded timing signal. The coded timing signal is supplied to the subcarrier time modulator for information modulation. Traditional impulse systems have used baseband modulation without subcarriers. In other words, the information itself was used for modulation. However, in the present invention, the information source supplies the information signal to the subcarrier generator and the modulator. The information signal can be any kind of intelligence, including digital bits, analog signals, or synthetic signals that represent voice, data, images, or the like. The subcarrier generator and modulator of the present invention generate a modulated subcarrier signal modulated by an information signal, and supply the modulated subcarrier signal to the subcarrier time modulator. That is, the modulated subcarrier signal is used in the subcarrier time modulator to modulate the carrier, which in this example is the coded timing signal. Modulation of a coding time signal by a subcarrier time modulator produces a modulation coding timing signal sent to the output stage. The output stage uses a modulation-encoded timing signal as a trigger to generate a monocycle pulse. In the radio frequency embodiment, the monocycle pulse is sent to the transmitting antenna via a transmitting line connected to the antenna. The monocycle pulse is converted into a propagating electromagnetic pulse by the transmitting antenna. The transmitted signal propagates through the propagation medium, eg, in the air in the radio frequency embodiment, to the impulse radio receiver. In a preferred embodiment, the radiated signal is a wideband or ultra wideband signal.<u style="single">Can be changed</u>.. This filtering causes each monocycle pulse to have many zero intersections in the time domain. In this case, the impulse radio receiver needs to use a similar waveform in the cross-correlator to be efficient. There are several examples of impulse radio receivers. Each impulse radio receiver is generally cross-correlated (coross) Includes correlator), decode source, decode timing modulator and adjustable time base, and subcarrier decoder. The decode source generates a decode control signal corresponding to the PN code used in the impulse radio transmitter that communicates the impulse radio signal. The adjustable time base produces a periodic timing signal containing a template signal pulse train with a waveform substantially equivalent to each pulse of the received (impulse radio) signal. The decode timing modulator uses the decode control signal to temporally position the periodic timing signal and generate the decode signal. Since the decoded signal will timely match the known PN code of the transmitter, the received signal can be detected by the cross correlator. The decoded signal is used to generate a template signal with a waveform designed to match the received signal. The template signal is temporally aligned according to the transmitter's known PN code and cross-correlated with the received signal. Continuous cross-correlation output signals are integrated to restore a noise-free impulse radio signal. When taken out in this way, the signal can be demodulated to remove subcarriers and an information signal can be obtained. The baseband signal is also input to the low-pass filter. A control loop containing a lowpass filter is used to generate the error signal, provide fine phase adjustment to the adjustable time base, and temporally place the periodic timing signal relative to the position of the received signal. In a preferred embodiment, the impulse radio subcarrier converts (or shifts) the baseband signal to a higher frequency. Subcarrier generators and modulators include frequency modulation (FM) technology, amplitude modulation (AM), phase modulation, frequency shift keying (FSK), phase shift keying (PSK:). Phase shift keying), pulse FM, etc. generate a signal that is modulated by an information signal. Other non-sinusoidal and / or discontinuous waveforms can also be used as subcarriers in the context of the present invention. The modulated subcarrier signal is used to time-shift the pulse position of the coded or periodic timing signal. That is, the signal that triggers the output stage is a pulse position modulated pulse train. In another embodiment, direct digital modulation using Manchester coding is used as the subcarrier. The combination of these subcarrier technologies will also be described. Modulation transfer Using a cross-correlation function as function) has the effect of making the receiver output a non-linear function of the input amplitude. For baseband modulation, this is undesirable. However, in subcarriers such as FM, AM, FSK, PSK, and Manchester, the harmonics are filtered to eliminate any distortion. When such filtering uses baseband modulation, the harmonics cannot be removed because the baseband has harmonics, and therefore the signal cannot be restored. The addition of subcarriers also provides additional fidelity in the form of wider bandwidth and better signal-to-noise ratio compared to baseband modulation alone. This advantage is attributed to the fact that subcarriers inherently make information more resistant to noise. Subcarrier examples provide low signal compression and low signal distortion by reducing baseband noise in reliable voice, data, and / or image communications. The linearity requirement for modulation using cross-correlators is greatly relaxed by using the subcarrier technology of the present invention. The use of subcarriers in impulse radio also improves harmonic distortion due to the nonlinear modulation transfer function compared to baseband modulation. Modulation transfer characteristics need to be extremely linear in order to successfully transmit low distortion speech or music. This is extremely difficult to achieve in non-subcarrier baseband impulse systems. The aforementioned and other features and advantages of the present invention will become apparent from the following description of preferred embodiments of the present invention, as illustrated in the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS 1A and 1B show 2GHz center frequency monocycle pulses according to the invention in time and frequency domains, respectively. Figures 2A and 2B show 1mpps systems with 1 nanosecond pulses according to the invention in each of the time and frequency domains. FIG. 3 shows a modulated signal that changes the pulse repetition interval (PRI) in proportion to the modulation according to the present invention. FIG. 4 is a plot showing the effect of pseudo-random dither on the energy distribution in the frequency domain according to the present invention. FIG. 5 shows the result of overlaying a narrowband sine wave (interference) signal on the impulse radio signal according to the present invention. FIG. 6 shows the cross-correlator transfer function of the impulse radio receiver according to the present invention. FIG. 7 shows the impulse radio multipath effect according to the present invention. FIG. 8 shows the phase of the multipath pulse according to the present invention. FIG. 9 shows a schematic block diagram of an impulse radio electrical system using one subcarrier channel according to the present invention. FIG. 10 shows an impulse radio transmitter of an impulse radio communication system according to the present invention. FIG. 11 shows another embodiment of the impulse radio transmitter according to the present invention. FIG. 12 shows another transmitter embodiment according to the present invention. FIG. 13 shows yet another embodiment according to the present invention. FIG. 14 shows an impulse radio receiver according to the present invention. FIG. 15 shows a representative plot of the pulses corresponding to the received signal in the context of the receiver 1400 according to the invention. FIG. 16 shows the cross-correlation process according to the present invention. FIG. 17 shows a schematic diagram of an impulse radio transmitter with three subcarrier generators / modulators according to the present invention. FIG. 18 is a representative analog example showing a cross-correlator according to the present invention and a plurality of analog FM demodulation branches following it. FIG. 19 shows a digital embodiment according to the present invention. FIG. 20 is a plot showing the delay time (in picoseconds) and the binary (ie, numerical) input values for a conventional binary-time delay generator according to the invention. Figure 21 shows the book It is a high-level block diagram which shows the said linearization method by invention. FIG. 22 is a functional diagram showing the linearized ROM 2110 according to the present invention. FIG. 23 shows the combined PN code and linearized EPROM according to the present invention. FIG. 24 shows a further embodiment of the impulse radio receiver according to the present invention. 25A to 25H are time (t) vs. voltage plots of the various signals numbered in FIG. 24 according to the present invention.<u style="single">25I-25L show frequency vs. amplitude plots corresponding to FIGS. 25E-25H according to the present invention.</u>.. 26 and 27 show typical pseudo-Manchester coding and decoding waveforms according to the present invention, respectively. FIG. 28 is a high-level block diagram of a lock-requiring operation performed on an impulse radio receiver according to the present invention. FIG. 29 shows the signal measured at 3 meters according to the present invention as well as the peripheral signal. FIG. 30 is a curve showing a particular example of the projected trade-off between the free space range and the bit rate according to the present invention. FIG. 31 shows that it is easy to resolve a multipath impulse signal in the time domain according to the present invention. In the drawings, the same reference numbers refer to the same or functionally similar elements. Further, the leftmost numerical value of the reference number represents the drawing in which the reference number first appears. Detailed Description of Preferred Examples Contents I. Overview II. Technical Foundation II.1 Gauss Monocycle II.2 Pulse Train II.3 Modulation II.4 Coding for Energy Smoothing and Channelization II.5 Reception and Demodulation II.6 Interference Resistance II.7 Processing Gain II.8 Capacity II.9 Multipath and Propagation III. Invention of subcarriers III.1 Behavior theory III.2 Improvement of baseband alone in one channel III2.a Transmitter III2.b Receiver III.3 2 or more subcarrier channels (Example: voice, digital data, and control information) IV. Time modulator V. Linearization V.1 Transmitter V.2 Receiver VI. Pseudo-Manchester Modulation VII. Lock Acquisition Method VIII. Real World Performance IX. Conclusion I. Overview Impulse radio communication systems operate in the time domain and use one or more subcarriers. An ultra-wideband time domain system that provides channelization, smoothing, and fidelity. A single impulse radio transmission (eg, a link) can communicate multiple independent channels at the same time by using different subcarriers on each channel. The impulse radio transmitter according to the invention uses a modulated subcarrier that time-places a periodic or coded timing signal. Alternatively, the coded timing signal can be mixed (or added) with the modulation subcarrier and the resulting signal can be used for time modulation of the periodic timing signal. Direct digital modulation of data is another aspect of subcarrier modulation of impulse radio signals. Direct digital modulation can be used alone for time modulation of a periodic timing signal, or a directly digitally modulated periodic timing signal can be modulated by one or more additional modulated subcarrier signals. The impulse radio technology according to the present invention can be widely applied to wireless communication applications. Since impulse radio is not a system with continuous wave (CW) carriers, the use of subcarriers is elegant and is an unintuitive addition to the time domain impulse radio design. The signal-to-noise ratio is significantly improved compared to non-subcarrier impulse radio transmission. First, the addition of subcarriers to the impulse radio communication system seems redundant. However, the layering of information modulation and subcarrier modulation for PN code smoothing in impulse radio systems has elegant consequences. Impulse radios are generally: short sustained pulses, typically center frequencies between 50MHz and 10GHz (gigahertz); ultra-wideband widths above 100% of the center frequency; It is extremely resistant to interference from other systems and multipath fading. In addition, impulse radios have extraordinary multipathing properties and can be manufactured relatively easily and at low cost, especially as compared to diffusion spectrum radios. Impulse radio systems consume substantially less power than known conventional radios. In addition, impulse radio systems occupy less space than existing portable telecommunications transceivers. Because of these characteristics, impulse radio is the technology of choice for a wide range of applications, including personal and indoor communication systems. Each section of Part II-Part VIII below is a detailed description of the invention. Part II relates to the technical foundation and provides the reader with an introduction to the concept of impulse radio and other relevant aspects of communication theory. This part contains sections related to the Gaussian monocycle pulse, the pulse sequence of the Gaussian monocycle pulse, modulation, coding, and the quantitative properties of these concepts. Part III relates to the use of subcarriers in impulse radio communication systems. This part contains sections related to the theory of operation of subcarriers in impulse radio transmitters and receivers. The description is divided to explain an example of one channel improved for baseband alone and an example of two or more subcarrier channels. Part IV deals with code time delays, subcarrier time delays, and time modulators used for combinations of both. The operation and structure of some embodiments for using the time modulator for subcarrier impulse radio communication will be described. Part V deals with time modulator linearization on both impulse radio transmitters and receivers. The linearization of the time modulator allows the impulse radio transmitter and receiver to generate a time delay with the accuracy required for impulse radio communication. Part VI relates to pseudo-Manchester coding for the modulation of digital data using impulse radio communication. Part VII is Impul A lock acquisition method in which a radio receiver acquires and maintains a lock on an impulse radio signal. Part VIII describes the performance of impulse radio communication systems in the real world with reference to the data collected by the inventor by testing prototypes. II. Technical Foundation This section relates to the technical foundation and provides the reader with an introduction to the concept of impulse radio and other relevant aspects of communication theory. This section contains sections related to Gaussian monocycle pulses, pulse trains of Gaussian monocycle pulses, modulation, coding, and the quantitative properties of these concepts. Impulse radio transmitters emit short Gauss monocycle pulses with tightly controlled average pulse intervals. Impulse radio transmitters use pulse widths between 20 and 0.1 nanoseconds (ns) and pulse intervals between 2 and 5000 nanoseconds. These narrow monocycle pulses inherently have wideband frequency characteristics. The impulse radio system uses pulse position modulation, and the actual pulse interval varies from pulse to pulse due to two components, the information element and the pseudo-random code element. Unlike diffusion spectrum systems, pseudo-random codes are not required for energy diffusion (because the impulse itself is wideband in nature), channelization, energy smoothing in the frequency domain, and anti-jamming. Needed for sex. The impulse radio receiver is a homodyne receiver with a cross-correlator front end. The front end coherently converts the electromagnetic pulse train into a baseband signal in one stage. The impulse radio receiver integrates a number of pulses to restore each bit of transmitted information. II.1 Gauss Monocycle The most basic element of impulse radio technology is the Gauss Monocycle. Technical Foundation This section relates to the technical foundation and provides the reader with an introduction to the concept of impulse radio and other relevant aspects of communication theory. This section contains sections related to Gaussian monocycle pulses, pulse trains of Gaussian monocycle pulses, modulation, coding, and the quantitative properties of these concepts. Impulse radio transmitters emit short Gauss monocycle pulses with tightly controlled average pulse intervals. Impulse radio transmitters use pulse widths between 20 and 0.1 nanoseconds (ns) and pulse intervals between 2 and 5000 nanoseconds. These narrow monocycle pulses inherently have wideband frequency characteristics. The impulse radio system uses pulse position modulation, and the actual pulse interval varies from pulse to pulse due to two components, the information element and the pseudo-random code element. Unlike diffusion spectrum systems, pseudo-random codes are not required for energy diffusion (because the impulse itself is wideband in nature), channelization, energy smoothing in the frequency domain, and anti-jamming. Needed for sex. The impulse radio receiver is a homodyne receiver with a cross-correlator front end. The front end coherently converts the electromagnetic pulse train into a baseband signal in one stage. The impulse radio receiver integrates a number of pulses to restore each bit of transmitted information. II.1 Gauss Monocycle The most basic element of impulse radio technology is the Gauss Monocycle. Technical Foundation This section relates to the technical foundation and provides the reader with an introduction to the concept of impulse radio and other relevant aspects of communication theory. This section contains sections related to Gaussian monocycle pulses, pulse trains of Gaussian monocycle pulses, modulation, coding, and the quantitative properties of these concepts. Impulse radio transmitters emit short Gauss monocycle pulses with tightly controlled average pulse intervals. Impulse radio transmitters use pulse widths between 20 and 0.1 nanoseconds (ns) and pulse intervals between 2 and 5000 nanoseconds. These narrow monocycle pulses inherently have wideband frequency characteristics. The impulse radio system uses pulse position modulation, and the actual pulse interval varies from pulse to pulse due to two components, the information element and the pseudo-random code element. Unlike diffusion spectrum systems, pseudo-random codes are not required for energy diffusion (because the impulse itself is wideband in nature), channelization, energy smoothing in the frequency domain, and anti-jamming. Needed for sex. The impulse radio receiver is a homodyne receiver with a cross-correlator front end. The front end coherently converts the electromagnetic pulse train into a baseband signal in one stage. The impulse radio receiver integrates a number of pulses to restore each bit of transmitted information. II.1 Gauss Monocycle The most basic element of impulse radio technology is the Gauss Monocycle. Use pulse widths between 1 nanoseconds (ns) and pulse intervals between 2 and 5000 nanoseconds. These narrow monocycle pulses inherently have wideband frequency characteristics. The impulse radio system uses pulse position modulation, and the actual pulse interval varies from pulse to pulse due to two components, the information element and the pseudo-random code element. Unlike diffusion spectrum systems, pseudo-random codes are not required for energy diffusion (because the impulse itself is wideband in nature), channelization, energy smoothing in the frequency domain, and anti-jamming. Needed for sex. The impulse radio receiver is a homodyne receiver with a cross-correlator front end. The front end coherently converts the electromagnetic pulse train into a baseband signal in one stage. The impulse radio receiver integrates a number of pulses to restore each bit of transmitted information. II.1 Gauss Monocycle The most basic element of impulse radio technology is the Gauss Monocycle. Use pulse widths between 1 nanoseconds (ns) and pulse intervals between 2 and 5000 nanoseconds. These narrow monocycle pulses inherently have wideband frequency characteristics. The impulse radio system uses pulse position modulation, and the actual pulse interval varies from pulse to pulse due to two components, the information element and the pseudo-random code element. Unlike diffusion spectrum systems, pseudo-random codes are not required for energy diffusion (because the impulse itself is wideband in nature), channelization, energy smoothing in the frequency domain, and anti-jamming. Needed for sex. The impulse radio receiver is a homodyne receiver with a cross-correlator front end. The front end coherently converts the electromagnetic pulse train into a baseband signal in one stage. The impulse radio receiver integrates a number of pulses to restore each bit of transmitted information. II.1 Gauss Monocycle The most basic element of impulse radio technology is the Gauss Monocycle.<u style="single">It is a practical practice (referred to herein as a Gauss monocycle pulse).</u>.. The Gaussian monocycle is the first derivative of the Gaussian function. Figures 1A and 1B show monocycle pulses with a center frequency of 2 GHz (ie, pulse width 0.5 nanoseconds) in the time and frequency domains (see 102 and 104, respectively). (<u style="single">In practical implementation</u>Complete gauss monocycle<u style="single">Transmission is interrupted</u>.. In the frequency domain, this results in a slight reduction in signal bandwidth.) These monocycles are sometimes referred to as impulses.<u style="single">Not a gated sine wave</u>.. The Gauss monocycle waveform is essentially a wide bandwidth signal, with center frequency and bandwidth completely dependent on pulse width. In the time domain, the Gauss monocycle is mathematically described by the following equation:<img file="JP3781428B2_D0001.tif" />Where A is the peak amplitude of the pulse, t is the time, τ is the time delay constant frequency domain, and the Gauss monocycle envelope is<img file="JP3781428B2_D0002.tif" />Therefore, the center frequency is<img file="JP3781428B2_D0003.tif" />Regarding c, the point (power) that decreased by 3 dB is<img file="JP3781428B2_D0004.tif" />That is, the bandwidth is about 160% of the center frequency. Since τ also defines the pulse width, the pulse width specifies both the center frequency and the bandwidth. In practice, the center frequency of a monocycle pulse is roughly this length<u style="single">Reciprocal</u>The bandwidth of this is roughly equal to 1.6 times the center frequency. Therefore, with the "0.5 nanosecond" pulse illustrated in Figures 1A and 1B,<img file="JP3781428B2_D0005.tif" />Will be. II.2 Pulse trains Impulse radio systems use pulse trains for communication rather than signal pulses. As described in detail in Part III, the impulse radio transmitter generates and outputs a pulse train for each bit of information. The inventor-created prototype has a pulse repetition frequency between 0.7 and 10 megapulses per second (mpps, where each megapulse is 10 6 pulses). Figures 2A and 2B are drawings of a 1mpps system with 1 nanosecond pulses (uncoded and unmodulated) in the time and frequency domains (see 102 and 104, respectively). In the frequency domain, this high constant pulse train produces energy spikes (comb-shaped lines 204) at 1 MHz intervals. That is, low power is already diffused between the comb-shaped wires 204. This pulse train does not transmit information and can interfere with conventional radio systems over short distances due to the stationarity of the energy spikes. Impulse radio systems have very low duty cycles and the average power time domain is significantly lower than the peak power in the time domain. In the examples of Figures 2A and 2B, for example, the impulse transmitter operates at 0.1% of the time (ie, 1 nanosecond per microsecond (μs)). Further processing is required to modulate the pulse train so that the impulse radio system can actually communicate information. The additional processing smoothes the energy distribution in the frequency domain so that impulse radio transmissions (eg signals) interfere minimally with traditional radio systems. II.3 Modulation amplitude and frequency / phase modulation are not suitable for this particular form of impulse communication. The only suitable choice is pulse position modulation, which allows the receiver to use a matched filter (ie cross-correlator). As illustrated in FIG. 3, the modulated signal is in proportion to the modulation with a pulse repetition interval (PRI :). pulse repetition interval) is changed. If the modulated signal has three levels, the first level shifts the pulse generation forward by δ picoseconds (ps) from the nominal position. The second level does not shift the pulse position from the nominal position at all. The third level delays the pulse by δps. This will be a digital modulation scheme. Analog modulation allows continuous shifts between PRI-δ and PRI + δ. In an impulse radio system, the maximum value of δ is t / 4, where t is the pulse time. In the frequency domain, pulse position modulation distributes energy over many frequencies. For example, in the case of a 1mpps system, if the modulation dither (d) is 100ps, the PRI is 1,000,000 hertz (Hz), and the additional frequency components are 999,800.04Hz, 999,900.01Hz, 1,000,100.01Hz, 1,000,200.04Hz. (Dither is a term for impulse radio communication that moves the pulse position in time.) Transmission energy is distributed between many spikes (comb-shaped lines) in the frequency domain. If the total transmit energy remains constant, the energy at each frequency spike decreases as the number of possible pulse positions increases, i.e., the energy disperses more smoothly in the frequency domain. II.4 Coding for energy smoothing and channelization Since the receiver is a cross-correlator, the amount of time position modulation required for 100% modulation is calculated as the reciprocal of fc / 4 (fc is the center frequency). ). Center frequency 1. In a 3GHz monocycle, for example, this corresponds to ± 157 (ps) time position modulation. The spectral smoothing effect at this level of time dither is negligible. Impulse radio achieves optimum smoothing by applying PN code dither to each pulse, which is much larger than the modulation dither. Figure 4 is a plot showing the effect of pseudo-random dither on the energy distribution in the frequency domain. FIG. 4 shows the effect of using the 256-position PN code on the uncoded signal when compared with FIG. 2B. PN dithering also provides channelization (channelization is a technique used to divide a communication path into multiple channels). In uncoded systems, it is very difficult to distinguish between independent transmitters. The PN code creates a channel when the code itself is relatively orthogonal (ie, low correlation and / or low interference between the codes used). II.5 Reception and Demodulation Obviously, mutual interference can occur when a large number of impulse radio users are present in a limited area. In addition, the PN code<u style="single">To</u>Reduces interference as the number of users increases, but individual pulses from one user's sequence are received at the same time as pulses from another user's sequence.<u style="single">probability</u>Will increase. Well, the realization of impulse radios according to the present invention does not depend on the reception of all pulses. Impulse radio receivers perform correlative, synchronous reception (at RF level) using statistical sampling of multiple pulses to restore transmitted information. Impulse radio receivers typically integrate 200 or more pulses to extract demodulated output. The optimum number of pulses that the receiver integrates depends on a number of variables, including pulse rate, bit rate, jamming level, and range. II.6 Interference Resistance Apart from channelization and energy smoothing, PN coding increases the resistance of impulse radio to interference from all radio communication systems, including other impulse radio systems. This is essential when some other signal acts as a source of interference to the impulse radio within the band occupied by the impulse signal. There is an unallocated band above 1GHz available in the impulse system<u style="single">do not do</u>Therefore, it is necessary to share the spectrum with other conventional and impulse radios without harmful effects. The PN code assists the impulse system in distinguishing between the intended transmission of impulses and transmissions from others. FIG. 5 shows the result of the narrow band sinusoidal interference signal 502 superimposed on the impulse radio signal 504. With an impulse radio receiver,<u style="single">Mutual</u>The inputs to the correlator include this narrowband signal 502 as well as the received ultra wideband impulse radio signal 504. Without PN coding, the cross-correlator would sample the jamming signal 502 in a stationary manner such that the jamming signal would cause significant damage to the impulse radio receiver. However, when the transmitted impulse signal is encoded with PN code dither (and the impulse radio receiver is synchronized with the same PN code dither), the jamming signal is randomly sampled. According to the present invention, integration over a large number of pulses cancels out the effects of interference. Statistically, temporal pseudo-randomization of receive processing produces a stream of randomly distributed values with a mean of zero (for interfering signals). All that is required to eliminate the effects of the interfering source is to sample over a sufficient number of pulses (ie, integrate over a sufficiently large number of pulses) to bring the effect of the interfering signal close to zero. II.7 Processing gain Impulse radio is anti-jamming due to its large processing gain<u style="single">There is</u>.. In a diffuse spectrum system, the definition of processing gain is the ratio of the bandwidth of a channel to the bandwidth of an information signal, which quantifies the reduction of channel interference when using wideband communication. For example, a direct sequence diffusion spectrum system with an information bandwidth of 10 kHz and a channel bandwidth of 16 MHz provides a processing gain of 1600 or 32 dB. However, an impulse radio system with the same 10 kHz information bandwidth and 2 GHz channel bandwidth provides even greater processing gain, with a processing gain of 200,000 or 53 dB. A duty cycle (eg 0.5%) provides a processing gain of 28.3 dB. (Processing gain is generally the ratio of the bandwidth of the received signal to the hand width of the received information signal) 28.3 dB for effective oversampling to restore information from integration over multiple pulses (eg, integration over 200 pulses) A processing gain is obtained. That is, dividing 2 GHz by a 10 mpps link transmitting 50 kilobits per second (kbps) yields a processing gain of 49 dB (ie, dividing a pulse width of 0.5 nanoseconds by a pulse repetition interval of 100 nanoseconds gives a duty cycle of 0.5%. Is obtained, and dividing 10 mpps by 50,000 bps gives 200 pulses per bit). II. Theoretical analysis suggests that an 8-capacity impulse radio system can have thousands of audio channels per cell. Understanding the capacitance of an impulse radio system requires careful examination of the capabilities of the cross-correlator. Figure 6 shows the cross correlater transfer function 602. This represents the output value of the impulse radio receiver cross-correlator for any received pulse. As illustrated in 604, the output of the cross-correlator is 0 volts when the pulse arrives outside the cross-correlation window 606. As the receive pulse 608 slides through the window, the cross-correlator output changes. It has a maximum value when the pulse is τ / 4 ahead of the center of the window (eg 1 volt) and a minimum value when it is τ / 4 behind the center (eg -1 volt). When the intended transmitter and system are in sync, the cross-correlator output has an amplitude between ± 1 volt (as a function of transmitter modulation). Other in-band transmissions cause changes in the output value of the cross-correlator. This change is a random variable and can be modeled as a Gaussian white noise signal with a mean of 0. As the number of interferences increases, the change increases linearly. By integrating over a large number of pulses, the receiver produces an estimate of the modulation value of the transmitted signal. In other words function) 602 is shown. This represents the output value of the impulse radio receiver cross-correlator for any received pulse. As illustrated in 604, the output of the cross-correlator is 0 volts when the pulse arrives outside the cross-correlation window 606. As the receive pulse 608 slides through the window, the cross-correlator output changes. It has a maximum value when the pulse is τ / 4 ahead of the center of the window (eg 1 volt) and a minimum value when it is τ / 4 behind the center (eg -1 volt). When the intended transmitter and system are in sync, the cross-correlator output has an amplitude between ± 1 volt (as a function of transmitter modulation). Other in-band transmissions cause changes in the output value of the cross-correlator. This change is a random variable and can be modeled as a Gaussian white noise signal with a mean of 0. As the number of interferences increases, the change increases linearly. By integrating over a large number of pulses, the receiver produces an estimate of the modulation value of the transmitted signal. In other words function) 602 is shown. This represents the output value of the impulse radio receiver cross-correlator for any received pulse. As illustrated in 604, the output of the cross-correlator is 0 volts when the pulse arrives outside the cross-correlation window 606. As the receive pulse 608 slides through the window, the cross-correlator output changes. It has a maximum value when the pulse is τ / 4 ahead of the center of the window (eg 1 volt) and a minimum value when it is τ / 4 behind the center (eg -1 volt). When the intended transmitter and system are in sync, the cross-correlator output has an amplitude between ± 1 volt (as a function of transmitter modulation). Other in-band transmissions cause changes in the output value of the cross-correlator. This change is a random variable and can be modeled as a Gaussian white noise signal with a mean of 0. As the number of interferences increases, the change increases linearly. By integrating over a large number of pulses, the receiver produces an estimate of the modulation value of the transmitted signal. In other words<img file="JP3781428B2_D0006.tif" />Where N = number of interferences, σ is the total interference for a single cross-correlation<u style="single">Distributed</u>, Z is the number of pulses for the receiver to integrate and restore modulation. This is a good relationship in communication systems as the number of concurrent users increases, with gradual (rather than abrupt) deterioration in link quality. II.9 Multipath fading, which is a problem for multipath and propagating sinusoidal systems, is a significantly smaller (several orders of magnitude smaller) problem for impulse systems than for traditional radio systems. In fact, the remarkable Rayleigh fading in cellular communication is a continuous wave phenomenon, not an impulse communication phenomenon. Special conditions must be met for the multipath effect to appear in an impulse radio system. The path length through which the dispersed pulses pass must be less than the speed of light x pulse width, and / or the pulses radiated continuously (as a sequence) at the transmitter must arrive at the receiver at the same time. For the former of 1 nanosecond pulse, it is equal to 0.3 meters or 1 foot (ie 1 nanosecond x 300,000, 000m / sec). (See Figure 7. In this case, the pulse passing through "pass 1" arrives half the pulse width after the pulse in the direct path). In the latter case of 1 megapulse per second, it is equivalent to transmitting extra 300, 600, 900 meters, etc. However, since each of the separate pulses undergoes pseudo-random dither, these pulses do not correlate. The pulses propagating between these intervals do not cause self-interference (figure 7 in pulse transmission path 2). On the other hand, the most elongated ellipse, as shown in Figure 7, the pulse travels along the grazing path, creating the multipath effect of impulse radio. As illustrated in 802 of FIG. 8, as the multipath pulse travels further by half the pulse width, it increases the power level of the received signal (the phase of the multipath pulse is inverted by the reflective surface). .. Further progression of the pulse by less than half the pulse width creates destructive interference, as shown in 804. For example, for a 1 nanosecond pulse, destructive interference occurs as the multipath pulse travels between 0 and 15 centimeters (0 to 6 inches). Impulse radio system testing (including impulse radar testing) suggests that multipath does not pose a major problem in production. In addition, shorter pulse widths are expected, in which case the probability of destructive interference is further reduced (because the reflection path length required for destructive interference is shorter). III. Invention of Subcarriers This section relates to the use of subcarriers in impulse radio communication systems. This section contains subsections related to the theory of operation of subcarriers in impulse radio transmitters and receivers. The description is divided to explain an example of one channel improved with respect to baseband alone and an example of two or more subcarrier channels. III.1 Behavior theory According to the present invention, additional channels<u style="single">To</u>Developed to include one or more subcarriers for smoothing, and fidelity. The following ultra wideband time domain impulse radio communication architecture operates according to the general impulse radio theory described in Part II. The following three specific embodiments, a one-channel system, a two-channel system, and three or more channel systems will be described. The examples of the three impulse radio receivers described below are used as examples for explaining the present invention, not as a limitation, and those skilled in the art can create and use the present invention. These technologies include at least the areas of communications, discrete analog, digital, and integrated circuit design and implementation, digital signal processing, and PN code theory. The realization of various elements and blocks will be apparent to those skilled in the art. III.2 Baseband Alone Improvements on 1 Channel This section describes an impulse radio communication architecture that uses a 1 channel subcarrier channel with improved baseband performance alone. Radio frequency (RF: radio) according to the present invention frequency) Examples are the most common. Applications of typical RF impulse radio systems include cellular phone phones, radiotelephones, wireless PBX / local area networks, and more. The propagation of RF impulse radio signals, defined as the process by which the signal travels from transmitter to receiver, is typically from the transmitting antenna to the receiving antenna via air or space. This is considered as a radio RF impulse radio. Suitable antennas for impulse radios are fully described in US Patent Application No. 07 / 368,831. However, the present invention is also suitable for transmission via a coaxial cable. In this embodiment, transmit and receive antennas are excluded. A typical block diagram of an impulse radio electrical system using one subcarrier channel is illustrated in FIG. Transmitter 901 and receiver 903 using a single subcarrier ultra wideband impulse radio channel are illustrated. The transmitter 901 and the receiver 903 are separated by a propagation medium 905, such as an air, space, or other medium capable of propagating ultra-wideband signals. The transmitted impulse radio signal 907 propagates from the transmitter 901 to the receiver 903 via the propagation medium 905. III.2.a Transmitter A preferred embodiment of an impulse radio transmitter in an impulse radio communication system having one subcarrier channel will be described with reference to FIG. Transmitter 901 includes a time-based 1002 that produces a periodic timing signal 1004. Time-based 1002 has a high timing of about picoseconds<u style="single">accuracy</u>Includes a voltage controlled oscillator circuit or similar circuit with. The voltage control that regulates the VCO center frequency is set during calibration to the desired center frequency used to define the transmitter's undivided pulse repetition rate. The periodic timing signal 1004 is supplied to the code source 1006 and the code time modulator 1008. The code source 1006 includes storage devices such as random access memory (RAM) and read-only memory (ROM) for storing the orthogonal PN code and outputting the PN code as the code signal 1010. Alternatively, the maximum length shift register can be used to generate the PN code. The code source 1006 can monitor the periodic timing signal 1004 and synchronize the code signal 1010 with the code time modulator 1008. The code time modulator 1008 uses the code signal 1010 to modulate the periodic timing signal 1004 for channelization and smoothing of the signal 1012 that is finally emitted. The output of the code time modulator 1008 is called the coded timing signal 1014. The coded timing signal 1014 is supplied to the subcarrier time modulator 1016 for information modulation thereof. In conventional impulse systems, information modulation is performed using the information itself as the modulation source. However, in the present invention, the information source 1018 supplies the information signal 1020 to the subcarrier generator and modulator 1022. The information signal 1020 can be some kind of intelligence, including digital bits representing audio, data, images, etc., analog signals, or synthetic signals. Both the encoded timing signal 1014 and the subcarrier time modulator 1016 can be implemented using a voltage, current, or digital source as the modulation input, as evidenced by those skilled in the art. As defined by Dixon, a subcarrier is "a carrier that is modulated with information independent of carrier modulation and modulates the carrier." Subcarrier generator and modulator 102 of the present invention 2 generates a modulated subcarrier signal 1024 modulated by the information signal 1020, and supplies the modulated subcarrier signal 1024 to the subcarrier time modulator 1016. That is, the modulated subcarrier signal 1024, in this case, is the subcarrier time modulator 1016 to modulate the carrier which is the encoded timing signal 1014.<u style="single">Used by</u>.. Modulation of the coded timing signal 1014 by the subcarrier time modulator 1016 produces a modulated coded timing signal 1026, which is sent to the output stage 1028. The output stage 1028 uses the modulation-encoded timing signal 1026 as a trigger to generate an electrical monocycle pulse. The electrical monocycle pulse is transmitted to the transmitting antenna 1030 via the transmitting line 1032. The electrical monocycle pulse is transformed into an electromagnetic pulse propagated by the transmitting antenna 1030. In this embodiment, the electromagnetic pulse is called a radiation signal 1012, and in the radio frequency embodiment, it propagates to an impulse radio receiver (not shown) via a propagation medium 905 such as air. In a preferred embodiment, the radiation signal 1012 is a wideband or ultra wideband signal. However, the radiated signal 1012 has a spectrum due to monocycle pulse filtering.<u style="single">Can be changed</u>.. With this bandpass filtering, each monocycle pulse will have many zero crossings in the time domain. In this case, the impulse radio receiver needs to use a similar waveform in the cross-correlator to be effective. The addition of the subcarrier generator and modulator "stage" 1022 to transmitter 901 has many advantages. Information signal-modulated subcarriers provide additional channelization and smoothing to the system, allowing the addition of many new and separate impulse radio channels. The addition of subcarriers provides more fidelity to the information signal 1020 in the form of additional bandwidth and better signal-to-noise ratio compared to baseband modulation alone. The use of subcarriers in impulse radio is a non-linear modulation transfer function compared to baseband modulation. Harmonic distortion due to function) is also improved. The nonlinear modulation transfer function is described below in relation to the cross-correlation processing performed by the impulse radio receiver. Since impulse radio is not a CW carrier system, the use of subcarriers is an elegant and unintuitive addition to the time domain impulse radio design. The signal-to-noise ratio is improved by 5 to 20 dB (depending on the signal-to-noise ratio of the narrow pulse modulated carrier) compared to non-subcarrier impulse radio transmission. Using subcarriers on impulse radio translates (or shifts) the baseband signal to a higher frequency. In a preferred embodiment, the subcarrier generator and modulator 1022 are frequency modulation (FM), amplitude modulation (AM), phase modulation, frequency shift keying (FSK), phase shift keying (PSK). It is modulated by the information signal 1020 by keying), pulse FM, and other techniques. In another embodiment, direct digital modulation is used in the subcarrier technology. In this other embodiment, Manchester coding of digital data produces a modulated subcarrier signal of 1024. The subcarrier time modulator 1016 uses the modulated subcarrier signal 1024 to perform pulse position modulation of the coded timing signal 1014. Other non-sinusoidal and / or discontinuous waveforms can also be used as subcarriers in the context of the present invention. The modulated subcarrier signal 1024 is used by the subcarrier time modulator 1016 to time shift the pulse position of the coded timing signal 1014. That is, the output<u style="single">Step</u>Signal to trigger<u style="single">(In this case, the modulation-coded timing signal 1026)</u>Is a pulse position-modulated pulse train. Subcarriers with different frequencies or waveforms can be used to add channelization of impulse radio signals. That is, the impulse radio link can communicate on a large number of independent channels at the same time by using a different subcarrier for each channel. To demonstrate this, consider a pair of independent impulse radio users operating on the same PN code. The first user pair communicates on an impulse radio with a subcarrier generator and modulator 1022 that generate one sinusoidal subcarrier of the first discrete frequency. The second user pair communicates on an impulse radio with a subcarrier generator and modulator 1022 that generate one sinusoidal subcarrier of the second discrete frequency, separate from the first frequency. Each user pair can have communications isolated from each other by setting up two sets of impulse radios (as described below) to reproduce only the information transmitted at the appropriate subcarrier frequency. For this illustration, more impulse radio channels are available by using the impulse radio of carrier technology. Otherwise, the two pairs of impulse radio users have isolated communication if each pair uses the same subcarrier as a different PN code. In addition, channelization can be achieved by operating the radio set independently of the PN code and / or subcarriers at a different pulse repetition rate. The result of the new subcarrier stage is the expansion of information channel fidelity. This advantage is attributed to the fact that subcarriers inherently make information noise resistant. The following III. As described in detail in section 2 (b), a template signal with a waveform designed to match the monocycle pulse received by the impulse radio receiver is generated. The template signal is temporally positioned according to the transmitter's known PN code and cross-correlates with the received impulse radio signal. Integrate the cross-correlation output to restore a noise-free impulse radio signal. After taking out in this way, the signal is demodulated to exclude subcarriers and an information signal is obtained. Another embodiment of the impulse radio transmitter according to the present invention is illustrated in FIG. In this embodiment, the code time modulator 1008 and the subcarrier time modulator 1016 are reversed. As illustrated in FIG. 11, the information source 1018 outputs the information signal 1020 to the subcarrier generator and modulator 1022. Further, the subcarrier generator and the modulator 1022 output the modulated subcarrier signal 1024 to the subcarrier time modulator 1016. The subcarrier time modulator 1016 uses the modulated subcarrier signal 1024 to time position-modulate the periodic timing signal 1004 to generate the modulation timing signal 1140. Any of the subcarrier modulation techniques described above in connection with FIG. 10 can be used. The code source 1006 receives the periodic timing signal 1004 for synchronization and outputs the code signal 1010 to the code time modulator 1008. The code time modulator 1008 further time-position-modulates the modulation timing signal 1140 using the code signal 1010, and outputs the modulation-encoded timing signal 1142. The modulation-coded timing signal 1142 illustrated in FIG. 11 is provided to the output stage 1028 in a manner similar to that of the embodiment illustrated in FIG. As already described in the context of FIG. 10, the impulse radio transmitter outputs the radiation signal 1012. The above description of FIG. 11 is an example of many changes that can be made to an impulse radio transmitter to provide the necessary coding and subcarrier modulation for the signal intended to be transmitted through the impulse radio transmitter. .. Figure 10 and The above embodiment described in connection with FIG. 11 is shown as an example and is not limited thereto. A configuration similar to the block diagram of FIGS. 10 and 11 of the impulse radio transmitter will be apparent to those skilled in the art based on the above disclosure without departing from the scope of the present invention. An example of another transmitter is illustrated in FIG. In this embodiment, the adder 1202 or a similar circuit is used to add the code signal 1010 and the information modulation subcarrier signal 1204. The adder 1202 outputs the code modulation subcarrier signal 1206 to the code timing modulator 1208. The code timing modulator 1208 performs the functions of the code time modulator and subcarrier time modulator 1016 of FIG. The code timing modulator 1208 modulates the periodic timing signal 1004 using the code modulation subcarrier signal 1206 to generate a modulation encoded timing signal 1026. The remaining elements of the receiver in Figure 12 behave as discussed in connection with Figure 10. Any of the subcarrier modulation techniques described above in the context of FIG. 10 can be used. In yet another embodiment, the modulation can be done by directly modulating the code signal 1010 with the information signal 1020. This is illustrated in FIG. The adder 1202 is configured to modulate (add) the code signal 1010 and the information signal 1020 to generate the modulated signal 1302. The code timing modulator 1208 modulates the periodic timing signal 1004 using the modulation signal 1302 to generate a modulation-encoded timing signal 1026. The remaining elements of the receiver of FIG. 13 behave as discussed in connection with FIG. Subcarriers that are not modulated with information can also be used to modulate the coded timing signal, or the coded timing signal itself can be transmitted unmodulated. These two examples can be used to communicate the mere presence of an impulse radio, such as a beacon or transponder. Split another impulse radio unit into another PN code and another subcarrier Many operating applications can be realized by guessing. III.2. Receiver The impulse radio receiver 903 for a 1-channel subcarrier impulse radio communication system will be described with reference to FIG. The impulse radio receiver (hereinafter referred to as the receiver) 1400 includes a receiving antenna 1402 for receiving the propagating impulse radio signal 1404. The received signal 1406 is input to the cross-correlator 1408 via the receiver transmission line 1410 connected to the receiving antenna 1402. Receiver 1400 also includes a decode source 1410 and an adjustable time base 1414. The decode source 1410 produces a decode control signal 1412 corresponding to the PN code used by the associated impulse radio transmitter (not shown) that transmitted the propagated signal 1404. The adjustable time base 1414 produces a periodic timing signal 1416 containing a pulse sequence of template signals having a waveform substantially equivalent to each pulse of the received signal 1406. Each pulse of the received signal 1406 is similar to the derivative of a Gauss monocycle pulse. FIG. 15 shows a representative plot of pulse 1502 corresponding to signal 1406 received in relation to receiver 1400. The pulse 1502 corresponds to a radiation signal (monocycle pulse) having a waveform similar to that of the pulse 302 in FIG. When an electromagnetic monocycle pulse having a waveform similar to that of the pulse 302 is incident on the receiving antenna 1402, the receiving antenna has the unique property that the electrical waveform appearing at the output has the shape of the pulse 1502. When the impulse radio antenna is inverted, the pulse 1502 becomes the inverted voltage. FIG. 16 shows the cross-correlation process. FIG. 16 shows the waveform of the template signal pulse 1602 and the waveform of the (impulse radio pulse) signal 1406 received at the time increment Δt. Curve 1604 is not a continuous waveform, but the received signal 1406 slides from the lock by the template signal pulse 1602 to represent the correlated voltage obtained in each Δt time array. Each Δt of the received signal 1406 is a pal Note that it is an inverting voltage when compared to the 1502). The temporal position of the template signal pulse used to correlate with the received signal 1406 is set by the decode timing modulator 1418. The effect of using a cross-correlation function as the modulation transfer function is to make the receiver output a non-linear function of the input amplitude. For baseband modulation, this is not desirable. However, with subcarriers such as FM, PSK, FSK, and Manchester, the harmonics can be easily filtered, eliminating any distortion. In such filtering, when baseband modulation is used, the harmonics remain in the baseband, so that the harmonics cannot be removed and the signal cannot be restored. Returning to FIG. 14, the decode control signal 1412 and the periodic timing signal 1416 are received by the decode timing modulator 1418. The decode timing modulator 1418 uses the decode control signal 1412 to temporally arrange the periodic timing signal 1416 to generate the decode signal 1420. The decoded signal 1420 timely matches the known PN code of the transmitter, allowing the received signal 1406 to be detected by the cross-correlator 1408. The detection process performed by the cross-correlator 1408 includes a decoded signal 1420 and a received signal 1406 cross-correlation operation. The integration of the cross-correlation over time produces the baseband signal 1422. II. As already discussed in Section A, the impulse radio signal is extracted from the noise by integrating the cross-correlation signal over time. In this embodiment, the baseband signal 1422 is demodulated by the subcarrier decoder 1424 to remove the subcarriers and obtain the demodulated information signal 1426. The demodulated information signal 1426 is substantially identical to the transmitter information signal (see 1018 in Figure 10). The baseband signal 1422 is also input to the lowpass filter 1428. A control loop 1429, including a lowpass filter 1428, generates an error signal 1430 to provide minor phase adjustments to the adjustable time base 1414 to temporally position the periodic timing signal 1416 relative to the position of the received signal 1406. Use. Subcarrier examples provide less signal compression and lower signal distortion by reducing baseband noise for highly reliable voice, data, and / or image communication. The linearity requirement for modulation using a cross-correlator is significantly relaxed by using the subcarrier technique of the present invention. Modulation transfer characteristics need to be extremely linear in order to successfully transmit low distortion speech or music. This is very difficult to achieve with non-subcarrier baseband impulse systems. Information signals are easily disturbed by noise Most of the noise is concentrated in the baseband and decreases as the frequency increases up to the Nyquist frequency. For example, 1. For impulse radios using a 4 megapulse rate, the Nyquist frequency is about 700 kHz. In this example, subcarriers up to about 700 kHz can be used to make the impulse radio system substantially noise resistant. The FM subcarrier embodiment uses a phase locked loop (PLL) frequency decoder. The characteristics of the phase-locked loop determine the bandwidth capture and other basic aspects of the received signal. An optional bandpass filter can be used in series prior to the phase-locked loop to narrow the spectrum of demodulation performed in the phase-locked loop. III.3 2 or more subcarrier channels (eg voice, digital data, and control information) A major advantage of the subcarrier impulse radio of the present invention is that many subcarriers have the same code for simultaneous transmission. It is possible to pack it in the conversion timing signal. Examples of three subcarriers in one impulse radio ultra wideband transmission are illustrated in FIGS. 17-19 for both analog and digital implementations. FIG. 17 shows a schematic representation of an impulse radio transmitter with three subcarrier generators / modulators (SC GEN / MOD) 1702, 1704, 1706, each with a different subcarrier frequency. The basic architecture of the transmitter is based on the example in Figure 10. For example, the main subcarrier generator / modulator 1720 (shown with a dashed line) is similar to the subcarrier generator / modulator 1022. However, this example can be modified to work with any of the transmitters and their equivalents disclosed above. Audio source (VIS: voice information source) 1708 goes to the subcarrier generator / modulator (abbreviated as SC GEN / MOD in Figure 17) 1702 via line 1722 to modulate the first subcarrier signal (not shown). Be supplied. The first subcarrier signal is generated internally or externally by the subcarrier generator / modulator 1702 and supplied as an input to the main subcarrier generator / modulator 1720. Similarly, digital data sources such as modem output or fax transmission (DDS: digital data source) 1710 is a second subcarrier generator / modulator (abbreviated as SC GEN / MOD in Figure 17) via line (or bus) 1724 for modulation of the second subcarrier signal 1704. Is supplied to. Finally, the Digital Control Information Source (CIS) 1712 is a third subcarrier generator / modulator (abbreviated as SC GEN / MOD in Figure 17) via line (or bus) 1726 for modulation of the third subcarrier signal. It is supplied to 1706. The second and third subcarrier signals are generated by the subcarrier generators / modulators 1704 and 1706, respectively, or supplied externally as inputs to the subcarrier generator / modulator 1720. The digital CIS1712 provides control information to the impulse radio receiver. In cellular phone transceiver type systems, such digital control information is routing information, scheduling information, bell signal (ring). signal) etc. can be included. Virtually any kind of control signal, or intelligence about it, can be used to modulate the subcarrier signal. The three modulated subcarrier signals are output from the three subcarrier generators / modulators 1702, 1704, 1706 via lines 1728, 1730, 1732 and added by the adder circuit 1714. The resulting signal 1716 is sent to the subcarrier time modulator 1016, where it is used to modulate the coded timing signal 1014 to generate a modulated coded timing signal 1026. The modulation-encoded timing signal 1026 output from the subcarrier time modulator 1016 is supplied to the output stage 1028 and transmitted as a radiation signal 1012 as described above. Two typical multi-carrier channel impulse radio receivers are illustrated in FIGS. 18 and 19. Each receiver has a component for demodulating three subcarrier channels, for example transmitted from the transmitter of FIG. The basic architecture of the receivers of FIGS. 18 and 19 is based on the embodiment of FIG. 14 or its equivalent. FIG. 18 illustrates the cross-correlator 1408 followed by multiple analog FM demodulation branches in a typical analog embodiment. The cross-correlated baseband signal 1422 is generated from the received signal 1406, as discussed in the context of FIG. 14 (using control loop elements, not shown in FIGS. 18 and 19). Each branch is a bandpass filter 1802 (eg LC or switched capacitor) Demodulate one subcarrier using filter)) and phase locked loop block 1804. In other words, the three independent and simultaneously transmitted information signals are restored and can be used by OUTPUT1 ~ 3. In the digital embodiment illustrated in FIG. 19, the cross-correlated baseband signal 1422 is converted to a digital signal using an analog-to-digital converter (A / DC) 1902. Encoded into signal 1903 using a digital signal processor (DSP) 1904, such as model number TMS320C40 (Texas Instruments, Dallas, Texas), and a known digital signal processor algorithm that uses a Fourier transform, etc. Digitally demodulate the three separate subcarriers. The digitally demodulated information is reconverted to analog using a digital-to-analog converter (D / AC) 1906. The audio signal is reconverted to the analog compatible part using the digital-to-analog converter 1906 and becomes available in OUTPUT1. The digital data signal will be output or otherwise available directly from the digital signal processor at OUTPUT2. Finally, the control signal is output or otherwise available directly from the digital signal processor at OUTPUT3 or after digital-to-analog conversion with the digital-to-analog converter 1906. The addition of multiple subcarriers does not affect the wideband characteristics of the impulse radio signal. IV. Time Modulator This section relates to code time delays, subcarrier time delays, and time modulators used for combinations of both. The operation and structure of some embodiments for using the time modulator for subcarrier impulse radio communication will be described. According to various embodiments of the present invention, the impulse radio transmitter includes a cord time modulator (eg 1008) and a subcarrier time modulator (eg 1016), as well as a cord and timing modulator (eg 1208). Each of these modulators has the function of delaying the signal (eg, the periodic timing signal 1004) in time according to the information transmitted by the trigger signal (eg, the code signal 1010 or the modulated subcarrier signal 1024). That is, each modulator (eg 1008, 1016, or 1208) is considered a delay generator. Delay generators that have a numeric input signal are called binary-to-time delay generators. Binary-time delay generators can be implemented using commercially available ICs currently available. A suitable delay generator with a numerical input is the MC100E196ECL (emitter-coupled logic) device manufactured by Motorola, Schaumburg, Illinois. However, in connection with the impulse radio signal according to the present invention, such a conventional binary-time delay generator provides an accurate time delay such that the impulse radio signal can be accurately restored in the impulse radio receiver. Not provided. In other words, a time delay of about 157 ps (picoseconds), which is a typical pulse duration of a monocycle pulse, cannot be accurately generated by using a conventional binary-time delay generator. V. Linearization This section relates to the linearization of time modulators on both impulse radio transmitters and receivers. The linearization of the time modulator allows the impulse radio transmitter and receiver to generate a time delay with the accuracy required for impulse radio communication. To solve the time delay problem described in Part IV, the inventor of the present invention performed a statistical analysis of the specifications (eg, performance curves) provided by the binary-time delay manufacturer. Based on this work, the inventor found that if the non-linear behavior of the device was known, the non-linear behavior of conventional binary-time delay generators could be compensated. That is, according to a further aspect of the invention, the impulse radio transmitter is a linearized reference read-only memory (ROM) to compensate for any non-linearity in the context of a conventional binary-time delay generator (Figure). Includes (not shown). This allows the impulse radio transmitter to generate a time delay with a system well below the 157 picosecond requirement. FIG. 20 is a plot showing the delay time (picoseconds) and the binary (ie numerical) input values in a conventional binary-time delay generator. Curve 2002 shows an example of the exact time delay output characteristics of a conventional binary-time delay generator. The desired output of the binary-time delay generator used in the present invention is shown by curve 2004. For example, with a binary input value of 18, point 2010 on curve 2002 is of a traditional binary-time delay generator. An interim delay can occur. FIG. 20 is a plot showing the delay time (picoseconds) and the binary (ie numerical) input values in a conventional binary-time delay generator. Curve 2002 shows an example of the exact time delay output characteristics of a conventional binary-time delay generator. The desired output of the binary-time delay generator used in the present invention is shown by curve 2004. For example, with a binary input value of 18, point 2010 on curve 2002 is of a traditional binary-time delay generator. An interim delay can occur. FIG. 20 is a plot showing the delay time (picoseconds) and the binary (ie numerical) input values in a conventional binary-time delay generator. Curve 2002 shows an example of the exact time delay output characteristics of a conventional binary-time delay generator. The desired output of the binary-time delay generator used in the present invention is shown by curve 2004. For example, with a binary input value of 18, point 2010 on curve 2002 is of a traditional binary-time delay generator.<u style="single">Actual</u>Represents the output. Binary input value 10 is for the output of a traditional binary-time delay generator<u style="single">50</u>This is a typical input that causes a time delay of ps. However, given a numeric input value of 10, conventional binary-time delay generators desire, as shown at point 2006.<u style="single">50</u>It may only generate an actual output value of about 15ps, not ps. So in this example<u style="single">50</u>To generate the ps delay, enter the numeric input value 18 and the desired delay, as shown at point 2010 on curve 2002.<u style="single">50</u>Need to generate ps. It is generally desirable to linearize the dither generators of the transmitter and receiver, but in practice it is only necessary to have the same dither-to-numerical input mapping linearity, not necessarily straight lines. According to the present invention, a conventional binary-time delay generator using the type of linearized data illustrated in FIG.<u style="single">Actual</u>Map the response to the desired time delay. This linearized data, or mapping, is stored in linearized read-only memory (ROM). To transmit 1s and 0s, the pulse is time-modulated either forward or backward in time. In other words, an impulse radio signal intended to generate a logical value of 1 when received by an impulse radio receiver is timed slightly forward on the impulse radio transmitter. Impulse radio signals that are intended to be received as a logical value of 0 are time-shifted slightly backwards at the impulse radio transmitter. The cross-correlator 1408 of the impulse radio receiver converts its time position into a more positive or more negative voltage increment. Maximize the signal-to-noise ratio of the data stream using a bandpass data filter. The bandwidth of the bandpass data filter should be set to approximately half of the transmit baud rate, as will be apparent to those skilled in the art. The comparator transforms these voltages into logical equivalents of 1 and 0. The pulse must be supplied for both 1 and 0 because in the absence of the pulse, noise at the comparator threshold can produce a random output. The larger the separation (ie, voltage difference) between the positive and negative information between the samples, the better the signal-to-noise ratio and the lower the bit error rate. Since 1 and 0 shift the signal in time, the linearization ROM needs to store the independent linearization information of the impulse radio signal for the logical value 1 and the independent linearization information of the impulse radio signal for the logical value 0. There is. At a given information (data) transmission rate, the impulse radio transmission logic values 1 and 0 are shifted back and forth by a finite amount, respectively, and the cross-correlator of the impulse radio receiver is the logic value 1 in the data stream. Must be able to be correctly identified from the logical value 0. 1. For the center frequency selected with a 3 GHz monocycle pulse, the desired forward shift at logical value 1 and the backward shift at logical value 0 is a shift value of 157 ps. When the center frequency is doubled, the amount of time shift is halved. In other words, the linearized ROM stores one (8-bit) digital value that represents the linearized numerical value, and when outputting from the linearized ROM to the code time modulator 1408, a correct time shift of 157 ps can be realized. Must be done. In a preferred embodiment, the linearized ROM stores one 8-bit number for a 157ps forward shift and a second 8-bit number for a 157ps backward shift. In addition to the 157ps shift, the linearized ROM needs to further store 8-bit numbers for forward and backward time shifts to achieve forward and backward shifts of some other time shift. Note that if the transmitter uses zero time shift (nominal value) and twice 157ps (corresponding to digital values 0 and 1 respectively) as the modulation value, the demodulation receiver also refers to the same. V. 1 Transmitter Figure 21 is a high-level block diagram showing the above linearization scheme according to the present invention. However, in contrast to the modulated coded timing signal 1026 generated by, for example, the code time modulator 1008 of FIG. 10, the direct digital coded timing signal 2102 is code time modulated, as illustrated in the block diagram of FIG. Occurs in vessel 1008. In this embodiment, the time base 1002 outputs the periodic timing signal 1004 to the code source 1006. The periodic timing signal 1004 is also provided for the code time modulator 1008, which is a binary-time delay generator in this embodiment. In this embodiment, the code source 1006 includes an address counter 2104 and two read-only memories (ROMs) 2106, 2110. The synchronization timing signal 1004 increments the address counter 2104 so that the counter outputs the multi-bit address 2105. In this embodiment, the address counter 2104 outputs a 15-bit wide address 2105 for each pulse of the periodic timing signal 1004. The PN code ROM2106 is accessed using the address 2105 provided by the address counter 2104. ROM2106 has a predetermined modulo PN (pseudo-random noise: noise) Memorize the code. (Other memory devices such as EEPROM, RAM, shift register, etc. can be used.) Each address 2105 output from the address counter 2104 accesses the storage location of ROM 2106, and ROM responds to this. Outputs the PN code 2108 (preferably a 15-bit PN code). (As mentioned above, the PN code is the time forward or backward of the pulse (eg, periodic time signal pulse or digital data signal pulse) due to the linearization and diffusion of the monocycle pulse of the impulse radio signal. Used for position modulation). The linearized data is stored in a location that can be addressed by the linearized ROM 2110. The linearized data is accessed by applying an address (eg, a 16-bit address) to the address input of the linearized ROM 2110. According to preferred embodiments of the invention, the 16-bit address is, for example, a 15-bit PN code 2108 output from the PN code ROM 2106 and a 1-bit digital data source provided by the information source 1018 (1024 in FIG. 10). (Illustrated with a broken line 2107) similar to). Alternatively, digital data provided by information source 1018 can be used to modulate subcarriers using the subcarrier generator / modulator 1022 as described herein. In this case, the subcarrier generator / modulator 1022 will provide a 1-bit digital data signal (see solid line 2109) to the linearized ROM 2110. In response to simultaneous reception of the full input address (16 bits in this example), the linearized ROM 2110 outputs a linearized modulation timing signal 2112 (similar to 1206 in FIG. 12 and 1302 in FIG. 13). The linearized modulation timing signal 2112 is preferably 8-bit wide and is provided to the code time modulator 1008 (ie, binary-time delay generator). The code time modulator 1008 uses the linearized modulation timing signal 2112 to generate the periodic timing signal 1004. It is delayed for a while and directly outputs the digitally coded timing signal 2102. The linearized ROM 2110 stores linearized data in order to correctly linearize the amount of time delay provided by the PN code ROM 2106. Each 15-bit pseudorandom code 2108 provided to the linearized ROM 2110 represents the dither time delay used to time-modulate the digital data bits 2107 provided simultaneously to the linearized ROM 2110. In this embodiment, 2 to the 15th power (23,768) different amounts of time delay can be used to time-modulate the forward time shift of logical value 1 or the backward time shift of logical value 0. Cross-correlation with impulse radio receivers<u style="single">Prior to</u>Data can be restored by modulation of the time delay composed of the PN noise code. A preferred embodiment of an impulse radio receiver that illustrates this operation will be described below. FIG. 22 is a functional diagram showing the linearized ROM 2110. Locations 2202 and 2204 in FIG. 22 represent storage locations inside the linearized ROM 2110, which are addressed with high and low addresses, respectively. In this example, each storage location can store 8-bit data.That is, in this example, the data stored inside the linearized ROM 2110 is divided into two groups, the data at location 2202 and the data at location 2204. The data in the first group (location 2202) represents the linearized data used when the digital data source 2107 is, for example, a logical value of 1, and the linearized data stored in the second group (location 2204). Represents the linearized data used when digital data source 2107 has a Boolean value of 0. That is, the logical values of the digital data source 2107 that make up the most significant bit of the ROM address indicate whether the linearized data is output from block 2202 or from block 2204. The 15-bit PN code 2108 applied to the lower 15 address inputs of the linearized ROM 2110 determines which specific ROM location inside either the selected location 2202 or 2204 set is output from the linearized ROM 2110. Used to select. In a further embodiment of the invention, the PN code can be mathematically combined with linearized data and the resulting numerical information can be stored directly in a single ROM or the like. This further embodiment avoids the need for two ROMs. The address counter 2104 simply inputs the address directly into a single PN code / linearized ROM. (In diffusion spectrum theory, each element of the PN code is called a "chip", that is, a PN code with a modular N length contains a total of N chips.) The first ROM has the desired delay value for each code chip. Instead of outputting and linearizing each delay value, a single ROM can be used to store the linearized version of the desired delay on each code chip. Yet another embodiment of the impulse radio transmitter is illustrated in the block diagram of FIG. In FIG. 23, the combination of the PN code and the linearized EPROM2302 is used to generate the 8-bit coded information signal 2304, which represents the amount of time delay to be generated by the code time modulator 1008. PN co The mode can be switched ON / OFF using the code switch 2306. The code can be eliminated for a variety of reasons, for example, in an independent arithmetic mode in which the impulse radio receiver can acquire and lock the accelerated signal. The code switch 2306 can be controlled by a simple switch, independent control logic, a microprocessor, and the like. When the code switch is on, the time base 1002 is used to clock the address generator 2104, as described above in the context of FIG. 21, as illustrated in FIG. However, in FIG. 23, the time base is illustrated to be implemented by the VCO2308 and the programmable divider 2310. The functions performed by the VCO2308 and the programmable divider 2310 will be apparent to those skilled in the art. According to the embodiment illustrated in FIG. 23, a counter start page block 2312, a counter stop page block 2314, and a counter limit comparator block 2316 are included. The counter start page block 2312 supplies an address (preferably 15 bits) to the address generator 2104 to indicate the start address. The counter stop page block 2314 supplies an address (also preferably 15 bits) to the counter limit comparator block 2316 to represent the stop address. The counter limit comparator block 2316 contains logic that compares the address generated by the address generator 2104 with the stop page address supplied by the counter stop page block 2314. The counter's limit comparator block 2316 generates a load signal 2317 and forwards the load signal 2317 to the address generator 2104 when the comparisons of these addresses are equal. In response to the reception of load signal 2317, the address generator 2104 is reset and starts counting again at the 15-bit address specified in counter start page block 2312. To do. The counting process from the start page address to the stop page address is repeated continuously. By repeating these addresses, the PN code and linearized EPROM2306 can modulate digital data with a PN code modulo of length determined by the difference between the start page of the counter and the stop page address of the counter. .. As described above, the combination of the PN code and the linearized EPROM2302 is used to generate the 8-bit coded information signal 2304, which represents the amount of time delay to be generated by the code time modulator 1008. The code time modulator 1008 time-position-modulates the coded information signal 2304 using the periodic timing signal 1004. The code time modulator 1008 directly outputs the digitally coded timing signal 2102, as described above in the context of FIG. The embodiment illustrated in FIG. 23 also includes an FM subcarrier modulator 2318. The FM subcarrier modulator 2318 produces a sinusoidal signal 2320. The sinusoidal signal 2320 is added by the adder 2322 to the baseband audio signal 2342 supplied by the baseband audio source 2344. Note that the baseband audio source is an example of the information source 1018. The adder 2322 outputs the modulator signal 2324 used by the subcarrier time modulator 1016 in a manner similar to that described above in the context of FIG. When decoded by the impulse radio receiver, the restored sine wave signal 2320 can be used as a control signal by the impulse radio receiver. That is, the example of the impulse radio transmitter illustrated in FIG. 23 transmits a signal that transmits three separate pieces of information in a single impulse radio transmission. The signals that carry these three pieces of information include digital data 2107, sinusoidal signal 2320, and baseband audio signal 2342. Alternatively, block 2344 in FIG. 23 is replaced or blocked by subcarrier generator / modulator 1022 as described above in the context of FIG. 1018 and 2318 can be replaced with one of the subcarrier generators / modulators 1702, 1704, 1706 as described above in the context of FIG. According to yet another embodiment, the direct digitally coded timing signal 2102 can be directly input to the output stage 1028. In this embodiment, Manchester coding is the only form of subcarrier modulation performed. Other configurations will be apparent to those skilled in the art after perusing this disclosure. V. 2 Receiver A further embodiment of the impulse radio receiver is illustrated in FIG. This example of an impulse radio receiver is similar in many respects to the receiver described above in the context of FIG. The receiver illustrated in FIG. 24 includes a cross-correlator 1408, a subcarrier decoder 1424, a lowpass filter 1428, an adjustable time base 1414, a decoding timing modulator / decoding source 2402, a pseudo Manchester decoder 2404, and a microprocessor 2406. Including. According to this embodiment, the propagating signal (1404) is received by the impulse radio receiving antenna 1402, which passes the received signal 1406 to the RF amplifier 2408. The RF amplifier 2408 amplifies the received signal and passes it to the cross-correlator 1408. The cross-correlator 1408 can include a multiplier 2410, a triggered waveform generator 2412, an amplifier 2414, an integrator 2416, a sample and hold unit 2418, and a delay unit 2420. The multiplier 2410 is a double balanced mixer suitable for operating in linear mode. mixer). The multiplier 2410 linearly multiplies the template signal 2422 generated by the triggered waveform generator 2412 with the received signal. The product signal 2415 of the multiplier 2410 is buffered by the amplifier 2414 and then integrated over time by the integrator 2416. The integrator is basically a first-order lowpass filter suitable for responding on a time scale similar to the width of a monocycle (ie 157ps). The integrator 2416 outputs the signal 2417 to a sample and hold unit 2418 that holds the peak value of the signal 2417. Delay unit 2420 is the unit for the correct triggering of sample and hold unit 2418. The delay unit 2420 allows for the delay and integrator stabilization time incurred by the multiplier 2410 and amplifier 2414. In one embodiment, the delay unit 2420 delays a trigger of about 10 to 15 nanoseconds after the peak value generated by the integrator 2416. As a result, sampling occurs before the integral value deteriorates. According to this embodiment of the impulse radio receiver, the decoded signal 1420 is generated in a manner similar to the generation of the direct digitally coded timing signal 2102 described above in the context of FIG. The main difference between block 2402 in the impulse radio receiver and in the impulse radio transmitter is that the digital source is not used to access the PN code / linearized ROM. The decode timing modulator / decode source 2402 includes a binary-time delay generator 2424, a PN code and linearized ROM 2426, and an address counter and limit logic block 2428. The start address and stop address signals are fed from microprocessor 2406 to address counter and limit logic block 2428 via lines 2430 and 2432, respectively. The address is output from the address counter and limit logic block 2428 via bus 2434. Address counter and re The mitt logic block 2428 provides an address to access the PN code and the linearized ROM 2426 when triggered by the periodic timing signal 1416 provided by the adjustable time base 1414. The PN code (corresponding to the known PN code used in the impulse radio transmitter) is output from the PN code and linearized ROM 2426 via bus 2436 and fed to the binary-time delay generator 2424. The binary-time delay generator 2424 time-tunes the periodic timing signal 1416 (received from the adjustable time base 1414) to generate the decode signal 1420. In this embodiment, the adjustable time base 1414 includes a programmable divider 2438 and a voltage controlled oscillator (VCO) 2440, which are used to output the periodic timing signal 1416. The voltage control signal is supplied from the microprocessor 2406 to the VCO 2440 via line 2442 to regulate the VCO output, as will be apparent to experts in the relevant technology. In this embodiment, the subcarrier demodulator 1424 includes a bandpass filter 2444, a phase locked loop 2446, and a lowpass filter 2448. The function performed in Phase Locked Loop 2446 is equivalent to the function performed in a similar Phase Locked Loop (2004) of FIG. In this case, the bandpass filter 2444 outputs the filtered signal 2445 to the phase locked loop 2446. The phase locked loop 2446 outputs the common mode prediction signal 2447 to the microprocessor 2406 via yet another low-pass filter 2449. The in-phase prediction signal 2447 provides the microprocessor 2407 with a prediction of subcarrier amplitude, which allows the microprocessor 2406 to evaluate the quality of the signal lock. The demodulated output signal 2450 of the phase locked loop 2446 is filtered by a lowpass filter 2448, which outputs the demodulated information signal 1426. In Figure 24 The overall function and operation of the subcarrier demodulator 1424 is substantially the same as described above in the context of FIG. The control loop 1429 has the same function as described above in the context of FIG. Further subcarrier modulation is achieved according to another aspect of the invention that uses pseudo-Manchester coding of digital data. It is called "pseudo" because traditional Manchester coding performs digital decoding. However, according to the present invention, decoding of Manchester-encoded signals is performed in the analog domain. Pseudo-Manchester coding shifts digital information from baseband to a frequency equal to an adjustable time-based integral submultiple or time-based integer multiple. This provides a coherent shift of digital data for correct restoration in the impulse radio receiver. In this embodiment, the pseudo-Manchester decoder 2404 includes a bandpass filter 2450 and an analog Manchester decoder 2452. The bandpass filter 2450 receives the baseband signal 1422 from the cross-correlator 1408. The filtered baseband signal 2454 is fed to the analog Manchester decoder 2452. Decoding performed by the analog Manchester decoder 2452 is best described after the description of the actual coding performed by the transmitter. In addition, the various signals numbered in FIG. 24 are illustrated as time (t) vs. voltage plots from FIGS. 25A to 25H. Further FIGS. 25I to 25L are plots of frequency vs. amplitude (P or logP) corresponding to FIGS. 25E to 25H. VI. Pseudo-Manchester Modulation This section relates to pseudo-Manchester coding for the modulation of digital data using impulse radio communication. Using the direct digital modulation approach already described in the context of Figure 24, problems arise when the data source produces a long string with Boolean "1" or Boolean "0". Since the data is restored using a phase-locked loop, the low frequency energy of such a long string of "1" or "0" will appear in the lowpass filter 1428, causing a phase error in error loop 1429. It will be introduced. A method is needed to separate the modulated frequency component from the frequency component expected in error loop 1429. Therefore, the inventor has developed a further subcarrier embodiment.<u style="single">this</u>A further subcarrier embodiment is a modulation scheme in which a square wave whose frequency is at least twice the frequency of the data signal (2 × clock) and the data are exclusively exclusive (XOR) by a Manchester coding method. including. Manchester coding is used in impulse radio systems because the data uses a 2x clock.<u style="single">taller than</u>Because it is modulated to frequency<u style="single">、</u>It is a subcarrier technology. Impulse radio receivers eliminate this modulation in analog rather than digitally as is done in true Manchester decoding. The voltage from the sample and hold (2418) is modulated by a synchronous 2x clock and processed sequentially by a lowpass filter and a subsequent comparator (not shown). The simplest embodiment is a lowpass filter with a cutoff set at frequencies above about half the bit rate. However, it requires the most complex filtering to be used with other conversion methods. That is, the pseudo-Manchester modulation technique according to this aspect of the present invention converts a non-return-to-zero (NRZ) digital signal into a return-to-zero (RZ) signal. Avoids errors in the phase locked loop of the impulse radio receiver. Return-to-zero encoders are pseudo-Manchester direct digital encoders and frequency shift keying encoders. encoder), N-phase phase shift encoder (eg, 4-phase phase shift keying (QPSK)) Keying) or quadrature amplitude modulation encoders, or other frequency conversion means, as will be apparent to those skilled in the art). The pseudo-Manchester coding scheme according to the invention uses a standard implementation of Manchester coding in an impulse radio transmitter. The digital data stream is, for example, Manchester-encoded before being used to address the PN code and linearized EPROM2302 (see Figure 23). The circuits for achieving Manchester coding of digital data streams will be apparent to those skilled in the art. 26 and 27 show typical waveforms of pseudo-Manchester coding and decoding according to the present invention, respectively. In FIG. 26, a sample digital data stream with logical values 1 and 0 is typically shown in waveform 2602. For impulse radio transmitters, the data is exclusively ORed with a (2 x clock) square wave that is at least twice the frequency of the data signal shown in waveform 2604. Waveform 2604 must have transitions that are synchronized and matched with the edges of the data bits. The result of the exclusive OR of waveforms 2602 and 2604 is generally illustrated in waveform 2606. This process guarantees a 0 to 1 or 1 to 0 transition in the middle of each bit interval and eliminates problems associated with long 1 or 0 runs. In connection with the pseudo-Manchester coding embodiment, the impulse radio receiver performs pseudo-Manchester decoding to decode the digital data signal. A set of waveforms representing the functions performed to restore the data is illustrated in Figure 27. The received impulse radio signal is received by the impulse radio receiver.<u style="single">Mutual</u>Once correlated, it passes through the bandpass filter 2450. The output 2454 of the bandpass filter 2450 resembles the typical waveform generally illustrated in 2702. The filtered baseband signal 2454 is fed to the first input of an analog multiplier (not shown). The second input of the analog multiplier receives a synchronized 2x clock signal (2704). The analog multiplier reverses the process performed by the impulse radio transmitter. The product signal is lowpass filtered and compared with a predetermined comparison level as generally illustrated by the waveform at 2708 to give the comparison data signal 2710. The comparative data signal 2710 is held at the peak of the impulse / response point of the filter (using the sample and hold unit) via the rising edge of the data strobe signal (commonly represented by waveform 2712). , Generates restored data 2465, which is generally illustrated in waveform 2714. The analog restoration technique according to the present invention utilizes a coherent link, that is, synchronous restoration, to reduce noise as much as logically possible by filtering. VII. Lock Acquisition Method This part relates to a lock acquisition method in which the impulse radio receiver acquires and maintains the lock of the impulse radio signal. Like all communication receivers, impulse radio receivers must first acquire and maintain a "lock" of the signal before recovering the data. FIG. 28 is a high level block diagram of the actions performed to acquire a lock on an impulse radio receiver. When the transmitter and receiver are turned on, as shown in steps 2802 and 2804, the microprocessor 2406 biases the VCO2440 (shown in step 2806), as shown in step 2808. Drift the lock loop 1429 at a programmed rate faster than the transmission interval of the distant transmitter. Some millions are typical offsets. The microprocessor 2406 then digitizes the voltage from the cross-correlator 1408 (received via the filter 1428) and searches for the non-zero average voltage, as illustrated in step 2808. The microprocessor reduces the rate difference (received VCO vs. transmitter VCO) and begins scanning for a time near the recognition time when the energy was detected, as illustrated in step 2810. Other than this, the digitization performed by the microprocessor 2406 can be done using independent A / D converter hardware. Similarly, filtering can be done by microprocessors, discrete components, or active filters, as will be apparent to those skilled in the art. When the time corresponding to the maximum correlation energy is detected, the microprocessor switches to a tracking algorithm in which the average voltage of the correlator is maintained at zero, as illustrated in step 2812. This tracking is similar to the four-phase locking algorithm used in traditional CW phase locked loop configurations, as will be apparent to those skilled in the art. That is, when the tracking algorithm is activated, the pseudo-Manchester decoder 2404 and subcarrier demodulator 142 Subcarrier demodulation of data according to 4 can be started. VIII. Real World Performance This section describes the performance of an impulse radio communication system in the real world with reference to the data collected by the inventor by testing the prototype. One impulse radio prototype created by the inventor has an average radiated power of 450 μW (microwatts). The center frequency is 675 MHz and is smoothed by a pseudo-random code at 256 positions. FIG. 29 shows the signal measured at 3 meters (see plot 2902) as well as the peripheral signal (see plot 2904). The measurements in this drawing were not tuned to guarantee antenna performance, and a 1.3GHz / 2mpps prototype was used with an average output power of 33μW. Just below 900MHz, the power spike 2906 is from two cellular phone base stations, one at a distance of about 400 meters and the other at a distance of about 1.6 kilometers. Spike 2908 between 360MHz and 720MHz is the dominant UHF television station. The 720MHz spike is a 2.2-megawatt EIRP channel 54 television station in Huntsville, Alabama, about 7 miles away. (The "bumpy" of the impulse spectrum measurements reflects the effect of multipath in the frequency domain. Moving the receiving antenna moves the NUL and peak locations. This does not affect the performance of the impulse system. ). Impulse radio performance was measured with a 1.3GHz / 2mpps prototype for two paths: 1) -9.6dBi transmit antenna embedded in a highly conductive medium with a total loss of 36dB in a 6cm pass, the inventor installed the impulse radio. It was used to transmit a 125 kbps pseudo-random bit stream to a 10 dBi receiving antenna through an additional 4 meters of air. The bit error rate was better than 0.5 × 10-5. 2) Set the bit rate to 7. It dropped to 8kbps and increased the range to 10 meters. The bit error rate was better than 10-6. Standard propagation modeling assumptions can be used to define the performance of a 1.3GHz / 2mpps simplex link in free space. Figure 30 shows 100 μW average power (-10 dBm), 10 dBi receiving antenna (about 90 ° beam), 2 dBi transmitting antenna (omnidirectional dipole pattern), S / N ratio 19.5 dB (BER about 10 to the -6th power), Assuming a margin of 6 dB, we represent curve 3002 showing the trade-off defined between the free space range and the bit rate. With reference to FIG. 31, this figure shows that it is easy to resolve a multipath impulse signal in the time domain. The measurements illustrated in Plot 3102 were made in a laboratory in a single story office complex. The lab included several feet of steel shelves, testing equipment, and metal file cabinets. On the other hand, the adjacent office space was occupied by a metal manufacturing company. The other was occupied by a personal computer sales office, along which was the company's warehouse (using steel shelves). The first arriving pulse (between 3 and 6 nanoseconds) has lower amplitude because it is transmitted through more walls than some later arriving pulses. IX. Conclusion Although the various embodiments of the present invention have been described above, it should be understood that these are presented as examples and are not limiting. That is, the gist and scope of the present invention should not be limited by any of the examples described above, but should be defined only according to the claims and their equivalents described below. The performance of 3GHz / 2mpps simplex link can be defined. Figure 30 shows 100 μW average power (-10 dBm), 10 dBi receiving antenna (about 90 ° beam), 2 dBi transmitting antenna (omnidirectional dipole pattern), S / N ratio 19.5 dB (BER about 10 to the -6th power), Assuming a margin of 6 dB, we represent curve 3002 showing the trade-off defined between the free space range and the bit rate. With reference to FIG. 31, this figure shows that it is easy to resolve a multipath impulse signal in the time domain. The measurements illustrated in Plot 3102 were made in a laboratory in a single story office complex. The lab included several feet of steel shelves, testing equipment, and metal file cabinets. On the other hand, the adjacent office space was occupied by a metal manufacturing company. The other was occupied by a personal computer sales office, along which was the company's warehouse (using steel shelves). The first arriving pulse (between 3 and 6 nanoseconds) has lower amplitude because it is transmitted through more walls than some later arriving pulses. IX. Conclusion Although the various embodiments of the present invention have been described above, it should be understood that these are presented as examples and are not limiting. That is, the gist and scope of the present invention should not be limited by any of the examples described above, but should be defined only according to the claims and their equivalents described below. The performance of 3GHz / 2mpps simplex link can be defined. Figure 30 shows 100 μW average power (-10 dBm), 10 dBi receiving antenna (about 90 ° beam), 2 dBi transmitting antenna (omnidirectional dipole pattern), S / N ratio 19.5 dB (BER about 10 to the -6th power), Assuming a margin of 6 dB, we represent curve 3002 showing the trade-off defined between the free space range and the bit rate. With reference to FIG. 31, this figure shows that it is easy to resolve a multipath impulse signal in the time domain. The measurements illustrated in Plot 3102 were made in a laboratory in a single story office complex. The lab included several feet of steel shelves, testing equipment, and metal file cabinets. On the other hand, the adjacent office space was occupied by a metal manufacturing company. The other was occupied by a personal computer sales office, along which was the company's warehouse (using steel shelves). The first arriving pulse (between 3 and 6 nanoseconds) has lower amplitude because it is transmitted through more walls than some later arriving pulses. IX. Conclusion Although the various embodiments of the present invention have been described above, it should be understood that these are presented as examples and are not limiting. That is, the gist and scope of the present invention should not be limited by any of the examples described above, but should be defined only according to the claims and their equivalents described below. Assuming 5 dB (BER about 10-6) and a margin of 6 dB, we represent curve 3002 showing the trade-off defined between the free space range and the bit rate. With reference to FIG. 31, this figure shows that it is easy to resolve a multipath impulse signal in the time domain. The measurements illustrated in Plot 3102 were made in a laboratory in a single story office complex. The lab included several feet of steel shelves, testing equipment, and metal file cabinets. On the other hand, the adjacent office space was occupied by a metal manufacturing company. The other was occupied by a personal computer sales office, along which was the company's warehouse (using steel shelves). The first arriving pulse (between 3 and 6 nanoseconds) has lower amplitude because it is transmitted through more walls than some later arriving pulses. IX. Conclusion Although the various embodiments of the present invention have been described above, it should be understood that these are presented as examples and are not limiting. That is, the gist and scope of the present invention should not be limited by any of the examples described above, but should be defined only according to the claims and their equivalents described below. Assuming 5 dB (BER about 10-6) and a margin of 6 dB, we represent curve 3002 showing the trade-off defined between the free space range and the bit rate. With reference to FIG. 31, this figure shows that it is easy to resolve a multipath impulse signal in the time domain. The measurements illustrated in Plot 3102 were made in a laboratory in a single story office complex. The lab included several feet of steel shelves, testing equipment, and metal file cabinets. On the other hand, the adjacent office space was occupied by a metal manufacturing company. The other was occupied by a personal computer sales office, along which was the company's warehouse (using steel shelves). The first arriving pulse (between 3 and 6 nanoseconds) has lower amplitude because it is transmitted through more walls than some later arriving pulses. IX. Conclusion Although the various embodiments of the present invention have been described above, it should be understood that these are presented as examples and are not limiting. That is, the gist and scope of the present invention should not be limited by any of the examples described above, but should be defined only according to the claims and their equivalents described below.