Switching power circuit

Description: TECHNICAL FIELD [Detailed description of the invention]

【0001】

[Technical field to which the invention belongs]

The present invention relates to a switching power supply circuit provided as a power supply in various electronic devices.

【0002】

[Conventional technology]

As a switching power supply circuit, a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular wave-like switching operation waveform, there is a limit to suppressing switching noise. It is also known that there is a limit to the improvement of power conversion efficiency due to its operating characteristics. Therefore, the applicant has previously proposed various switching power supply circuits using various resonant converters. The resonance type converter can easily obtain high power conversion efficiency, and low noise is realized by making the switching operation waveform sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

【0003】

The circuit diagram of FIG. 11 shows an example of a switching power supply circuit as a prior art that can be configured based on the invention previously proposed by the applicant. As the basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

【0004】

In the power supply circuit shown in this figure, the bridge rectifier circuit Di and the smoothing capacitor Ci generate a rectified smoothing voltage Ei corresponding to one level of the AC input voltage VAC from the commercial AC power supply (AC input voltage VAC).

【0005】

As the voltage resonance type converter that inputs and interrupts the rectified smoothing voltage Ei (DC input voltage), a single-ended method using one stone is adopted. In addition, a self-excited configuration is adopted as the drive system. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is selected as the main switching element Q1 that forms the voltage resonance converter. A primary side parallel resonant capacitor Cr is connected in parallel between the collector and the emitter of the main switching element Q1. A clamp diode DD is connected between the base and the emitter. Here, the parallel resonant capacitor Cr forms a primary side parallel resonant circuit together with the leakage inductance L1 obtained in the primary winding N1 of the isolated converter transformer PIT, so that the operation as a voltage resonant converter can be obtained. It has become. Then, a self-excited oscillation drive circuit including a drive winding NB-resonant capacitor CB-base current limiting resistor RB is connected to the base of the main switching element Q1. The main switching element Q1 is switched and driven by supplying a base current based on an oscillation signal generated by this self-excited oscillation drive circuit. At the time of start-up, it is started by the start-up current flowing from the line of the rectified smoothing voltage Ei to the base via the start-up resistor Rs-base current limiting resistor RB.

【0006】

The orthogonal type control transformer PRT is configured by winding the control winding Nc so that the winding direction of the drive winding NB and the current detection winding ND is orthogonal to the winding direction, and will be described later. It is provided to control the switching frequency of the primary side voltage resonance type converter.

【0007】

The isolated converter transformer PIT is provided to transmit the switching output of the switching converter obtained on the primary side to the secondary side. Although detailed description here is omitted, the isolated converter transformer PIT has a gap formed with respect to the core so that a loosely coupled state can be obtained.

【0008】

The winding start end of the primary winding N1 of the isolated converter transformer PIT is connected to the collector of the main switching element Q1, and the winding end is connected to the line of the rectified smoothing voltage Ei via the current detection winding ND. Will be done. Further, the winding start end of the secondary winding N2 is connected to the secondary ground, and the winding end is connected to the positive electrode terminal of the smoothing capacitor CO via the rectifying diode DO.

【0009】

Further, on the primary side of the circuit shown in this figure, the active clamp circuit 22 previously proposed by the applicant is provided. As shown in the figure, the active clamp circuit 22 is formed by connecting the series connection circuit of the clamp capacitor CCL-auxiliary switching element Q2 in parallel with the primary winding N1 of the insulation converter transformer PIT. Here, a MOS-FET is adopted for the auxiliary switching element Q2.

【0010】

As a drive circuit system of the auxiliary switching element Q2, as shown in the figure, a self-excited oscillation drive circuit in which a capacitor Cg-resistor Rg-drive winding Ng is connected in series is connected to the gate of the auxiliary switching element Q2. .. Further, the resistor R1 is inserted between the gate and the source of the auxiliary switching element Q2. The drive winding Ng in this case is formed so as to wind up the winding end end side of the primary winding N1 in the insulation converter transformer PIT, and the number of turns is, for example, 1T (turn). Then, the auxiliary switching element Q2 performs the switching operation so as to conduct the conduction in a predetermined period at the on / off timing synchronized with the switching cycle of the main switching element Q1. As a result, the main switching element Q1 is turned off. During this period, the parallel resonance pulse waveforms generated at both ends of the parallel resonance capacitor Cr are clamped to suppress the peak level.

【0011】

The switching output of the main switching element Q1 forming the primary voltage resonance type converter is transmitted to the primary winding N1 of the insulation converter transformer PIT having the above structure, and is further excited with respect to the secondary winding N2. It will be transmitted.

【0012】

In this case, on the secondary side of the isolated converter transformer PIT, the leakage inductance of the secondary winding N2 is formed by connecting the secondary side parallel resonant capacitor C2 in parallel with the secondary winding N2 as shown in the figure. Form a secondary parallel resonant circuit with L2. Then, a half-wave rectifier circuit composed of a rectifier diode DO and a smoothing capacitor CO is connected to the secondary side parallel resonant circuit according to the connection form shown in the figure, so that the secondary side DC output voltage EO is output.

【0013】

In the power supply circuit having such a configuration, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is provided with a parallel resonance circuit for obtaining the voltage resonance operation. Become. In this specification, a switching converter having a configuration in which resonance circuits are provided on the primary side and the secondary side to operate in this way is also referred to as a composite resonance type switching converter.

【0014】

The control circuit 1 is configured to pass a DC current variable according to the level of the secondary side DC output voltage EO as a control current through the control winding Nc of the orthogonal control transformer PRT. By varying the control current level flowing through the control winding Nc in this way, in the orthogonal control transformer PRT, the inductance LB of the drive winding NB is controlled to be variable. As a result, the resonance frequency of the resonance circuit composed of the drive winding NB-resonance capacitor CB in the self-excited oscillation drive circuit changes, and the switching frequency of the main switching element Q1 is variably controlled. By changing the switching frequency of the main switching element Q1 in this way, the secondary side DC output voltage EO is controlled to be constant. That is, the power supply is stabilized.

【0015】

Here, for reference, the specifications of the main parts of the power supply circuit shown in FIG. 11 are described below. The insulation converter transformer PIT uses an ER40 type ferrite core, and the primary winding N1 = 50T and the secondary winding N2 = 50T. In addition, the drive resonance capacitor Cg = 0.33μF, resistance Rg = 10Ω, resistance R1 = 82Ω, primary side parallel resonance capacitor Cr = 2200pF, clamp capacitor CCL = 0.047μF, secondary side parallel resonance capacitor C2 = 7500pF were selected. There is. With such specifications, a wide range of practically sufficient regulation characteristics from AC input voltage VAC = 80V to 288V can be satisfied under the condition of load power Po = 200W to 0W.

【0016】

FIG. 12 is a waveform diagram showing the operation of the primary side in the power supply circuit shown in FIG. 11 above. Figures 12 (a) to (g) show the operation when the AC input voltage VAC = 100V and the load power Po = 200W, respectively, and FIGS. 12 (h) to (n) show the operation when the AC input voltage VAC = 100V, respectively. , The operation when there is no load with the load power Pomin = 0W is shown. First, when the load power is Po = 200W, the primary side parallel resonance voltage V1 obtained at both ends of the parallel connection circuit of the main switching element Q1 // primary side parallel resonance capacitor Cr is as shown in FIG. 12 (a). The parallel resonance pulse waveform as shown in the figure is obtained at 0 level in the period TON1 when the main switching element Q1 is on and in the period TOFF1 when it is off, so that the switching operation of the main switching element Q1 is a voltage resonance type. It is shown that The collector current flowing through the collector of the main switching element Q1 has the waveform shown in FIG. 12B according to the on / off timing of the main switching element Q1.

【0017】

Further, an alternating voltage excited by the drive winding Ng is applied to the gate of the auxiliary switching element Q2 forming the active clamp circuit 22 via the series connection of the resistor Rg-capacitor Cg, whereby the auxiliary switching element is applied. As the gate-source voltage VGS of Q2, as shown in Fig. 12 (g), a waveform that becomes a trapezoidal pulse is obtained in the period TOFF1. At this time, the drive winding current Ig based on the waveform obtained by differentiating the gate-source voltage VGS by the series connection circuit of the resistor Rg-capacitor Cg flows as shown in FIG. 12 (f). This differential waveform is obtained in the start interval and the end interval in the period TOFF1, which are the rising and falling periods of the parallel resonance pulse waveform. By such an operation of the drive circuit system, the parallel circuit of the auxiliary switching element Q2 // clamp diode DD2 forming the active clamp circuit 22 conducts the switching operation in the period TON2 and turns off in the period TOFF2. become.

【0018】

Then, in the period TON2, the current originally flowing through the parallel resonance capacitor Cr is made to flow as the clamp current IQ2 with respect to the clamp capacitor CCL as shown in FIG. 12 (d). As a result, as shown in FIG. 12 (e), the current IC1 flowing through the parallel resonant capacitor Cr during the period TOFF1 when the main switching element Q1 is turned off is only the start section and the end section during the period TOFF1. The amount of charging current in the parallel resonance capacitor Cr decreases, and the peak level of the parallel resonance voltage V1 shown in FIG. 12A is suppressed so as to be clamped to about 1/2.

【0019】

Then, under the condition of no load, for example, when the secondary side parallel resonance voltage rises, the waveforms shown in FIGS. 12 (a) to 12 (g) above are shown in FIGS. 12 (h) to 12 (n), respectively. It changes as shown in. Here, as can be seen by comparing FIGS. 12 (a) and 12 (h), the switching frequency fs is controlled to increase as the load power Po decreases, and the main switching element Q1 The off period TOFF1 is fixed, and the switching frequency fs (switching cycle) is changed by changing the on period TON. Then, constant voltage control is performed by such an operation. Then, even in the state where the switching frequency is variable in this way, the main switching element Q1 and the active clamp circuit 22 perform on / off operation at the same timing as in the case of the load power Po = 200W. As a result, the peak of the parallel resonance voltage pulse shown in FIG. 12 (h) is suppressed. In this way, in the power supply circuit provided with the active clamp circuit, it is possible to significantly suppress the peak level of the voltage resonance pulse generated by the switching operation, thereby making the switching element, the resonance capacitor, etc. a low withstand voltage product. Can be selected. In addition, the characteristic that the control range is expanded is obtained. Further, as the drive circuit for the auxiliary switching element Q2 of the active clamp circuit 22, the self-excited oscillation drive circuit is used as shown in FIG. 11, so that the circuit configuration is simplified as compared with the case of the separately excited type, for example. There is.

【0020】

[Problems to be Solved by the Invention]

By the way, in the case of the configuration including the active clamp circuit 22 as shown in FIG. 11, MOS-FET is used as the auxiliary switching element of the active clamp circuit. Then, the auxiliary switching element Q2 is switched and driven by the self-excited oscillation drive circuit (Ng-Rg-Cg). However, such a configuration has the following problems. Due to the structure of MOS-FET, there is a capacitance between the gate and the source. Therefore, in the circuit shown in FIG. 11, when the auxiliary switching element Q2, which is the MOS-FET, conducts, a charge charge is generated in the capacitance between the gate and the source. However, this charge charge decreases as the current Ig (Fig. 12 (f), Fig. 12 (m)) is pulled out at turn-off, but the self-oscillation drive circuit (Ng-) shown in Fig. 11 In the Rg-Cg) configuration, this charge charge cannot be rapidly and sufficiently extracted in the corresponding short time at turn-off. Therefore, the descending time tf of the auxiliary switching element Q2 at the time of turn-off becomes long. Therefore, ideally, the clamp current IQ2 shown in FIGS. 12 (d) and 12 (k) should suddenly reach the 0 level at the turn-off of the auxiliary switching element Q2, but in reality, the turn-off period remains the same. The descent time is tf, and the clamp current has flowed.

【0021】

In this way, the clamp current flows at the time of turn-off, so that the switching loss in the auxiliary switching element Q2 in particular increases by about 3 W. As a result, the DC-DC power conversion efficiency of the power supply circuit is reduced to 95.5%, which is 97% in the case of the circuit configuration without the active clamp circuit, for example. And under the condition of load power of 200W, the input power increases by about 3W. In addition, since heat is generated due to the increase in power loss, it is necessary to attach a heat sink to ensure the reliability of the auxiliary switching element Q2, which is a MOS-FET, which makes the circuit board smaller and lighter. It is a hindrance.

【0022】

[Means for solving problems]

Therefore, in consideration of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, the switching means formed by providing a switching element that switches the DC input voltage by being switched driven by the self-excited oscillation drive circuit, and the output of the switching means obtained in the primary winding are sent to the secondary side. It is provided with an isolated converter transformer for transmission, a primary winding of the isolated converter transformer, and a primary side parallel resonant circuit formed by a primary side parallel resonant capacitor and provided so as to make the operation of the switching means a voltage resonance type. .. Further, it is obtained in a secondary side parallel resonant circuit formed by connecting a secondary side parallel resonant capacitor in parallel to a secondary winding wound around an isolated converter transformer, and in this secondary side parallel resonant circuit. It is provided with a DC output voltage generating means configured to obtain a secondary side DC output voltage by inputting an alternating voltage and performing a rectifying operation. In addition, a series connection circuit consisting of a clamp capacitor and a bipolar transistor is provided, and switching is driven by a drive resonance circuit so that the voltage generated in the primary side parallel resonance circuit is clamped during the period when the switching means is turned off. Provide active clamping means to be provided. Then, the first drive winding forming the self-excited oscillation drive circuit, the second drive winding forming the drive resonance circuit, and the switching means for the first and second drive windings. A detection winding for exciting the switching output and a control transformer around which the control winding is wound are provided, and the level is variable according to the level of the DC output voltage obtained by the DC output voltage generating means. A constant voltage control means for variably controlling the switching frequency of the switching element by passing a control current through the control winding of the control transformer to perform constant voltage control on the secondary DC output voltage. To prepare.

【0023】

The switching power supply circuit is also configured as follows. That is, the switching means formed by providing a switching element that switches the DC input voltage by being switched driven by the self-excited oscillation drive circuit, and the output of the switching means obtained in the primary winding are sent to the secondary side. It is provided with an isolated converter transformer for transmission, a primary winding of the isolated converter transformer, and a primary side parallel resonant circuit formed by a primary side parallel resonant capacitor and provided so as to make the operation of the switching means a voltage resonance type. .. Further, it is obtained in a secondary side parallel resonant circuit formed by connecting a secondary side parallel resonant capacitor in parallel to a secondary winding wound around an isolated converter transformer, and in this secondary side parallel resonant circuit. It is provided with a DC output voltage generating means configured to obtain a secondary side DC output voltage by inputting an alternating voltage and performing a rectifying operation. Further, the switching is provided by providing a series connection circuit of a clamp capacitor and a MOS field effect transistor, and switching is driven by a drive resonance circuit formed by connecting a resonance capacitor in parallel with the drive winding. An active clamping means is provided so as to clamp the voltage generated in the primary side parallel resonant circuit during the period when the means are turned off. Then, the first drive winding forming the self-excited oscillation drive circuit, the second drive winding forming the drive resonance circuit, and the switching means for the first and second drive windings. The detection winding for exciting the switching output of the device, the control transformer around which the control winding is wound, and the control current at a level that is variable according to the level of the DC output voltage obtained by the DC output voltage generating means. Provided with a constant voltage control means for variably controlling the switching frequency of the switching element by flowing it through the control winding of the control transformer to perform constant voltage control on the secondary side DC output voltage. And said.

【0024】

The switching power supply circuit is also configured as follows. That is, it can be obtained in the primary winding, the oscillating means for outputting the oscillation frequency signal, the switching means formed with the switching element for switching the DC input voltage by switching driving based on the oscillating frequency signal, and the primary winding. It is formed by an isolated converter transformer that transmits the output of the switching means to the secondary side, a primary winding of the isolated converter transformer, and a parallel resonant capacitor on the primary side, so that the operation of the switching means is a voltage resonance type. It is provided with a primary side parallel resonant circuit provided. Further, it is obtained in a secondary side parallel resonant circuit formed by connecting a secondary side parallel resonant capacitor in parallel to a secondary winding wound around an isolated converter transformer, and in this secondary side parallel resonant circuit. It is provided with a DC output voltage generating means configured to obtain a secondary side DC output voltage by inputting an alternating voltage and performing a rectifying operation. Then, switching drive is provided by a drive resonance circuit formed by including a series connection circuit of a clamp capacitor and a bipolar transistor, a drive winding, and a resonance capacitor and a resistor connected in series with the drive winding. By doing so, an active clamping means provided to clamp the voltage generated in the primary side parallel resonance circuit during the period when the switching means is turned off is provided, and the switching output of the switching means is transmitted to the drive winding. By variably controlling the switching frequency of the switching element according to the level of the drive transformer and the secondary side DC output voltage, the constant voltage control of the secondary side DC output voltage is performed. It was decided to provide a constant voltage control means.

【0025】

The switching power supply circuit is also configured as follows. That is, it can be obtained in the primary winding, the oscillating means for outputting the oscillation frequency signal, the switching means formed with the switching element for switching the DC input voltage by switching driving based on the oscillating frequency signal, and the primary winding. It is formed by an isolated converter transformer that transmits the output of the switching means to the secondary side, a primary winding of the isolated converter transformer, and a parallel resonant capacitor on the primary side, and is provided so that the operation of the switching means is a voltage resonant type. It is equipped with a primary side parallel resonant circuit. Further, the secondary side parallel resonant circuit formed by connecting the secondary side parallel resonant capacitor in parallel to the secondary winding wound around the isolated converter transformer and the alternation obtained in the secondary side parallel resonant circuit. It is provided with a DC output voltage generating means configured to obtain a secondary side DC output voltage by inputting a voltage and performing a rectification operation, and a series connection circuit consisting of a clamp capacitor and a MOS type electric field effect transistor. By switching and driving by a drive resonance circuit formed with a resonance circuit including a drive winding and a resonance capacitor, the voltage generated in the primary side parallel resonance circuit is clamped during the period when the switching means is turned off. The active clamping means provided as described above and a drive transformer having a structure for transmitting the switching output of the switching means to the drive winding are provided. Further, the constant voltage control means for controlling the constant voltage of the secondary side DC output voltage by variably controlling the switching frequency of the switching element according to the level of the secondary side DC output voltage. It was decided to prepare.

【0026】

In addition, it was decided to configure the switching power supply circuit as follows. That is, the output of the switching means formed by including a switching element that switches the DC input voltage by being driven by a predetermined switching frequency by self-excited oscillation and the output of the switching means obtained in the primary winding are secondary. It is provided with an isolated converter transformer for transmission to the side, a primary winding of the isolated converter transformer, and a primary side parallel resonant circuit formed by a primary side parallel resonant capacitor and provided so that the operation of the switching means is a voltage resonant type. .. In addition, the secondary side parallel resonant circuit formed by connecting the secondary side parallel resonant capacitor in parallel to the secondary winding wound around the isolated converter transformer, and the secondary side parallel resonant circuit are obtained. It is provided with a DC output voltage generating means configured to obtain a secondary side DC output voltage by inputting an alternating voltage to be generated and performing a rectifying operation. Further, a rectifying diode that is provided with a series connection circuit of a clamp capacitor and a bipolar transistor and is switched and driven by a driving resonance circuit formed by at least a drive winding and a resonance capacitor to form a DC output voltage generating means. The active clamping means provided to clamp the voltage generated in the secondary side parallel resonant circuit during the period when the element is turned off, the drive winding of the drive resonant circuit, and the secondary winding with respect to this drive winding. A detection winding that excites the alternating voltage obtained in the above and a control transformer around which the control winding is wound are provided. Further, the conduction angle of the bipolar transistor is variably controlled by passing a control current having a level variable according to the level of the DC output voltage obtained by the DC output voltage generating means through the control winding of the control transformer. Therefore, it was decided to provide a constant voltage control means for performing constant voltage control on the secondary side DC output voltage.

【0027】

According to each of the above configurations, the primary side is provided with a primary side parallel resonance circuit for forming a voltage resonance type converter, and the secondary side is formed by a secondary side winding and a secondary side parallel resonance capacitor. A so-called composite resonance type switching converter configuration provided with a secondary side parallel resonance circuit can be obtained. The parallel resonance voltage level is suppressed by providing an active clamping means for clamping the parallel resonance voltage generated in the primary side parallel resonance circuit or the secondary side parallel resonance circuit with respect to the primary side or the secondary side. To be done. With such a configuration, the active clamping means of the present invention is switched and driven by a drive resonance circuit provided with a drive winding and a resonance circuit, and the drive resonance circuit is a primary side voltage resonance type. It is designed to be almost the same as the switching frequency of the converter. By adopting such a configuration, the current supplied to the control terminals (base, gate, etc.) of the transistor element forming the active clamp means can be sinusoidal, and as a result, the active clamp circuit is formed. It is possible to significantly shorten the descent time tf at the time of turn-off of the transistor element.

【0028】

BEST MODE FOR CARRYING OUT THE INVENTION

FIG. 1 shows a configuration example of a switching power supply circuit (corresponding to claims 1 and 2) as the first embodiment of the present invention. The power supply circuit shown in FIG. 1 is configured as a composite resonance type switching converter having a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side.

【0029】

In the power supply circuit shown in this figure, a full-wave rectifier circuit consisting of a bridge rectifier circuit Di and a smoothing capacitor Ci is provided as a rectification and smoothing circuit for inputting a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage. The rectified smoothing voltage Ei corresponding to the level equal to the AC input voltage VAC is generated.

【0030】

The voltage resonance type switching converter provided in this power supply circuit adopts a self-excited configuration equipped with one main switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is used for the main switching element Q1.

【0031】

A series resonance circuit for self-excited oscillation drive consisting of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB is connected between the base of the main switching element Q1 and the primary side ground. In addition, the base of the main switching element Q1 is also connected to the positive electrode side of the smoothing capacitor Ci (rectified smoothing voltage Ei) via the starting resistor RS, so that the base current at startup is obtained from the rectified smoothing line. ..

【0032】

In addition, the clamp diode DD inserted between the base of the main switching element Q1 and the negative electrode (primary side ground) of the smoothing capacitor Ci forms a path for the clamp current that flows when the main switching element Q1 is off. Further, the collector of the main switching element Q1 is connected to one end of the primary winding N1 of the isolation converter transformer PIT via the resonance current detection winding NR wound around the orthogonal type control transformer PRT described later, and the emitter is Be grounded.

【0033】

Further, a parallel resonant capacitor Cr is connected in parallel between the collector and the emitter of the main switching element Q1. This parallel resonance capacitor Cr forms a primary side parallel resonance circuit of a voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulation converter transformer PIT described later. Although detailed description is omitted here, when the main switching element Q1 is turned off, the voltage V1 across the parallel resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of this parallel resonance circuit, and is a voltage resonance type. You can get the operation of.

【0034】

The orthogonal control transformer PRT shown in this figure is a saturable reactor in which a resonance current detection winding NR, drive windings NB, NB2, and control winding NC are wound. The orthogonal control transformer PRT in this case drives the main switching element Q1 and the auxiliary switching element Q2 forming the active clamp circuit 20 described later, and is provided for constant voltage control. Although not shown, the structure of this orthogonal control transformer PRT forms a three-dimensional core by joining the ends of the magnetic legs of two double U-shaped cores having four magnetic legs. .. Then, the resonance current detection winding NR and the drive windings NB and NB2 are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is further resonated as described above. It is configured by winding in the direction orthogonal to the current detection winding NR and the drive windings NB and NB2. Here, the drive winding NB2 is formed so as to wind up the winding end end side of the resonance current detection winding NR.

【0035】

In this case, the resonance current detection winding NR of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the isolation converter transformer PIT to switch the main switching element Q1. The output is transmitted to the resonance current detection winding NR via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding NR is induced in the drive windings NB and NB2 via the transformer coupling, so that each of the drive windings NB and NB2 is driven. Alternate voltage as voltage is generated. Drive voltage obtained in the driving winding NB is the series resonance circuit (NB, CB) Karabe to form a self-excited oscillation driving circuit for the main switching element Q1 via the over scan current limiting resistor RB, the main as a drive current It is output to the base of the switching element Q1. As a result, the main switching element Q1 performs the switching operation at the switching frequency determined by the resonance frequency of the series resonant circuit. Further, the drive voltage obtained in the drive winding NB2 is passed through an LCR series resonant circuit (NB2-Cg-Rg) forming a self-excited oscillation drive circuit for driving the auxiliary switching element Q2, as described later. , It is output to the base of the main switching element Q1 as the drive current. As a result, the auxiliary switching element Q2 also performs the switching operation at the switching frequency determined by the resonance frequency of the LCR series resonant circuit.

【0036】

The isolated converter transformer PIT transmits the switching output of the main switching element Q1 to the secondary side. As shown in FIG. 9, the insulation converter transformer PIT is provided with an EE-type core in which, for example, E-type cores CR1 and CR2 made of ferrite material are combined so that their magnetic legs face each other, and the central magnetic leg of this EE-type core is provided. On the other hand, the split bobbin B is used to wind the primary winding N1 and the secondary winding N2 in a split state. Then, a gap G is formed for the central magnetic leg as shown in the figure. This makes it possible to obtain loose coupling with the required coupling coefficient. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs. Further, as the coupling coefficient k, for example, a loosely coupled state of k0.85 is obtained, which makes it difficult to obtain a saturated state.

【0037】

By the way, as the operation on the secondary side of the isolated converter transformer PIT, the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection relationship of the rectifier diode DO, and the alternation excited by the secondary winding N2. Due to the change in voltage polarity, the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M operation mode (polarization mode: forward operation) and -M operation. It may be in a mode (inductance mode: flyback operation). For example, the mutual inductance is + M when it is equivalent to the circuit shown in FIG. 10 (a), and the mutual inductance is -M when it is equivalent to the circuit shown in FIG. 10 (b).

【0038】

The winding start end side of the primary winding N1 of the isolated converter transformer PIT is connected to the collector of the main switching element Q1, and the winding end end is the positive electrode of the smoothing capacitor Ci via the series connection of the resonance current detection winding NR. It is connected to (rectified smoothing voltage Ei).

【0039】

Further, an active clamp circuit 20 is provided on the primary side of the power supply circuit shown in this figure. The active clamp circuit 20 includes an auxiliary switching element Q2, a clamp capacitor CCL, and a clamp diode DD2. In this case, a bipolar transistor is selected for the auxiliary switching element Q2. For the clamp diode DD2, a high-speed recovery type with a long reverse recovery special time is selected as a characteristic.

【0040】

Further, the collector of the auxiliary switching element Q2 is connected to the connection point between the line of the rectified smoothing voltage Ei and the winding end end of the primary winding N1 via the clamp capacitor CCL. Further, the emitter of the auxiliary switching element Q2 is connected to the connection point between the winding start end side of the primary winding N1 and the collector of the main switching element Q1. In addition, the anode of the clamp diode DD2 is connected to the emitter of the auxiliary switching element Q2, and the cathode is connected to the base of the auxiliary switching element Q2. I am trying to form it. As described above, the active clamp circuit 20 of the present embodiment is formed by connecting the clamp capacitor CCL in series to the switching circuit including the auxiliary switching element Q2 and the clamp diode DD2. Then, the circuit formed in this way is connected in parallel with the primary winding N1 of the insulation converter transformer PIT.

【0041】

Further, as described above, as the drive circuit system of the auxiliary switching element Q2, the LCR series resonance formed by the series connection circuit of the resistor Rg-capacitor Cg-drive winding NB2 with respect to the base of the auxiliary switching element Q2. The circuit is connected. The resonance frequency of the LCR series resonant circuit (Rg-Cg-NB2) is set to be equivalent to that of the series resonant circuit (RB-CB-NB2) forming the self-excited oscillation drive circuit of the main switching element Q1. Set. That is, it is set so as to be substantially equal to the switching frequency of the main switching element Q1. Therefore, for this embodiment, for example, a resistor Rg = 1Ω and a capacitor Cg = 0.68μF are selected, and the drive winding NB2 = 3T is set.

【0042】

Here, the drive winding NB2 is wound around the orthogonal control transformer PRT, and is excited by the alternating voltage obtained in the resonance current detection winding NR that detects the resonance current that is the switching output of the main switching element Q1. Alternate voltage is generated. Similarly, the drive winding NB also generates an alternation voltage excited by the alternation voltage obtained in the resonance current detection winding NR. Further, in this case, an alternating voltage having opposite polarities can be obtained between the drive winding NB and the drive winding NB2 due to the relationship of the winding directions.

【0043】

As will be described later, the active clamp circuit 20 operates so as to suppress the peak level of the parallel resonance voltage V1 generated across the parallel circuit of the main switching element Q1 // parallel resonance capacitor Cr. By suppressing the peak level of the parallel resonance voltage V1, it becomes possible to select low withstand voltage products for component elements such as the main switching element Q1 and the parallel resonance capacitor Cr. This makes it possible to select, for example, the main switching element. Since the switching characteristics of Q1 are improved, power loss can be reduced and reliability as a circuit can be improved. In addition, since the component size is reduced by making it a low withstand voltage product, it is possible to promote the reduction in size and weight of the circuit board.

【0044】

On the secondary side of the isolated converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2 are used. A parallel resonant circuit is formed. By this parallel resonance circuit, the alternating voltage induced in the secondary winding N2 becomes the resonance voltage. That is, a voltage resonance operation can be obtained on the secondary side.

【0045】

That is, in this power supply circuit, a parallel resonance circuit for making the switching operation a voltage resonance type is provided on the primary side, and a parallel resonance circuit for obtaining a voltage resonance operation is provided on the secondary side. It has a configuration as a "resonant switching converter".

【0046】

On the secondary side of the power supply circuit formed as described above, by connecting the rectifier diode DO and the smoothing capacitor CO as shown in the figure, half-wave rectification is performed by flyback operation (see Fig. 10 (b)). The secondary side DC output voltage EO is obtained by forming a half-wave rectifier circuit that operates. The DC output voltage EO is also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO is used as the detection voltage and the operating power source of the control circuit 1.

【0047】

In the control circuit 1, the drive wound around the orthogonal control transformer PRT is driven by varying the control current (DC current) level flowing through the control winding NC according to the change in the DC output voltage level EO on the secondary side. Variable control of the inductance LB of the winding NB. As a result, the resonance condition of the series resonance circuit in the self-oscillation drive circuit for the main switching element Q1 formed including the inductance LB of the drive winding NB changes. This is an operation of varying the switching frequency of the main switching element Q1, and this operation stabilizes the DC output voltage on the secondary side.

【0048】

Here, the auxiliary switching element Q2 in the active clamp circuit 20 is driven based on the voltage excited by the drive winding NB2. Further, the main switching element Q is driven based on the alternating voltage obtained in the drive winding NB. As described above, the alternating voltages obtained in the drive winding NB2 and the drive winding NB have opposite polarities, so that the timing of the switching operation of the auxiliary switching element Q2 is the main switching element. In synchronization with Q1, the main switching element Q1 operates so as to be turned on within the period when it is turned off. Then, in the present embodiment, in the orthogonal type control transformer PRT, the drive winding NB and the drive winding NB2 are wound as controlled windings. Therefore, as described above, the switching frequency of the main switching element Q1 is variably controlled according to the change in the inductance of the drive winding NB, and at the same time, the inductance of the drive winding NB2 also changes, so that auxiliary switching is performed. The switching frequency of the element Q2 is also variably controlled so as to be synchronized with the main switching element Q1.

【0049】

The waveform diagram of FIG. 2 shows the operation of the main part in the power supply circuit shown in FIG. 1 above. The operation shown in FIG. 2 is assumed to be obtained when the circuit shown in FIG. 1 is configured to correspond to the AC100V system. In FIGS. 2 (a) to 2 (g), the AC input voltage VAC = 100V, The operation of each part is shown under the condition that the load power Po = 200W, and Fig. 2 (h) to (n) show Fig. 2 under the condition that the AC input voltage VAC = 100V and the load power Po = 20W. The operation of the same part as (a) to (g) is shown.

【0050】

First, the operation when the load power Po = 200 W shown in FIGS. 2 (a) to 2 (g) will be described. In this figure, five operation modes from mode to mode are shown for the operation modes within one switching cycle. The main switching element Q1 is controlled to be turned on during the period TON1, and the operation as a mode is obtained in this period TON1. The auxiliary switching element Q2 is controlled to be in the off state during this period TON1.

【0051】

In the mode (period TON1), the switching output current IQ1 flows through the collector of the main switching element Q1 according to the waveform shown in Fig. 2 (b). This switching output current IQ1 is the primary winding of the isolated converter transformer PIT. It flows to the main switching element Q1 via the leakage inductance L1 obtained in N1. The switching output current IQ1 at this time has a waveform that reverses from the negative direction to the positive direction at the initial stage, as shown in the period TON1 in FIG. 2 (b). Here, during the period in which the switching output current IQ1 flows in the negative direction, the clamp diode DD becomes conductive when the discharge in the parallel resonance capacitor Cr ends at the end of the immediately preceding period td2, and the clamp diode DD the primary winding N1 By passing the switching output current IQ1 through the mode, the power is regenerated to the power supply side. Then, at the timing when the switching output current IQ1 (Fig. 2 (b)) reverses from the negative direction to the positive direction, the main switching element Q1 is ZVS (Zero Volt Switching) and ZCS (Zero Current). Turn on by Switching).

【0052】

Then, in the next period td1, the operation is performed as a mode. During this period, the main switching element Q1 turns off, so that the current flowing through the primary winding N1 flows through the parallel resonant capacitor Cr. At this time, the current Icr flowing through the parallel resonance capacitor Cr shows a waveform that appears in a pulsed manner due to the positive electrode property, as shown in FIG. 2 (e). This is an operation as a partial resonance mode. Further, at this time, since the parallel resonant capacitor Cr is connected in parallel with the main switching element Q1, the main switching element Q1 is turned off by ZVS.

【0053】

Subsequently, the auxiliary switching element Q2 is controlled to be in the on state and the main switching element Q1 is controlled to be in the off state, which is the auxiliary switching element shown in FIG. 2 (c). It corresponds to the period TON2 when the voltage V2 across Q2 becomes 0 level. This period TON2 is the operation period of the active clamp circuit 20, and the operation as a mode is first performed and then the operation as a mode is performed.

【0054】

In the operation of the previous mode, the parallel resonance capacitor Cr is charged by the current flowing from the primary winding N1, but as a mode operation, the voltage level obtained in the primary winding N1 is initially set. The potential is equal to or higher than the voltage level across the clamp capacitor CCL (at the start of period TON2). As a result, the conduction condition of the clamp diode DD2 connected in parallel to the auxiliary switching element Q2 is satisfied and the current flows in the path of the clamp diode DD2 the base of the auxiliary switching element Q2 the collector the clamp capacitor CCL. Therefore, as the clamp current IQ2, a serrated waveform approaching 0 level from the negative direction with the passage of time can be obtained after the start of the period TON2 in FIG. 2 (d). Here, for example, if the capacitance of the clamp capacitor CCL is selected to be 50 times or more the capacitance of the parallel resonance capacitor Cr, most of the current will be applied to the clamp capacitor CCL as the clamp current IQ2 depending on the operation in this mode. It is made to flow with respect to it, and hardly flows with respect to the parallel resonance capacitor Cr. As a result, the slope of the parallel resonance voltage V1 (Fig. 2 (a)) applied to the main switching element Q1 during this period TON2 is made gentle, and as a result, for example, when the active clamp circuit 20 is not provided. It is suppressed to about 1/2 and the conduction angle is widened. That is, a clamping operation with respect to the parallel resonance voltage V1 can be obtained.

【0055】

Then, when the above mode ends in the period TON2, the operation as the mode continues. At the start of this mode, the timing is such that the clamp current IQ2 shown in FIG. 2 (d) reverses from the negative direction to the positive direction. At this timing, the auxiliary switching element Q2 is turned on by ZVS and ZCS at the timing when the clamp current IQ2 reverses from the negative direction to the positive direction. In the state where the auxiliary switching element Q2 is turned on in this way, due to the resonance action of the primary side parallel resonant circuit obtained at this time, the collector emitter of the auxiliary switching element Q2 is passed through the primary winding N1 the clamp capacitor CCL. The clamping current IQ2 flows through the coil, and as shown in Fig. 2 (d), a waveform that increases in the positive direction is obtained.

【0056】

The operation of the above mode is made to end at the timing when the voltage V2 across the auxiliary switching element Q2, which has been set to 0 level in the period TOFF1 until now, starts to rise because the auxiliary switching element Q2 is turned on. Then, it shifts to the operation as a mode in the period td2. In the mode, the parallel resonant capacitor Cr can operate to pass a discharge current through the primary winding N1. That is, a partial resonance operation can be obtained. Then, in response to this partial resonance operation, the current Icr flowing through the parallel resonance capacitor Cr can obtain a waveform that appears in a pulsed manner due to the negative electrode property, as shown in FIG. 2 (e). Further, at this time, the parallel resonance voltage V1 applied to the main switching element Q1 has a large slope due to the small capacitance of the parallel resonance capacitor Cr as described above, and is shown in FIG. 2 (a). In this way, it descends rapidly toward the 0 level. Then, the auxiliary switching element Q2 starts turn-off at the timing when the above mode ends and the mode starts. At this time, the parallel resonance voltage V1 falls down with a certain inclination as described above. Then, it becomes a turn-off operation by ZVS. Further, the voltage generated by the turn-off of the auxiliary switching element Q2 is prevented from rising sharply by discharging the parallel resonant capacitor Cr as described above. This operation is performed as a waveform that transitions from 0 level to peak level with a certain slope with a period td2 (in mode), as shown by, for example, the voltage V2 across the auxiliary switching element Q2 in FIG. 2 (c). It is shown. As the voltage V2 across the auxiliary switching element Q2, for example, the peak level is almost maintained during the period TOFF2 when the auxiliary switching element Q2 is turned off, and the period td1 (in mode) which is the start period of this period TOFF2. From 0 level to peak level as described above, with the period td2 (in mode), which is the end period. It becomes a waveform that transitions with. After that, the mode ~ operation is repeated every one switching cycle.

【0057】

Here, the base-emitter voltage VBE of the auxiliary switching element Q2 rises in the period td1 and falls in the period td2, resulting in a pulsed waveform that peaks positively in the period TON2. The period td1 and the period td2 are set to a threshold period in which both the main switching element Q1 and the auxiliary switching element Q2 are turned off, and this threshold period is maintained by the flow of the base inflow current Ig.

【0058】

The base inflow current Ig that flows through the base of the auxiliary switching element Q2 is formed by the resistor Rg-capacitor Cg-drive winding NB2, and has a resonance frequency that is almost the same as the switching frequency of the main switching element Q1. By inserting the LCR series resonant circuit, the base inflow current Ig is at a positive level during the mode ~, as shown in Fig. 2 (f), due to the resonant action of this series resonant circuit. Therefore, a sine wave with a negative level can be obtained during the period corresponding to the mode. That is, it is a sine wave having a frequency equivalent to the switching frequency of the primary side voltage resonance type converter. On the other hand, in the case of the power supply circuit shown in FIG. 11, the current Ig (FIG. 12 (f)) flowing into the control terminal (gate) of the auxiliary switching element Q2 is the period corresponding to the period td1 and the period td2. Was obtained as a differential waveform pulse.

【0059】

In the present embodiment, as shown in FIG. 2 (f) above, a sinusoidal base inflow current Ig is passed through the base which is the control terminal of the auxiliary switching element Q2. In addition, in the auxiliary switching element Q2, it is possible to significantly shorten the descent time tf of the clamp current (collector current) IQ2 at the time of turn-off. That is, as shown in FIG. 2 (d), the switching output current IQ1, which is the collector current of the auxiliary switching element Q2, suddenly becomes 0 level at the turn-off time of the auxiliary switching element Q2 (at the end of the period TON2). It is something that is done. On the other hand, in the power supply circuit shown in FIG. 11, as shown as the gate inflow current Ig in FIG. 12 (f), the partial resonance period immediately after the turn-off appeared as the descending time tf.

【0060】

Further, the operation waveforms of the respective parts shown in FIGS. 2 (a) to 2 (g) above are shown in FIGS. 2 (h) to (n) under the condition that the load power is Po = 0 W and no load is applied. ) Changes.

【0061】

Here, for example, as can be seen by comparing the primary side parallel resonance voltage V1 in FIGS. 2 (a) and 2 (h), the waveform shown in FIG. 2 (h) is the period during which the main switching element Q1 is turned on. TON1 is significantly shorter, which makes the switching frequency higher than at the maximum load power shown in Fig. 2 (a). That is, the switching frequency of the main switching element Q1 is variably controlled as the constant voltage control operation by the orthogonal type control transformer PRT described above. Also, as can be seen from the voltage V2 in Fig. 2 (j), the gate inflow current Ig in Fig. 2 (m), and the base-emitter voltage VBE in Fig. 2 (n), which show the operation of the auxiliary switching element Q2 side. The switching frequency of the auxiliary switching element Q2 is also controlled. Therefore, in the present embodiment, the auxiliary switching element Q2 is variably controlled in the switching frequency so as to be synchronized with the control of the switching operation of the main switching element Q1.

【0062】

Then, even under the no-load condition shown in FIGS. 2 (h) to 2 (n), the mode ~ operation is performed at the timing shown in the figure, so that the peak level of the primary side parallel resonance voltage V1 is suppressed, and the peak level is also suppressed. The peak level of the voltage V2 across the auxiliary switching element Q2 is also suppressed to, for example, about 1/2.

【0063】

Then, as understood from the above description, in the present embodiment, the auxiliary switching element Q2 is turned off as a result by providing the resonance circuit as the self-excited oscillation drive circuit system of the active clamp circuit 20. The descent time tf at the time of the descent time is significantly shortened, and thereby the switching loss due to this descent time tf is also reduced. In reality, the DC-DC power conversion efficiency was improved from 95.5% in the circuit shown in FIG. 11 to 96.8% in the circuit shown in FIG. 1 as the present embodiment. The result is obtained. Along with this, the input power can be reduced by about 3W, and the invalid input power at no load (load power Po = 0W) can be reduced from 4.7W to 3.2W. .. Since the switching loss in the auxiliary switching element Q2 is reduced in this way and the heat generation is suppressed, for example, the heat sink provided for the auxiliary switching element Q2 can be eliminated.

【0064】

Further, as in the case where the main switching element Q1 and the auxiliary switching element Q2 are both bipolar transistors as in the power supply circuit shown in FIG. 1, when the characteristics are the same, as shown in FIG. Therefore, the main switching element Q1 and the auxiliary switching element Q2 can be packaged to form a composite transistor 30 as one component, which can be used. In the composite transistor 30 shown in this figure, the main switching element Q1 and the auxiliary switching element Q2 are connected in series with the [clamp diode DD-Zener diode ZD1] and the [clamp diode DD2-Zener diode, respectively]. The series connection circuit of ZD2] is connected by the form shown in the figure. The Zener diodes ZD1 and ZD2 are provided for protection when a reverse current is about to flow through the clamp diode.

【0065】

FIG. 4 shows a configuration example of a switching power supply circuit (corresponding to claims 4 and 5) as a second embodiment of the present invention. In this figure, the same parts as those in FIG. 1 are designated by the same reference numerals, and the description thereof will be omitted. Further, as the power supply circuit of the second embodiment, the insulation converter transformer PIT shown in FIG. 9 is provided, and the configuration of the composite resonance type switching converter is adopted. Note that this point is the same for each embodiment described below.

【0066】

In the power supply circuit shown in FIG. 4, first, a smoothing capacitor connected in series by connecting [rectifier diodes Di1, Di2, smoothing capacitors Ci1, Ci2] to the commercial AC power supply AC by the connection form shown in the figure. A rectified smoothing voltage Ei corresponding to twice the AC input voltage VAC of Ci1-Ci2 is generated and supplied to the primary voltage resonance type converter.

【0067】

Further, in the case of the present embodiment, the primary side voltage resonance type converter adopts a separately excited single-ended type configuration. In this case, an IGBT (insulated gate bipolar transistor) is used as the main switching element Q1. For this IGBT, for example, an 800V medium withstand voltage product is selected. The collector of the main switching element Q1 as an IGBT is connected to the end of the primary winding N1 and the emitter is connected to the primary ground. Further, the parallel resonant capacitor Cr is connected in parallel between the collector and the emitter of the main switching element Q1. The clamp diode DD is also connected in parallel between the collector and the emitter of the main switching element Q1.

【0068】

The switching drive unit 10 is provided to drive the main switching element Q1 by a separately excited type and to control the switching frequency, and can be configured as, for example, a single IC. The switching drive unit 10 includes an oscillation circuit 11 and a drive circuit 12. At the time of start-up, the switching drive unit 10 is adapted to obtain start-up power from the line of the rectified smoothing voltage Ei via the start-up resistor Rs.

【0069】

The oscillation circuit 11 generates an oscillation signal and outputs it to the drive circuit 12. In the drive circuit 12, the input oscillation signal is converted into a drive voltage that can drive the main switching element Q1 which is an IGBT, and is output to the gate of the main switching element Q1. As a result, the main switching element Q1 is switched and driven. Further, the oscillation circuit 11 is configured to change the frequency of the oscillation signal according to the error detection output of the secondary side DC output voltage EO output from the control circuit 1. In this way, the main switching element Q1 is driven based on the oscillation signal whose frequency is variable, so that the switching frequency of the main switching element Q1 is variable, thereby stabilizing the secondary side DC output voltage EO. Will be planned.

【0070】

Further, as the active clamp circuit 20A in this case, a drive transformer DT is provided as a drive circuit system. A resonance current detection winding NR is wound around the primary side of the drive transformer DT, and a drive winding NB2 is wound around the secondary side. As a result, the alternating voltage obtained in the resonance current detection winding NR is excited for the drive winding NB2, and as a result, as in the case of FIG. 1, the LCR resonance circuit (Rg) -Cg-NB2) can drive the auxiliary switching element Q2. Further, since the switching output of the main switching element Q1 whose switching frequency is controlled according to the load fluctuation or the like is obtained in the resonance current detection winding NR, it is driven based on the drive winding NB2 excited by this. The auxiliary switching element Q2 also follows the switching operation so as to have a switching frequency equivalent to that of the main switching element Q1. Even with such a configuration, the power supply circuit shown in FIG. 1 is provided with an LCR resonance circuit (Rg-Cg-NB2) as a self-excited oscillation drive circuit in the active clamp circuit 20A. A similar effect can be obtained.

【0071】

FIG. 5 shows a configuration example of a switching power supply circuit (corresponding to claims 7 and 8) as a third embodiment. In this figure, the same parts as those in FIGS. 1 and 4 are designated by the same reference numerals and the description thereof will be omitted. The primary side shown in this figure is provided with a voltage resonance type converter in the same manner as in each of the above-described embodiments, but the orthogonal type control transformer PRT is omitted. Therefore, as shown in the figure, the drive winding NB forming the self-excited oscillation drive circuit of the main switching element Q1 is wound around the primary side of the insulation converter transformer PIT, so that the alternation obtained in the primary winding N1 can be obtained. It is designed to be excited by a voltage. In such a configuration, since the frequency control of the main switching element Q1 is not performed, the main switching element Q1 is based on the resonance circuit (NB-CB) formed in the self-excited oscillation drive circuit. The switching operation is performed by a fixed switching frequency determined by the resonance frequency.

【0072】

Then, in the third embodiment, the active clamp circuit is provided not on the primary side but on the secondary side. In the active clamp circuit 20B provided on the secondary side, a bipolar transistor is selected as the auxiliary switching element Q2, as in the active clamp circuit 20 shown in FIGS. 1 and 2. As a drive circuit system by self-excited oscillation, as shown in the figure, a configuration including an orthogonal control transformer PRT and an LCR resonance circuit (Rg-Cg-NB2) is adopted.

【0073】

In this case, the collector of the auxiliary switching element Q2 is connected via the clamp capacitor CCL to the connection point between the end of the secondary winding N2 and the anode of the rectifier diode DO2, and the emitter is secondary. Connected to the side ground.

【0074】

Further, in the orthogonal type control transformer PRT in this case, as shown in the figure, the control winding Nc is wound so that the winding direction of the current detection winding NR and the drive winding NB2 is orthogonal to the winding direction. Be disguised. In this case, one end of the drive winding NB2 is connected to the secondary ground, and the other end is connected to the base of the auxiliary switching element Q2 via the capacitor Cg-resistor Rg. Further, the resonance current detection winding NR is inserted in series between the winding start end of the secondary winding N2 and the secondary ground. The detection output of the control circuit 1, that is, the control current, is supplied to the control winding Nc of the orthogonal control transformer PRT provided on the secondary side. In such a configuration, for example, the switching frequency is fixed, and the on period is variably controlled with the off period of the auxiliary switching element Q2 being constant according to the level change of the secondary side DC output voltage EO due to load fluctuation or the like. You can get the action of doing. That is, with respect to the switching operation of the auxiliary switching element Q2, an operation of variably controlling the conduction angle can be obtained. Here, for example, assuming that the level of the secondary side DC output voltage EO rises in a light load state, the conduction angle control is performed so that the ON period of the auxiliary switching element Q2 is extended.

【0075】

Then, as a result of performing PWM control as described above, as a voltage induced in the secondary winding N2 of the isolated converter transformer PIT, the pulse width of the negative waveform is expanded, and the pulse width of the positive waveform is expanded. Becomes shorter. In the secondary side rectifying diode DO, since this secondary side parallel resonance voltage is input and rectification is performed by forward operation, the period during which the secondary side rectifying diode DO is conducted and turned on is shortened, and one of them is turned off. The period will be extended. In this way, as a result, the conduction angle of the rectifier diode DO1 is controlled, so that the secondary side DC output voltage can be stabilized.

【0076】

In the configuration in which the active clamp circuit is provided on the secondary side in this way, the active clamp is applied to the peak level of the resonance pulse of the secondary side parallel resonant circuit (N2 // C2) generated during the period when the rectifier diode DO1 is off. Compared with the configuration when no circuit is provided, it can be reduced to about 1/2. Further, in the present embodiment, the self-excited oscillation drive circuit in the active clamp circuit 20A provided on the secondary side also has the configuration of the LCR resonance circuit (Rg-Cg-Lg). Similar to the power supply circuit of the above, the switching loss due to the auxiliary switching element Q2 is reduced, and the DC-DC power conversion efficiency of the power supply circuit is improved to the same level as in the case where the active clamp circuit is not provided. ..

【0077】

FIG. 6 shows a configuration example of a switching power supply circuit (corresponding to claim 2) as a fourth embodiment of the present invention. In this figure, the same parts as those in FIGS. 1, 4, and 5 are designated by the same reference numerals and the description thereof will be omitted. The power supply circuit shown in this figure is configured as a composite resonance type converter having a voltage resonance type converter and an active clamp circuit on the primary side and a parallel resonance circuit and a half-wave rectifier circuit on the secondary side. Overall, the configuration is similar to that of the power supply circuit shown in Fig. 1. However, the configuration of the active clamp circuit provided on the primary side is different.

【0078】

In the active clamp circuit 21 shown in FIG. 6, MOS-FET is selected as the auxiliary switching element Q2. When the auxiliary switching element Q2 is a MOS-FET in this way, the drain is connected to the clamp capacitor CCL, and the source is connected to the winding start end side of the primary winding N1. Further, the clamp diode DD2 is connected in parallel between the drain and the source of the auxiliary switching element Q2. In this case, the clamp diode DD2 uses a so-called body diode built in the auxiliary switching element Q2 as a MOS-FET.

【0079】

Further, in the first to third embodiments (power supply circuits of FIGS. 1, 4 and 5), the LCR series resonance circuit is used as the resonance circuit for driving the auxiliary switching element Q2. Therefore, in the fourth embodiment, a parallel resonant circuit is used. That is, a parallel resonance circuit is formed by connecting a resonance capacitor Cg in parallel with the drive winding NB2. Then, the output point of this parallel resonant circuit and the gate of the auxiliary switching element Q2 are connected via the resistor Rg. In this case, the resistor R1 is connected in parallel between the gate and the source of the auxiliary switching element Q2. Further, also in this case, the resonance frequency of the parallel resonance circuit (NB2 // Cg) is set to be equal to the switching frequency of the main switching element Q1, and for this purpose, a self-excited oscillation drive circuit is formed. A self-excited oscillation drive circuit is formed by selecting NB2 = 3T, Cg = 1μF, Rg = 22Ω, and R1 = 82Ω for each element to be powered.

【0080】

Then, as the operation of the power supply circuit with such a configuration, the same waveform as each part shown in FIG. 2 can be obtained. Here, in the case of the present embodiment, the auxiliary switching element Q2 may be a MOS-FET, but even in this case, the gate-source voltage of the auxiliary switching element Q2 is shown in FIGS. 2 (g) and 2 (). Similar to n), it has a rectangular wavy shape. The same waveform as in FIGS. 2 (d) and 2 (k) can be obtained for the clamp current IQ2, but in this case, the negative electrode positive clamp current IQ2 that flows corresponding to the mode period is the primary winding. The current flows in the path of wire N1 clamp diode DD2 clamp capacitor CCL. Further, also in the case of this embodiment, the gate inflow current Ig has a sinusoidal shape corresponding to the switching frequency as shown in FIGS. 2 (f) and 2 (m). The descent time tf of the clamp current (drain current) IQ2 (Fig. 2 (d) (k)) generated at turn-off is significantly shortened. Therefore, the switching loss is also reduced, and as a result, the DC-DC conversion efficiency of the entire circuit is improved. In the case of the circuit shown in FIG. 6, the DC-DC power conversion efficiency of 96.5% was obtained according to the actual experimental results, which is 1.0% higher than that of the circuit shown in FIG. In addition, the input power was reduced by about 2.3W, and the invalid input power at no load (load power Po = 0W) was reduced from 4.7W to 3.5W. Further, also in the case of this embodiment, since the power loss is reduced as described above, the heat sink provided for the auxiliary switching element Q2 can be eliminated.

【0081】

FIG. 7 shows a configuration example of a switching power supply circuit (corresponding to claim 6) as the fifth embodiment of the present invention. In this figure, the same parts as those in FIGS. 1, 4, 5 and 6 are designated by the same reference numerals and the description thereof will be omitted. The basic configuration of the power supply circuit shown in this figure is substantially the same as that of the circuit of the second embodiment shown in FIG. That is, the primary side voltage resonance type converter is driven by the separately excited type by providing the switching drive unit 10. However, in this case, a MOS-FET is provided as the main switching element Q1 instead of the IGBT, and therefore a body diode is used for the clamp diode DD2 with respect to the drain-source of the main switching element Q1. Connected in parallel. Further, in this figure, the active clamp circuit 21A provided on the primary side has a configuration provided with a drive transformer DT as in the active clamp circuit shown in FIG. 4, and is auxiliary. MOS-FET is adopted as the switching element Q2. Further, the self-excited oscillation drive circuit system for the auxiliary switching element Q2 is the same as in FIG. That is, it is configured by connecting a parallel resonant circuit (NB2 // Cg) to the resistor Rg.

【0082】

FIG. 8 shows a configuration example of a switching power supply circuit (corresponding to claims 7 and 9) as the sixth embodiment. In this figure, the same parts as those in FIGS. 1, 4, 5, 6, and 7 are designated by the same reference numerals and the description thereof will be omitted. The overall configuration of the power supply circuit shown in this figure is substantially the same as that of the power supply circuit as the third embodiment shown in FIG. Further, as the active clamp circuit 21B provided on the secondary side, a self-excited oscillation drive circuit provided with an orthogonal control transformer PRT on the secondary side and a resonance circuit according to the active clamp circuit 21 shown in FIG. The configuration is adopted. However, in this active clamp circuit 21A, an IGBT is provided as the auxiliary switching element Q2. Further, the self-excited oscillation drive circuit is configured by connecting a parallel resonant circuit (NB2 // Cg) to the resistor Rg as in the case shown in FIGS. 6 and 7.

【0083】

Even in each of the configurations shown in FIGS. 7 and 8 above, the descent time tf at turn-off of the clamp current (drain current or collector current) flowing through the auxiliary switching element Q2 in the active clamp circuits 21A and 21B is large. As a result, the same effects as those of the respective embodiments described above can be obtained.

【0084】

In the present embodiment, an orthogonal type control transformer is used as a control transformer for performing constant voltage control, but instead of this orthogonal type control transformer, the oblique crossing previously proposed by the present applicant. A shape control transformer can be adopted. Although not shown here, the structure of the oblique control transformer is three-dimensional by combining two sets of double U-shaped cores having four magnetic legs, as in the case of an orthogonal control transformer, for example. Form a mold core. Then, the control winding NC and the drive winding NB are wound around this three-dimensional core, and at this time, the relationship between the winding directions of the control winding and the drive winding is diagonally intersecting. To be done. Specifically, one of the control winding NC and the drive winding NB is wound around two magnetic legs that are adjacent to each other among the four magnetic legs. The other winding is wound around two magnetic legs that are said to be in a diagonal positional relationship. When such an oblique control transformer is provided, the inductance of the drive winding increases even when the alternating current flowing through the drive winding changes from a negative current level to a positive current level. Is obtained. As a result, the current level in the negative direction for turning off the driven switching element increases, and the accumulation time of the switching element is shortened. Is also shortened, and the power loss of the switching element can be further reduced.

【0085】

Further, the embodiment of the present invention is not limited to the configuration shown in each figure. For example, in the above embodiment, bipolar transistors, MOS-FETs, IGBTs, etc. are used for the main switching element and the auxiliary switching element, but in addition, for example, SIT (static induction thyristor), etc. It is also conceivable to adopt other elements. Further, the configuration of the switching drive unit for driving the main switching element Q1 by the separately excited type does not have to be limited to the one shown in each figure, and may be appropriately changed to an appropriate circuit configuration. Further, the combination of the types of elements for the main switching element Q1 and the auxiliary switching element Q2 is not limited to the configurations shown in the above figures. Further, the secondary side rectifier circuit formed including the secondary side resonance circuit is not limited to the configuration shown in each figure as the embodiment, and other circuit configurations may be adopted. There is no such thing.

【0086】

[Effect of the invention]

As described above, the present invention forms a composite resonance type switching converter by providing a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side, and is active with respect to the primary side or the secondary side. A clamp circuit is provided. Then, on this configuration, the self-excited oscillation drive circuit that drives the auxiliary switching element in the active clamp circuit is provided with an LCR series resonance circuit or a parallel resonance circuit (drive resonance circuit). Further, the drive resonance circuit is set to have a resonance frequency equivalent to the switching frequency of the main switching element. With such a configuration, the drop time of the switching output current (drain current or collector current) at the time of turn-off of the auxiliary switching element can be significantly shortened, and the switching loss can be reduced. As a result, the DC-DC conversion efficiency of the switching power supply circuit can be improved to the same level as when the active clamp circuit is not provided, for example. In addition, the input power can be reduced accordingly. Furthermore, since the heat generation of the auxiliary switching element is reduced by reducing the switching loss, it is possible to eliminate the heat sink that should be provided for the auxiliary switching element, resulting in cost reduction and circuit miniaturization and weight reduction. It becomes possible to plan.

[Simple explanation of drawings]

[Figure 1]

It is a circuit diagram which shows the structure of the switching power supply circuit as the 1st Embodiment of this invention.

[Figure 2]

It is a waveform diagram which shows the operation of the main part in the power supply circuit shown in FIG.

[Fig. 3]

It is a circuit diagram which shows the internal structure example of a composite transistor.

[Fig. 4]

It is a circuit diagram which shows the structure of the switching power supply circuit as the 2nd Embodiment.

[Fig. 5]

It is a circuit diagram which shows the structure of the switching power supply circuit as the 3rd Embodiment.

[Fig. 6]

It is a circuit diagram which shows the structure of the switching power supply circuit as 4th Embodiment.

[Fig. 7]

It is a circuit diagram which shows the structure of the switching power supply circuit as the 5th Embodiment.

[Fig. 8]

It is a circuit diagram which shows the structure of the switching power supply circuit as the 6th Embodiment.

[Fig. 9]

It is sectional drawing which shows the structure of the insulation converter transformer.

[Fig. 10]

It is an equivalent circuit diagram which shows each operation when the mutual inductance is + M / -M.

[Fig. 11]

It is a circuit diagram which shows the structure of the switching power supply circuit as a prior art.

[Fig. 12]

It is a waveform diagram which shows the operation of the main part in the power supply circuit shown in FIG.

[Explanation of symbols]

1 control circuit, 10 switching drive, 11 oscillator circuit, 12 drive circuit, 20,20A, 20B, 21,21A, 21B, 22 active capacitor circuit, 30 composite transistor, Q1 main switching element, Q2 auxiliary switching element, PIT insulation Converter transformer, PRT orthogonal control transformer, Cr primary side parallel resonance capacitor, C2 secondary side parallel resonance capacitor, CCL clamp capacitor, NB2 drive winding, Cg resonance capacitor, Rg, R1 resistor