Switching power circuit

Description: TECHNICAL FIELD [Detailed description of the invention]

【0001】

[Technical field to which the invention belongs]

The present invention relates to a switching power supply circuit provided as a power supply in various electronic devices.

【0002】

[Conventional technology]

As a switching power supply circuit, a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular wave-like switching operation waveform, there is a limit to suppressing switching noise. It is also known that there is a limit to the improvement of power conversion efficiency due to its operating characteristics. Therefore, the applicant has previously proposed various switching power supply circuits using a resonant converter. The resonance type converter can easily obtain high power conversion efficiency, and low noise is realized by making the switching operation waveform sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

【0003】

The circuit diagram of FIG. 7 shows an example of a switching power supply circuit as a prior art that can be configured based on the invention previously proposed by the applicant. As the basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

【0004】

In the power supply circuit shown in this figure, the bridge rectifier circuit Di and the smoothing capacitor Ci generate a rectified smoothing voltage Ei corresponding to one level of the AC input voltage VAC from the commercial AC power supply (AC input voltage VAC).

【0005】

On the primary side of this power supply circuit, a self-excited configuration is shown as a voltage resonance type converter circuit that performs single-ended operation by a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is used for the switching element Q1. The base of the switching element Q1 is connected to the positive electrode side of the smoothing capacitor Ci (rectified smoothing voltage Ei) via the starting resistor RS so that the base current at startup is obtained from the rectified smoothing line.

【0006】

Also, between the base of the switching element Q1 and the ground on the primary side, the drive winding NB provided on the primary side of the insulation converter transformer PIT with a number of turns of 1T (turn) and the inductor LB-resonant capacitor CB-base current limiting resistor RB A series resonance circuit for self-excited oscillation drive consisting of the series circuit of is connected. This self-excited oscillation circuit generates a switching frequency fs that turns on / off the switching element Q1.

【0007】

In addition, the clamp diode DD1 inserted between the base of the switching element Q1 and the negative electrode (primary side ground) of the smoothing capacitor Ci forms a path for the clamp current that flows when the switching element Q1 is off. , The collector of the switching element Q1 is connected to the winding start end of the primary winding N1 of the isolated converter transformer PIT, and the emitter is grounded.

【0008】

Further, a parallel resonant capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. Also in this case, the capacitance of the parallel resonant capacitor Cr itself and the leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT form a primary side parallel resonant circuit of the voltage resonant converter .

【0009】

The isolated converter transformer PIT is provided to transmit the switching output of the switching converter obtained on the primary side to the secondary side. As the structure of this isolated converter transformer PIT, for example, as shown in FIG. 8, an EE type core composed of ferrite E type cores CR1 and CR2 is provided. Then, as shown in the figure, the split bobbin B is used to wind the primary winding N1 and the secondary winding N2, both of which are litz wires, around the divided regions. Here, both the primary winding N1 and the secondary winding N2 are wound in the same winding direction. Then, a gap G is formed with respect to the central magnetic leg of the EE type core as shown in the figure. The leakage inductance in the insulation converter transformer PIT is determined by the gap length of this gap G, and loose coupling with the required coupling coefficient can be obtained. As the coupling coefficient k here, for example, a loosely coupled state of k0.85 is obtained, which makes it difficult to obtain a saturated state. This gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs, and the gap length in this case is about 1 mm.

【0010】

By the way, as an operation in the isolated converter transformer PIT, the inductance of the primary winding N1 and the inductance of the secondary winding N2 depend on the connection relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO. Regarding the mutual inductance M with L2, there are cases where the operation mode is + M (polarization mode: forward operation) and the operation mode is -M (depolarization mode: flyback operation). For example, assuming that the primary winding N1 and the secondary winding N2 have the same polarity (winding direction), the mutual inductance becomes + M when the circuit is equivalent to the circuit shown in FIG. 9 (a), and FIG. 9 (b). The mutual inductance is -M when it is equivalent to the circuit shown in.

【0011】

As shown in FIG. 7, the winding start end of the primary winding N1 of the isolated converter transformer PIT is connected to the collector of the main switching element Q1, and the winding end is connected to the line of the rectifying smoothing voltage Ei. Further, the winding start end of the secondary winding N2 is connected to the secondary ground, and the winding end is connected to the positive electrode terminal of the smoothing capacitor CO1 via the rectifying diode DO1. In such a connection form, the primary winding N1 and the secondary winding N2 of the isolated converter transformer PIT are connected by polarity, which is the equivalent circuit shown in FIG. 9 (a). It corresponds to.

【0012】

The switching output of the main switching element Q1 forming the primary voltage resonance type converter is transmitted to the primary winding N1 of the insulation converter transformer PIT having the above structure, and is further excited with respect to the secondary winding N2. It will be transmitted. In this case, on the secondary side of the isolated converter transformer PIT, the leakage inductance of the secondary winding N2 is formed by connecting the secondary side parallel resonant capacitor C2 in parallel with the secondary winding N2 as shown in the figure. Form a secondary parallel resonant circuit with L2. Then, a half-wave rectifier circuit composed of a rectifier diode DO1 and a smoothing capacitor CO1 is connected to the secondary side parallel resonant circuit according to the connection form shown in the figure, thereby outputting the secondary side DC output voltage EO1.

【0013】

In the power supply circuit having such a configuration, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is provided with a parallel resonance circuit for obtaining the voltage resonance operation. Become. In this specification, a switching converter having a configuration in which resonance circuits are provided on the primary side and the secondary side to operate in this way is also referred to as a composite resonance type switching converter.

【0014】

Further, in this power supply circuit, an active clamp circuit 20 is provided on the secondary side. That is, the secondary side active clamp circuit 20 includes an auxiliary switching element Q2 for MOS-FET, a clamp capacitor C3, and a clamp diode DD2 for a body diode. Further, as a drive circuit system for driving the auxiliary switching element Q2, a drive winding Ng1, a capacitor Cg1, and a resistor Rg1 are provided.

【0015】

A clamp diode DD2 is connected in parallel between the drain and the source of the auxiliary switching element Q2. The connection form is such that the anode of the clamp diode DD2 is connected to the source and the cathode is connected to the drain. Further, the drain of the auxiliary switching element Q2 is connected to the connection point between the winding end end of the secondary winding N2 and the anode of the rectifying diode DO1 via the clamp capacitor C3. Further, the source of the auxiliary switching element Q2 is connected to the secondary ground. Therefore, the secondary side active clamp circuit 20 is formed by connecting the clamp capacitor C3 in series with the parallel connection circuit of the auxiliary switching element Q3 and the clamp diode DD2. Then, the circuit formed in this way is further connected in parallel to the secondary side parallel resonant circuit.

【0016】

As a drive circuit system for the auxiliary switching element Q2, as shown in the figure, a series connection circuit of a capacitor Cg1-resistor Rg1-drive winding Ng1 is connected to the gate of the auxiliary switching element Q2. This series connection circuit forms a self-excited drive circuit for the auxiliary switching element Q2. That is, the signal voltage from this self-excited drive circuit is applied to the gate of the switching element Q2 to perform the switching operation. The drive winding Ng1 in this case is formed on the winding start end side of the secondary winding N2, and the number of turns in this case is, for example, 1T (turn). As a result, a voltage excited by the alternating voltage obtained in the primary winding N1 is generated in the drive winding Ng1. Further, in this case, a voltage having opposite polarities between the secondary winding N2 and the drive winding Ng1 can be obtained due to the relationship of the winding directions.

【0017】

In addition, the switching operation of the auxiliary switching element Q2 is PWM-controlled by the control circuit 1 provided on the secondary side. That is, the secondary side DC output voltage EO1 is supplied to the control circuit 1, and the control circuit 1 applies the corresponding DC control voltage to the gate of the auxiliary switching element Q2 to control the conduction angle of the auxiliary switching element Q2. .. As a result, the DC output voltage E01 is made constant with respect to fluctuations in the AC input voltage VAC and the load power Po.

【0018】

[Problems to be Solved by the Invention]

By the way, in the power supply circuit shown in FIG. 7, the winding directions of the primary winding N1 and the secondary winding N2 are the same as shown by the structure of the isolated converter transformer PIT in FIG. Therefore, the primary winding current I1 flowing through the primary winding N1 creates a magnetomotive force in the primary winding N1, and similarly, the secondary winding current I2 flowing through the secondary winding N2 causes the secondary winding N2. A magnetomotive force is generated against the coil. As a result, as shown in FIG. 8, a primary magnetic flux φ1 is generated on the primary side, and a secondary magnetic flux φ2 is generated on the secondary side. Here, as described above, since the primary winding N1 and the secondary winding N2 in the circuit of FIG. 7 are connected with a positive polarity, the primary magnetic flux φ1 and the secondary magnetic flux φ2 described above are connected to each other. Therefore, the magnetic flux represented by φ1 + φ2 is generated in the central magnetic leg of the insulating converter transformer PIT.

【0019】

That is, since the primary winding N1 and the secondary winding N2 are in the same winding direction and have a positive polarity connection, the primary magnetic flux φ1 and the secondary magnetic flux φ2 are added to the central magnetic leg, which is relatively large. Magnetic flux will be generated. Here, assuming that a gap is not formed in the central magnetic leg of the core of the insulation converter transformer PIT (gap length = 0), for example, under the condition of load power Po = 100 W or more, the saturation region of the magnetization curve of the ferrite core Will enter. In addition, in this specification, the term "saturation" refers to a state of entering the saturation region of such a magnetization curve. As a result, the inductance of the core drops sharply, increasing the possibility that the main switching element Q1 of the BJT will be destroyed. Therefore, as the isolated converter transformer PIT, by forming a gap G as shown in FIG. 8, a loosely coupled state with a required coupling coefficient can be obtained so that saturation does not occur. I have to.

【0020】

Then, in the case of the power supply circuit having the configuration shown in FIG. 7, in order to avoid the above phenomenon and satisfy the regulation range, the gap length of the gap G formed in the insulation converter transformer PIT has an accuracy of 1 mm ± 0.1 mm. It is necessary to manage with. In order to satisfy the above-mentioned accuracy of the gap length, it is necessary for the E-type cores CR1 and CR2 to be manufactured and controlled with an accuracy of 0.5 mm ± 0.05 mm by polishing the ends of each central magnetic leg. .. Therefore, it takes a long time to manufacture because the process of polishing the central magnetic leg of the E-type core with high accuracy is required, and it corresponds to the case where the insulation converter transformers having the same E-type core with different gap lengths are manufactured. It also becomes difficult to manage the product. In other words, the need to form a gap leads to a decrease in manufacturing efficiency.

【0021】

Further, when a gap G is formed in the isolated converter transformer PIT, a leakage magnetic flux called a fringe magnetic flux is generated in the vicinity of the gap G, so that eddy currents are generated in the primary winding N1 and the secondary winding N2, which are litz wires. Loss occurs and local heat generation occurs. This heat generation is conducted to the winding having a low temperature, and the temperature of the winding itself is raised. It is known that this increases the so-called copper loss, which is a power loss, and lowers the power conversion efficiency. In particular, in the circuit shown in FIG. 7, since the primary winding current I1 flowing through the primary winding N1 and the high-frequency current flowing through the secondary winding current I2 flowing through the secondary winding N2 are large, the primary winding current I1 The heat generated by the DC resistance as a litz wire and the above-mentioned eddy current loss in the secondary winding N2 is remarkable.

【0022】

Furthermore, in the circuit shown in FIG. 7, when the level of the AC input voltage VAC drops to about 75V to 85V in the AC100 system under heavy load conditions, ZVS (Zero) is used as the operation of the primary side main switching element Q1. It is also a problem that there is a period of abnormal operation that does not result in Voltage Switching) operation. If such a phenomenon continues, the main switching element Q1 may generate heat and be destroyed in a short time.

【0023】

[Means for solving problems]

Therefore, in consideration of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a switching means formed with a main switching element for intermittently outputting the DC input voltage and a primary side parallel resonance circuit having the operation of the switching means as a voltage resonance type are formed. It has a structure in which a parallel resonance capacitor on the primary side is obtained and a required coupling coefficient that is loosely coupled on the primary side and the secondary side can be obtained, and the output of the switching means obtained on the primary side is transmitted to the secondary side. Obtained for the insulation converter transformer, the secondary resonance circuit formed by connecting the secondary resonance capacitor to the secondary winding of the insulation converter transformer, and the secondary winding of the insulation converter transformer. The DC output voltage generating means configured to obtain the secondary side DC output voltage by inputting the alternating voltage to be performed and performing the rectification operation, and the clamp capacitor and the secondary side in parallel with the above secondary side resonance capacitor. A secondary side active clamping means formed by providing a series connection circuit with a side auxiliary switching element, and a DC control signal based on the secondary side DC output voltage is applied to the secondary side auxiliary switching element to be applied to the secondary side auxiliary switching element. It is provided with a constant voltage conversion means for converting the secondary side DC output voltage to a constant voltage by executing the conduction angle control of the side auxiliary switching element. The isolated converter transformer is provided with a core that is not provided with a gap for suppressing saturation, and the primary winding and the secondary winding are wound around the core in opposite winding directions. , The primary winding and the secondary winding are polarly connected.

【0024】

According to the above configuration, the primary side is provided with a primary side parallel resonance circuit for forming a voltage resonance type converter, and the secondary side is a secondary formed by a secondary winding and a secondary side parallel resonance capacitor. A so-called composite resonance type switching converter configuration provided with a side parallel resonance circuit can be obtained. Further, an active clamp circuit is provided on the secondary side, and constant voltage control is performed by controlling the conduction angle of the auxiliary switching element of the active clamp circuit. Based on this configuration, for the isolated converter transformer, the primary winding and the secondary winding are wound in opposite winding directions, and then the primary winding and the secondary winding are polar-connected. .. As a result, the magnetic fluxes obtained in the primary winding and the secondary winding act to cancel each other out, so that the magnetic flux generated in the core can be made small, and the saturation state can be suppressed accordingly. .. Therefore, the core of the isolated converter transformer in the switching power supply circuit of the present invention is not provided with a gap for the purpose of suppressing saturation.

【0025】

BEST MODE FOR CARRYING OUT THE INVENTION

FIG. 1 shows a configuration example of a switching power supply circuit as the first embodiment of the present invention. The power supply circuit shown in FIG. 1 is configured as a composite resonance type switching converter having a voltage resonance type converter on the primary side and an active clamp circuit and a voltage resonance circuit on the secondary side.

【0026】

In the power supply circuit shown in this figure, the bridge rectifier circuit Di and the smoothing capacitor Ci generate a rectified smoothing voltage Ei corresponding to one level of the AC input voltage VAC from the commercial AC power supply (AC input voltage VAC).

【0027】

On the primary side of this power supply circuit, a self-excited configuration is shown as a voltage resonance type converter circuit that performs single-ended operation by a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is used for the switching element Q1. The base of the switching element Q1 is connected to the positive electrode side of the smoothing capacitor Ci (rectified smoothing voltage Ei) via the starting resistor RS so that the base current at startup is obtained from the rectified smoothing line.

【0028】

Also, between the base of the switching element Q1 and the ground on the primary side, the drive winding NB provided on the primary side of the insulation converter transformer PIT with a number of turns of 1T (turn) and the inductor LB-resonant capacitor CB-base current limiting resistor RB A series resonance circuit for self-excited oscillation drive consisting of the series circuit of is connected. This self-excited oscillation circuit generates a switching frequency fs that turns on / off the switching element Q1. The switching frequency fs is fixed at about 100 KHz.

【0029】

In addition, the clamp diode DD1 inserted between the base of the switching element Q1 and the negative electrode (primary side ground) of the smoothing capacitor Ci forms a path for the clamp current that flows when the switching element Q1 is off. , The collector of the switching element Q1 is connected to the end of the winding end of the primary winding N1 of the isolated converter transformer PIT, and the emitter is grounded.

【0030】

Further, a parallel resonant capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. Also in this case, the capacitance of the parallel resonant capacitor Cr itself and the leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT form a primary side parallel resonant circuit of the voltage resonant converter.

【0031】

The isolated converter transformer PIT transmits the switching output of the main switching element Q1 to the secondary side. The present embodiment is characterized by the structure of the isolated converter transformer PIT, which will be described later. The winding end end of the primary winding N1 of the isolated converter transformer PIT is connected to the collector of the main switching element Q1, and the winding start end is connected to the positive electrode (rectified smoothing voltage Ei) of the smoothing capacitor Ci.

【0032】

On the secondary side of the isolated converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2 are used. A parallel resonant circuit is formed. By this parallel resonance circuit, the alternating voltage induced in the secondary winding N2 becomes the resonance voltage. That is, a voltage resonance operation can be obtained on the secondary side. That is, in this power supply circuit, a parallel resonance circuit for making the switching operation a voltage resonance type is provided on the primary side, and a parallel resonance circuit for obtaining a voltage resonance operation is provided on the secondary side. It has a configuration as a "resonant switching converter".

【0033】

For the secondary side of the power supply circuit formed as described above, a half-wave rectifier circuit composed of a rectifier diode DO1 and a smoothing capacitor CO1 is provided to obtain a secondary side DC output voltage EO1. The DC output voltage EO1 is also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as the detection voltage and the operating power source of the control circuit 1.

【0034】

Further, in this power supply circuit, an active clamp circuit 20 is provided on the secondary side. That is, the secondary side active clamp circuit 20 includes an auxiliary switching element Q2 for MOS-FET, a clamp capacitor C3, and a clamp diode DD2 for a body diode. Further, as a drive circuit system for driving the auxiliary switching element Q2, a drive winding Ng1, a capacitor Cg1, and a resistor Rg1 are provided.

【0035】

A clamp diode DD2 is connected in parallel between the drain and the source of the auxiliary switching element Q2. The connection form is such that the anode of the clamp diode DD2 is connected to the source and the cathode is connected to the drain. Further, the drain of the auxiliary switching element Q2 is connected to the connection point between the winding end end of the secondary winding N2 and the anode of the rectifying diode DO1 via the clamp capacitor C3. Further, the source of the auxiliary switching element Q2 is connected to the secondary ground. Therefore, the secondary side active clamp circuit 20 is formed by connecting the clamp capacitor C3 in series with the parallel connection circuit of the auxiliary switching element Q3 and the clamp diode DD2. Then, the circuit formed in this way is further connected in parallel to the secondary side parallel resonance circuit (resonance capacitor C2).

【0036】

As a drive circuit system for the auxiliary switching element Q2, as shown in the figure, a series connection circuit of a capacitor Cg1-resistor Rg1-drive winding Ng1 is connected to the gate of the auxiliary switching element Q2. This series connection circuit forms a self-excited drive circuit for the auxiliary switching element Q2. That is, the signal voltage from this self-excited drive circuit is applied to the gate of the switching element Q2 to perform the switching operation. The drive winding Ng1 in this case is formed on the winding start end side of the secondary winding N2, and the number of turns in this case is, for example, 1T (turn). As a result, a voltage excited by the alternating voltage obtained in the primary winding N1 is generated in the drive winding Ng1. Further, in this case, a voltage having opposite polarities between the secondary winding N2 and the drive winding Ng1 can be obtained due to the relationship of the winding directions. Even if the drive winding Ng1 is used, its operation is guaranteed if the number of turns is 1T, but the operation is not limited to this.

【0037】

In addition, the switching operation of the auxiliary switching element Q2 is PWM-controlled by the control circuit 1 provided on the secondary side. That is, the secondary side DC output voltage EO1 is supplied to the control circuit 1, and the control circuit 1 applies the corresponding DC control voltage to the gate of the auxiliary switching element Q2 to control the conduction angle of the auxiliary switching element Q2. .. As a result, the DC output voltage E01 is made constant with respect to fluctuations in the AC input voltage VAC and the load power Po. This is a system with a very fast transient response to sudden fluctuations in load power.

【0038】

In the case of such a configuration, for example, the switching frequency is fixed, and the on period is variable while the off period of the auxiliary switching element Q2 is constant according to the level change of the secondary side DC output voltage EO1 due to load fluctuation or the like. The operation of controlling is obtained. That is, with respect to the switching operation of the auxiliary switching element Q2, an operation of variably controlling the conduction angle can be obtained. Here, for example, assuming that the level of the secondary side DC output voltage EO1 rises in a light load state, the conduction angle control is performed so that the ON period of the auxiliary switching element Q2 is extended.

【0039】

Then, as a result of performing PWM control as described above, as a voltage induced in the secondary winding N2 of the isolated converter transformer PIT, the pulse width of the negative waveform is expanded, and the pulse width of the positive waveform is expanded. Becomes shorter. In the secondary side rectifying diode DO1, this secondary side parallel resonance voltage is input and rectification is performed by forward operation. Therefore, the period during which the secondary side rectifying diode DO1 is conducted and turned on is shortened, and one of them is turned off. The period will be extended. In this way, as a result, the conduction angle of the rectifier diode DO1 is controlled, so that the secondary side DC output voltage can be stabilized.

【0040】

In the configuration in which the active clamp circuit 20 is provided on the secondary side in this way, the peak level of the resonance pulse of the secondary side parallel resonant circuit (N2 // C2) generated during the period when the rectifier diode DO1 is turned off is active. Compared with the configuration when the clamp circuit is not provided, the configuration can be reduced to about 1/2. Further, in the present embodiment, the self-excited oscillation drive circuit in the active clamp circuit 20A provided on the secondary side also has an LCR resonance circuit (Rg-Cg-Lg) configuration, so that auxiliary switching is performed. The switching loss due to the element Q2 is reduced, and the DC-DC power conversion efficiency of the power supply circuit is improved to the same level as in the case where the active clamp circuit is not provided.

【0041】

FIG. 2 shows the structure of the isolated converter transformer PIT provided in the power supply circuit shown in FIG. In this figure, for convenience of explanation, the drive winding Ng1 is not shown, and the primary winding N1 and the secondary winding N2 are shown. As shown in the figure, the insulation converter transformer PIT constitutes an EE type core with two E type cores CR1 and CR2. Then, a split bobbin B is provided for this EE type core, and as shown in the figure, for example, the primary winding N1 is wound around the winding region on the E type core CR1 side of the split bobbin B, and the E type core is wound. The secondary winding N2 is wound around the winding area on the CR2 side. Then, in the case of the present embodiment, the winding directions of the primary winding N1 and the secondary winding N2 are opposite to each other as shown by the arrows on the left and right outer sides of the core in the drawing. , So-called reverse winding structure. Further, in the case of the present embodiment, no gap is formed at the opposite portions of the central magnetic legs of the E-type cores CR1 and CR2.

【0042】

Here again, with reference to FIG. 1, the connection between the primary winding N1 and the secondary winding N2 of the isolated converter transformer PIT will be described. As shown in FIG. 1, for example, the connection between the winding start end and the winding end end of the primary winding N1 is the reverse of the case of the circuit shown in FIG. 7 as the prior art. That is, in the circuit shown in FIG. 1, the winding start end of the primary winding N1 is connected to the positive electrode terminal of the smoothing capacitor Ci, and the winding end is connected to the collector of the main switching element Q1. is there. Further, as the secondary winding N2, the winding end end is connected to the positive electrode terminal of the smoothing capacitor CO1 via the rectifying diode DO1, and the winding start end is connected to the secondary ground. .. That is, even when the power supply circuit shown in FIG. 1 is provided with an isolated converter transformer PIT in which the primary winding N1 and the secondary winding N2 have a reverse winding structure as shown in FIG. 2 above. The primary winding N1 and the secondary winding N2 are connected so as to have the polarity shown in the equivalent circuit of FIG. 9 (a).

【0043】

According to such a configuration, the polarity of the primary magnetic flux φ1 generated by the primary winding current I1 flowing through the primary winding N1 and the polarity of the secondary magnetic flux φ2 generated by the secondary winding current I2 flowing through the secondary winding N2. Is like the arrow shown in the core in Figure 2. This is because the polarity of the primary magnetic flux φ1 is reversed as compared with the case of the isolated converter transformer PIT shown in FIG. 8 as the prior art. The polarity of the secondary magnetic flux φ2 is the same as that of the isolated converter transformer PIT in FIG. In the present embodiment, the polar relationship between the primary magnetic flux φ1 and the secondary magnetic flux φ2 as shown in FIG. 2 can be obtained. As a result, the primary magnetic flux φ1 and the secondary magnetic flux φ2 can be obtained. , Both act to cancel each other out. That is, the magnetic flux (Δφ) obtained in the central magnetic leg of the isolated converter transformer PIT is | φ1-φ2 | = Δφ Will be represented by. This indicates that, as described above, the primary magnetic flux φ1 and the secondary magnetic flux φ2 cancel each other out, and do not add to each other as in the case of the circuit of FIG. 7, for example. Therefore, in the present embodiment, the magnetic flux obtained in the central magnetic leg of the insulating converter transformer PIT can be weaker than before. As a result, for example, a loosely coupled state of about k = 0.8 to 0.9 can be obtained as the coupling coefficient k between the primary side and the secondary side. As a result, the isolated converter transformer PIT of the present embodiment can prevent the core from being saturated even if a gap is not intentionally formed with respect to the central magnetic leg, and as a result, as shown in FIG. Therefore, no gap is provided. Actually, for example, on the joint surface of the central magnetic leg with the gap length set to 0, so-called core squeal of audible sound may occur. Therefore, for example, a Mylar film is applied to the joint surface of the central magnetic leg. Therefore, a gap having a gap length of 0.1 mm or less may be formed.

【0044】

Then, by configuring the isolated converter transformer PIT in this way, the magnetic flux obtained in the central magnetic leg becomes much weaker than before, so that the periphery of the gap, for example, as in the case of FIG. The temperature rise of the winding due to the fringe magnetic flux generated in the above and the accompanying decrease in the power conversion efficiency are also eliminated.

【0045】

Further, in the insulated converter transformer PIT of the present embodiment, since the magnetic flux (Δφ) obtained in the central magnetic leg is weak, the leakage inductance of the primary winding N1 and the secondary winding N2 is also reduced. Become. As a result, stable ZVS operation can be realized as the main switching element Q1 even under a heavy load condition of, for example, a load power of about 200 W.

【0046】

FIG. 3 is a waveform diagram showing the operation of a main part in the power supply circuit of FIG. 1 according to the above configuration. Here, the operation is shown under the conditions of AC input voltage VAC = 220V and load power Po = 200W. For comparison, the waveform in the case of the power supply circuit shown in FIG. 7 is also shown by the alternate long and short dash line.

【0047】

Due to the switching operation of the main switching element Q1, a resonance pulse voltage VQ1 is generated at both ends of the primary side parallel resonance capacitor Cr at a fixed cycle as shown in FIG. 3 (a). At this time, the collector current I1 flowing through the main switching element Q1 is as shown in FIG. 3 (b). That is, when the main switching element Q1 is turned on, a damper current (negative direction) flows through the primary winding N1 via the clamp diode DD1 and the base-collector of the main switching element Q1, and when the period in which this damper current flows ends, the collector current I1 The level rises sharply from negative to positive. By performing such a switching operation, the resonance current I2 shown in Fig. 3 (d) flows through the secondary winding N2 of the isolated converter transformer PIT, and the resonance current I2 shown in Fig. 3 (c) flows through the secondary side parallel resonant capacitor C2. Such resonance voltage V2 is generated. Then, during the positive period in which the rectifier diode D01 operates, a voltage clamped to the voltage E01 level is obtained.

【0048】

In addition, when the active clamp circuit 20 becomes conductive, the clamp current IQ2 flows in the path from the clamp diode DD2 to the clamp capacitor C3, which is a saw tooth that flows from the negative direction to the positive direction with the passage of time as shown in Fig. 3 (e). It becomes a diode. When the active clamp circuit 20 is conducting, most of the current flows through the clamp capacitor C3 as the clamp current IQ2, and hardly flows through the secondary side parallel resonant capacitor C2. Therefore, the resonance voltage V2 is clamped even during the period when the active clamping circuit 20 is conducting, and as a result, the negative voltage level is limited as shown in FIG. 3 (c).

【0049】

Further, FIG. 4A shows the primary side parallel resonance voltage VQ1 and the switching output current IQ1 flowing through the main switching element Q1. The conditions at this time are when the load power is Po = 200W and the AC input voltage of the AC100V system is reduced to about 75V to 85V. Note that FIG. 4 (b) shows the waveform of the power supply circuit of FIG. 7 for comparison.

【0050】

As can be seen from the waveform in Fig. 4 (b), in the case of the circuit in Fig. 7, the primary side parallel resonance voltage VQ1 and the switching output current are at the timing when the switching output current IQ1 reverses from the negative electrode level to the positive electrode level in the period TON. There is a phenomenon that IQ1 appears as a pulse at a positive level. In other words, it is an abnormal operation that does not result in ZVS operation. On the other hand, in the circuit shown in FIG. 1, as shown in FIG. 4 (a), the pulse of the primary side parallel resonance voltage VQ1 in the period TON disappears, and the waveform of the switching output current IQ1 also disappears. , The pulse does not appear and it is normal. That is, in this embodiment, it is shown that the ZVS operation is normally performed even under the condition of a heavy load and a low AC input voltage.

【0051】

Here, the specifications of the main parts of the power supply circuit shown in FIG. 1 will be shown. First, for the isolated converter transformer PIT, after adopting the core of EE-40, the gap length is set to Gap = 0, and the number of turns of the primary winding N1 and the secondary winding N2 is N1 = 50T, N2 = It is set to 45T. Further, the primary side parallel resonance capacitor Cr = 5600pF, the secondary side parallel resonance capacitor C2 = 8200pF, and the clamp capacitor C3 = 0.27μF. Further, in the power supply circuit shown in FIG. 7 above, the core of EE-40 is similarly adopted as the isolated converter transformer PIT, and Gap = 1 mm. Further, the primary winding N1 = secondary winding N2 = 45T, the primary side parallel resonance capacitor Cr = 6800pF, the secondary side parallel resonance capacitor C2 = 0.01μF, and the clamp capacitor C3 = 0.33μF.

【0052】

The power conversion efficiency of the power supply circuit shown in Fig. 1 was 91.9% when the load power was Po = 200W and VAC = 100V. On the other hand, in the circuit shown in Fig. 7, it was 90.8% at VAC = 100V under the load condition of load power Po = 200W. In other words, the power conversion efficiency is improved by 1.1% compared to the prior art. This is a reduction of about 2W in power loss.

【0053】

In addition, the temperature rise value in the isolated converter transformer PIT of the circuit shown in Fig. 1 shows a significant decrease of about 4 ° C for both the primary winding N1 and the secondary winding N2 compared to the circuit shown in Fig. 7. Has been done. Specifically, the primary winding N1 decreased from 45 ° C to 41 ° C, and the secondary winding N2 decreased from 52 ° C to 48 ° C.

【0054】

FIG. 5 shows a configuration example of a switching power supply circuit as a second embodiment of the present invention. In this figure, the same parts as those in FIG. 1 are designated by the same reference numerals, and the description thereof will be omitted. This is an example of a composite resonance type converter circuit in which the primary side is a separately excited oscillation type voltage resonance converter using an IC and MOS-FET, and the secondary side is an active clamp circuit 20 using MOS-FET and a voltage doubler rectification type current resonance circuit. It becomes.

【0055】

The primary side voltage resonance type converter of the power supply circuit shown in this figure adopts a single-ended system configuration by a separately excited type. In this case, a MOS-FET is used as the main switching element Q1. The drain of the main switching element Q1 as this MOS-FET is connected to the winding end end of the primary winding N1, and the source is connected to the primary side ground. Further, the parallel resonant capacitor Cr is connected in parallel between the drain and the source of the main switching element Q1. The clamp diode DD1 is also connected in parallel with respect to the drain and source of the main switching element Q1.

【0056】

The switching drive unit 2 is provided to drive the main switching element Q1 by a separately excited type, and can be configured as, for example, a one-stone IC. The switching drive unit 2 includes an oscillation circuit 3 and a drive circuit 4. At the time of start-up, the switching drive unit 2 is adapted to obtain start-up power from the line of the rectified smoothing voltage Ei via the start-up resistor Rs.

【0057】

The oscillation circuit 3 generates an oscillation signal and outputs it to the drive circuit 4. In the drive circuit 4, the input oscillation signal is converted into a drive voltage that can drive the main switching element Q1 which is a MOS-FET, and is output to the gate of the main switching element Q1. As a result, the main switching element Q1 is switched and driven at a predetermined switching frequency fs based on the oscillation signal.

【0058】

Further, on the secondary side of the circuit shown in FIG. 5, the secondary side series resonant capacitor Cs is connected in series to the winding start end of the secondary winding N2, and the leakage inductance of the secondary winding N2. The secondary side series resonance circuit (current resonance circuit) is formed by the capacitance of L2 and the secondary side series resonance capacitor Cs. That is, the power supply circuit shown in this figure adopts a configuration in which a voltage resonance circuit is provided on the primary side and a current resonance circuit is provided on the secondary side as a composite resonance type switching converter.

【0059】

The secondary rectifier circuit in this case is formed by connecting two rectifier diodes DO1 and D02 and a smoothing capacitor C01 as shown in the figure. Depending on such a connection form, a configuration as a so-called voltage doubler half-wave rectifier circuit can be obtained. In this voltage doubler half-wave rectifier circuit, the secondary side series resonant capacitor Cs is charged by the current rectified by the rectifying diode DO2 in the half cycle of the alternating voltage obtained in the secondary winding N2. Then, in the next half cycle, the rectifier diode D01 conducts with the potential obtained to the secondary side series resonance capacitor Cs, and the smoothing capacitor CO1 is charged. By such an operation, the secondary side DC output voltage EO1 which is the voltage across the smoothing capacitor CO1 can be obtained at a level corresponding to twice the alternating voltage level obtained in the secondary winding N2. .. Therefore, when the voltage doubler half-wave rectifier circuit is provided on the secondary side in this way, the secondary side DC output voltage EO1 may be the same level as that obtained by the equal voltage voltage rectifier circuit. , The number of turns of the secondary winding N2 can be reduced to about half of the normal number.

【0060】

Further, the components of the active clamp circuit 20 are the same as those in the example of FIG. Then, the series circuit of the clamp capacitor C3 and the auxiliary switching element Q2 constituting the active clamp circuit 20 is connected in parallel with the resonance capacitor Cs. That is, the clamp capacitor C3 is connected to the connection point between the secondary winding N2 and the resonance capacitor Cs. As a result, the charge charge to the resonance capacitor Cs due to the resonance current generated in the secondary winding N2 is controlled. In addition, the switching operation of the auxiliary switching element Q2 is PWM-controlled by the control circuit 1 provided on the secondary side. That is, the secondary side DC output voltage EO1 is supplied to the control circuit 1, and the control circuit 1 applies the corresponding DC control voltage to the gate of the auxiliary switching element Q2 to control the conduction angle of the auxiliary switching element Q2. .. As a result, the DC output voltage E01 is made constant with respect to fluctuations in the AC input voltage VAC and the load power Po.

【0061】

Even in such a configuration, the isolated converter transformer PIT having the structure shown in FIG. 2 is provided, and the connection between the primary winding N1 and the secondary winding N2 is a polar connection. The same effect as the power supply circuit of is obtained.

【0062】

FIG. 6 shows a configuration example of a switching power supply circuit as a third embodiment. In this figure, the same parts as those in FIGS. 1 and 5 are designated by the same reference numerals and the description thereof will be omitted. This is an input voltage voltage doubler rectifier circuit, which is a configuration example in which the primary side is a voltage resonance circuit using an IGBT (insulated gate bipolar transistor), and the secondary side is an active clamp circuit 20 using an IGBT and a half-wave rectification type voltage resonance circuit. .. IGBTs are known to have high switching characteristics.

【0063】

In the circuit shown in this figure, first, as a rectifying and smoothing circuit system for AC input, rectifying diodes Di1 and Di2 and smoothing capacitors Ci1 and Ci2 are connected as shown to form a so-called voltage doubler rectifying and smoothing circuit. Further, the main switching element Q1 by the IGBT forming the primary side voltage resonance type converter is provided with the switching drive unit 2 by the oscillation circuit 3 and the drive circuit 4 as in the example of FIG.

【0064】

The collector of the main switching element Q1 by this IGBT is connected to the winding end end of the primary winding N1, and the emitter is connected to the primary side ground. Further, the parallel resonant capacitor Cr is connected in parallel between the collector and the emitter of the main switching element Q1. The clamp diode DD1 is also connected in parallel between the collector and the emitter of the main switching element Q1.

【0065】

The switching drive unit 2 is provided to drive the main switching element Q1 by a separately excited type, and is configured as, for example, a one-stone IC. At start-up, the switching drive unit 2 obtains start-up power from the line of the rectified smoothing voltage Ei via the start-up resistor Rs. Then, the oscillation circuit 3 generates an oscillation signal and outputs it to the drive circuit 4, and the drive circuit 4 converts the input oscillation signal into a drive voltage that can drive the main switching element Q1 to perform main switching. Output to the gate of element Q1. As a result, the main switching element Q1 is switched and driven at a predetermined switching frequency fs based on the oscillation signal. In such a configuration, for example, SIT (static induction thyristor)) or the like may be adopted as the main switching element Q1.

【0066】

Further, the configuration of the secondary side of the power supply circuit shown in this figure is basically the same as the configuration of FIG. However, the difference is that the IGBT is adopted as the auxiliary switching element Q2 of the active clamp circuit 20.

【0067】

Even in such a configuration, the isolated converter transformer PIT having the structure shown in FIG. 2 is provided, and the connection between the primary winding N1 and the secondary winding N2 is a polar connection. And the same effect as the power supply circuit of FIG. 5 can be obtained.

【0068】

Although the embodiment has been described above, as the power supply circuit of the present invention, for example, a combination of a primary side voltage resonance type converter and a secondary side rectifier circuit may be considered in various ways other than those shown in each figure. Is. Further, the drive system of the active clamp circuit is not limited to the self-excited configuration shown in each figure, and may be a configuration adopting another self-excited system, a separately excited system, or the like.

【0069】

[Effect of the invention]

As described above, the present invention relates to an isolated converter transformer of a switching power supply circuit having a composite resonance type and an active clamp circuit provided on the secondary side, and the primary winding and the secondary winding are so-called reverse windings. After that, the primary winding and the secondary winding are connected in a positive polarity. In this structure, the primary magnetic flux and the secondary magnetic flux act to cancel each other out. Therefore, in the present invention, it is not necessary to provide a gap for suppressing saturation in the core of the insulating converter transformer. Become. Since it is not necessary to provide a gap to the core of the insulating converter transformer as described above, in the present invention, the step for forming the gap is omitted in the manufacture of the isolated converter transformer, and the manufacturing It will be easier to manage. That is, the manufacturing efficiency of the power supply circuit including the isolated converter transformer is improved.

【0070】

Further, since the gap is not provided, the generation of the fringe magnetic flux in the vicinity of the gap is eliminated, and the heat generation in the primary winding and the secondary winding can be significantly suppressed and the power loss can be reduced. Further, in particular, depending on the structure of the insulated converter transformer in the present invention, the amount of current flowing through the primary winding and the secondary winding can be made smaller than before, so that the heat generation can be suppressed and the power loss can be reduced as described above. It becomes possible to further promote the reduction of.

【0071】

Further, in the structure of the isolated converter transformer PIT in the present invention, since the leakage inductance is also reduced, ZVS operation is guaranteed even under the condition of low AC input voltage under load, and the reliability as a power supply circuit is improved.

[Simple explanation of drawings]

[Figure 1]

It is a circuit diagram which shows the structural example of the switching power supply circuit of 1st Embodiment of this invention.

[Figure 2]

It is sectional drawing which shows the structure of the insulation converter transformer provided in the switching power supply circuit of embodiment.

[Fig. 3]

It is a waveform diagram which shows the operation of the main part in the switching power supply circuit of embodiment.

[Fig. 4]

It is a waveform diagram for comparing the ZVS operation with respect to the embodiment and the switching power supply circuit of the prior art.

[Fig. 5]

It is a circuit diagram which shows the structural example of the switching power supply circuit of the 2nd Embodiment of this invention.

[Fig. 6]

It is a circuit diagram which shows the structural example of the switching power supply circuit of the 3rd Embodiment of this invention.

[Fig. 7]

It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art.

[Fig. 8]

It is sectional drawing which shows the structure of the insulation converter transformer provided in the switching power supply circuit of the prior art.

[Fig. 9]

It is an equivalent circuit diagram which shows each operation in the insulation converter transformer when the mutual inductance is a polar mode and a polar mode.

[Explanation of symbols]

1 Control circuit, 2 Switching drive, 3 Oscillator, 4 Drive circuit, 20 Active clamp circuit, Q1 Main switching element, Cr Primary side parallel resonance capacitor, DD1, DD2 Clamp diode, C2 Secondary side parallel resonance capacitor, PIT insulation Converter transformer, C3 clamp capacitor, Q2 auxiliary switching element