Method and encoder for channel coding a radio signal for a radio channel

Method and Encoder for Channel Coding a Radio Signal for a Radio Channel

DISCLAIMER

It is to be noted that only subject matter embraced in the scope of the claims appended hereto, whether in the manner defined in the claims or in a manner similar thereto and involving the main features as defined in the claims, is intended to be included in the scope of the present invention, while subject matter described and exemplified to provide background and better understanding of the invention, is not intended for inclusion as part of the present invention.

TECHNICAL FIELD

The present invention relates to digital radio communication systems. More particularly, and not by way of limitation, the present invention is directed to a transmitter and method for transmitting a sequence of transmitted symbols in a digital communication system utilizing soft pilot symbols.

BACKGROUND

In digital communication systems, the receiver must estimate some parameters in order to correctly demodulate the transmitted data. The receiver may also need to estimate a measure of signal quality to feed back to the transmitter. The estimation of parameters/signal quality generally falls into three categories:

(1) Blind estimation. Generally this approach relies on some signal or channel property/characteristic that is known a priori or learned in a slow manner (for example, second-order statistics). The biggest problem with blind estimation is performance. Blind estimation generally underperforms other approaches by a significant margin. Also, blind estimation algorithms may be more complex.

(2) Pilot-aided. This approach includes known (i.e., pilot) symbols in the transmitted signal. Pilot symbols can be embedded in the data sequence (for example, the midamble of GSM) or allocated a separate resource such as the pilot code in WCDMA, so long as the pilot symbols experience the same effective fading channel as the data. The pilot-aided approach generally offers the best performance. However, pilot symbols consume resources that might otherwise be devoted to transmitting useful data. Typically there is a tradeoff between having sufficient pilots for good estimation and maximizing data throughput.

(3) Data-aided. This approach uses demodulated data symbols as “extra” pilot symbols. Generally this approach is used in conjunction with either blind estimation or the pilot-aided approach. There are two problems associated with the data-aided approach. First, blind estimation or pilot-aided estimation (or both) is typically required as a first receiver step. Therefore, data-aided approaches require extra receiver complexity. Second, data-aided approaches can degrade receiver performance due to the effect of errors in demodulating data. In data-aided approaches, the demodulated data symbols are assumed to be correct and are used as additional pilot symbols. However, if the data symbols are incorrect, the parameter/signal quality estimation algorithms can produce incorrect results. The effects of incorrect symbol decision(s) can persist for more than one estimation interval, so data-aided approaches may need special mechanisms to avoid the effect of error propagation.

The data-aided approach has been utilized in a number of existing communication systems. For example, in Wideband Code Division Multiple Access (WCDMA) systems, the control channel on the uplink is demodulated/decoded, and the symbol decisions are used as effective pilots. This has also been proposed for the WCDMA control channel on the downlink. In the Digital Advanced Mobile Phone System (D-AMPS), the channel is first estimated over a synchronization word and then tracked over data during equalization. In the equalizer, early temporary unreliable decisions are fed to the tracker, and delayed better decisions are fed to the decoder. Also in D-AMPS and GSM, multi-pass (turbo) demodulation/decoding uses decoded/reencoded symbols as effective pilots in a second pass.

Some of these issues are addressed in the following prior art references:

EBERLE W ET AL: “80-Mb/s QPSK and 72-Mb/s 64-QAM Flexible and Scalable Digital OFDM Transceiver ASICs for Wireless Local Area Networks in the 5GHz Band” IEEE JOURNAL OF SOLID-STATE CIRCUITS, vol. 36, no. 11, 1

-3November 2001 (2001-11-01), XP011061618 IEEE SERVICE CENTER,

PISCATAWAY, NJ, US ISSN: 0018-9200

VALENTI M C ET AL: “A bandwidth efficient pilot symbol technique for coherent detection of turbo codes over fading channels” IEEE MILITARY COMMUNICATIONS CONFERENCE PROCEEDINGS, vol. 1, 31 October 1999 (1999-10-31), pages 81-85, XP010369653 PISCATAWAY, NJ, USA, IEEE, US DOI: 10.1109/MILCOM.1999.822647 ISBN: 978-0-7803-5538-5

US 2002/0186778 discloses a method and apparatus to adaptively puncture bits within QAM modulated data symbols transmitted in a communication system in order to effect a signaling channel. Inherent characteristics of a particular mapping scheme are used for the QAM constellation to selectively puncture particular bits within a data symbol with signaling information and predetermined binary values to selectively increase the log-likelihood ratio gains of those particular bits punctured with the signaling information obtained using the log-likelihood ratios.

US Patent No. 4,535,352 discloses a technique for generating a Semi-Compatible High Definition Television (SC-HDTV) signal which can be sent over two present-day Cable Television (CATV) channels. An HDTV signal is formed comprising a line scan rate which is a multiple of the line scan rate of a current conventional TV signal. The SC-HDTV signal is formed therefrom by transmitting a first HDTV line as is and a second HDTV line as a field differential signal transmitted via QAM on the IF or RF carrier of the first line signal at a line scan rate of a current conventional TV signal. Arrangements are also disclosed for deconverting the SC-HDTV signal back into either a HDTV or conventional TV signal.

US Patent No. 6,904,097 discloses a method and apparatus to adaptively puncture bits within QAM modulated data symbols transmitted in a communication system in order to effect a signaling channel. The method and apparatus utilize inherent characteristics of a particular mapping scheme for the QAM constellation to selectively puncture particular bits within a data symbol with signaling information and predetermined binary values to selectively increase the log-likelihood ratio gains of those particular bits punctured with the signaling information. The log-likelihood ratios are used to obtain the signaling information.

US Patent No. 7,000,173 discloses a turbo code based incremental redundancy includes a first step of puncturing a data stream for a first transmission to provide a set of first unpunctured trellis sections. A next step includes puncturing a data stream for a second transmission to provide a set of second unpunctured trellis sections. A next step includes incremental redundancy combining the first and second transmissions of the trellises to provide non-adjacent first and second unpunctured trellis sections.

-4US Patent No. 7,079,585 discloses a hierarchical QAM system allows the transmission of different sources by embedding the relative constellation points. At the transmitter the constellations generated by the different sources are remapped by a grey coder and then combined. At the receiver, the different source data are decoded by using a corresponding grey code demapper.

US Patent No. 7,835,321 discloses a method of allocating channels in a user equipment. The method includes determining whether a High Speed Downlink Shared Channel (HS-DSCH) is configured for the user equipment (UE) and determining a number of codes used by the DPCH and the E-DCH. The method further includes allocating the DPCH and the E-DCH channels to an I branch or a Q branch based on the number of codes used by the DPCH and the E-DCH and the HS-DSCH configuration.

US Patent No. 7,907,671 discloses a communication unit having a transmitter and a processor. The processor receives information representative of a configuration of physical channels and determines a scaling factor responsive to the information and facilitates scaling a signal provided by a combination of the channels utilizing the scaling factor. The scaled signal is provided to the transmitter.

US 2002/0101915 discloses a system and method of encoding a sequence of information bits comprising the steps of dividing the information bits into encoding bits and parallel bits; encoding the encoding bits to produce encoded bits; mapping the encoded bits and the parallel bits into first and second pulse amplitude modulation (PAM) signals; and generating a quadrature amplitude modulation (QAM) signal from these first and second PAM signals.

US 2005/0143004 disclose a method for transmitting data having primary and secondary data. A first transmit signal is transmitted from a first communication satellite wherein the first transmit signal is produced when the transmitter modulates the primary and secondary data on a first carrier wave associated with the first communication satellite, with secondary data having a first polarity. A second transmit signal is transmitted from a second communication satellite wherein said second transmit signal is produced when the transmitter modulates the primary and secondary data on a second carrier wave associated with the second communication satellite with secondary data having a second polarity opposite the first polarity.

US 2006/0023680 discloses a method and system for transmitting reference signals in a BWA communication system using a first transmit antenna, a second transmit antenna, and a plurality of sub-carriers. Each of the sub-carriers occupies a sub-frequency domain. Reference signals for identifying a first transmit antenna are transmitted through a second predetermined number of sub-carriers in a first mini sub-channel transmitted through the first transmit antenna and occupied by a predetermined time domain and sub-frequency

-5domains of a first predetermined number of sub-carriers, and reference signals for identifying a second transmit antenna are transmitted through the second predetermined number of the sub-carriers in a second mini sub-channel transmitted through the second transmit antenna and occupied by the time domain and the sub-frequency domains of the first predetermined number of sub-carriers.

US 20070071119 discloses a multicarrier transmitter performs bit puncturing for pilot subcarriers in two operations. The first operation may uniformly removes bits from an encoded bit stream to achieve a predetermined code rate reserving space for pilot subcarriers of a predetermined number of antennas. The second operation may remove bits from each spatial data stream after interleaving in a manner to achieve approximately equal spacing between the punctured bits before interleaving. In this way, a conventional interleaver may be used. The second operation reserves space for pilot subcarriers when a greater number of antennas than the predetermined number are used.

US 20070253324 discloses a method and apparatus are provided for reducing the number of pilot symbols within a MIMO-OFDM communication system, and for improving channel estimation within such a system. For each transmitting antenna in an OFDM transmitter, pilot symbols are encoded so as to be unique to the transmitting antenna. The encoded pilot symbols are then inserted into an OFDM frame to form a diamond lattice, the diamond lattices for the different transmitting antennae using the same frequencies but being offset from each other by a single symbol in the time domain. At the OFDM receiver, a channel response is estimated for a symbol central to each diamond of the diamond lattice using a two-dimensional interpolation. The estimated channel responses are smoothed in the frequency domain. The channel responses of remaining symbols are then estimated by interpolation in the frequency domain.

US 2007/0274407 discloses a Viterbi decoding system that interprets bits in received QAM constellations as many-valued parameters rather than binary valued parameters. Rather than applying a hard 0-1 function to the QAM data, the system uses a non-stepped linear or curved transfer function to assign values to the bits. In another aspect, a system differentiates between data bits based on their estimated reliability, giving more emphasis to decoding reliable bits than unreliable bits using any of a variety of techniques.

US 2007/0291853 discloses apparatuses and methods for transmitting/receiving uncompressed data in which, when the uncompressed data is transmitted or received over a wireless network, different code rates are applied to the data according to the significance of bits or bit groups included in the data.

US 2008/0151989 discloses replacement of data symbols at determined positions of a transmitted data stream by quasi-pilot symbols earring data modulated onto the carrier

-6with a different modulation scheme than the original symbols. The modulation scheme for the quasi-pilot symbols has a lower amplitude and/or phase ambiguity than the modulation scheme for the original data symbol.

US 2009/0132889 discloses apparatus and methods for storing data in a nonvolatile solid state memory device according to a rate-compatible code, such as a ratecompatible convolutional code (RPCC). An example of such a memory device is a flash memory device. Data can initially be block encoded for error correction and detection. The block-coded data can be further convolutionally encoded. Convolutional-coded data can be punctured and stored in the memory device. The puncturing decreases the amount of memory used to store the data. Depending on conditions, the amount of puncturing can vary from no puncturing to a relatively high amount of puncturing to vary the amount of additional error correction provided and memory used. The punctured data can be decoded when data is to be read from the memory device.

SUMMARY

According to the invention there are provided a method and encoder for channel coding a radio signal for a radio channel having the features of the respective independent claims.

The present invention overcomes the disadvantages of the prior art by transmitting some symbols with higher reliability than others. These so-called soft pilots are demodulated first and then used as known symbols for use in channel estimation and demodulation of higher-order modulation symbols (amplitude reference). These soft pilot symbols are more robust than the surrounding symbols, thereby enabling reliable decision-directed parameter estimation. Additionally, inserting a “constant envelope” modulation symbol among higher order modulation symbols is particularly helpful in establishing the amplitude reference essential in demodulating the higher order modulation symbols.

In one embodiment, the soft pilot symbols are modulated with a simpler, lower order modulation (for example, BPSK or QPSK) compared to the rest of the symbol sequence, which is likely a higher order modulation (for example, 16 Quadrature Amplitude Modulation (16QAM) or 64QAM). By using these soft pilots, the symbol can still carry some data, contrasted to a fixed pilot symbol, which allows no data throughput for the symbol. These specified soft symbol locations (time/frequency/code) and the modulation type(s) are shared with the receiver. The receiver may know the information a priori or through signaling.

-7Soft pilots provide an alternative to explicit data pilots for future releases of WCDMA. With soft pilot symbols, explicit pilot symbols are not necessary. With knowledge of the modulation type and the location of the soft pilots in time, frequency, and code, the receiver can maximize performance. This allows for better data rates than would otherwise be possible with explicit pilot symbols.

In another embodiment of the invention, the soft pilot symbols are generated by low-level puncturing of channel coded bits. The method includes inserting a set of soft pilot symbols by low-level puncturing of channel coded bits and replacing with known bit patterns, modulating the sequence, and transmitting the radio signal.

In a specific embodiment related to the High Speed Downlink Shared Channel (HS-DSCH), the soft pilot symbols are generated during the channel coding chain by low-level puncturing of channel coded bits after rearranging the modulation constellation and before mapping to a physical channel. In a specific embodiment related to the Enhanced Dedicated Channel (E-DCH), the soft pilot symbols are generated during the channel coding chain by low-level puncturing of channel coded bits after interleaving on the E-DCH and before mapping to a physical channel. With such a mechanism, the use of soft pilots requires no changes to the specification and implementation of the critical channel coding and rate matching procedures. This enhances compatibility with legacy equipment and allows reuse of existing transceiver implementations.

In another embodiment, the present invention is directed to a transmitter for transmitting a radio signal that includes a sequence of transmitted symbols. The transmitter includes means for inserting a set of soft pilot symbols by low-level puncturing of channel coded bits and replacing the punctured bits with known bit patterns; and means for modulating the sequence and transmitting the radio signal.

In another embodiment, the present invention is directed to a channel coder for channel coding a radio signal for a radio channel. The channel coder includes means for inserting soft pilot symbols by low-level puncturing of channel coded bits; and means for replacing the punctured channel coded bits with known bit patterns after channel interleaving. In a specific embodiment, the radio channel is an HS-DSCH. In another specific embodiment, the radio channel is an E-DCH.

According to another embodiment of the invention, the locations of the soft pilots in terms of time and code (or frequency) are designed to accommodate timevarying channel responses and to minimize undesirable impact on code performance and peak-to-average ratios.

-8BRIEF DESCRIPTION OF THE DRAWINGS

In the following section, the invention will be described with reference to exemplary embodiments illustrated in the figures, in which:

FIG. 1 is a flow chart illustrating the steps of an exemplary embodiment of the method of the present invention;

FIG. 2 shows the data bit mapping to points in the constellation for 16QAM in one exemplary embodiment of the present invention;

FIG. 3 shows the data bit mapping to points in the constellation for 16QAM in another exemplary embodiment of the present invention;

FIG. 4 (Prior Art) illustrates the existing channel coding chain for the HS-DSCH;

FIG. 5 illustrates the channel coding chain for the HS-DSCH in an exemplary embodiment of the present invention;

FIG. 6 is a flow diagram illustrating an overview of a soft pilot generation process in an exemplary embodiment of the present invention;

FIG. 7 is a flow diagram illustrating a soft pilot generation process for the HSDSCH in an exemplary embodiment of the present invention;

FIG. 8 is a flow diagram illustrating a soft pilot generation process for the EDCH in an exemplary embodiment of the present invention;

FIG. 9 is a functional block diagram of an exemplary embodiment of an interleaver structure for the E-DCH;

FIG. 10 illustrates a first exemplary embodiment of soft pilot symbol location;

FIG. 11 illustrates a second exemplary embodiment of soft pilot symbol location;

FIG. 12 is a functional block diagram of an exemplary embodiment of a twopass G-Rake receiver of the present invention; and

FIG. 13 is a flow chart illustrating an exemplary embodiment of a processing method performed by the two-pass G-Rake receiver of the present invention.

DETAILED DESCRIPTION

For high data rate communications, higher order modulations such as 16QAM and 64QAM are utilized to increase spectral efficiency. According to a first embodiment of the present invention, the transmitter designates certain symbols in the data sequence as so-called “soft pilot” symbols by using a specific alternative modulation for these symbols. The specific modulation order and the location of these symbols (in terms of time, code, and/or frequency) is known by or signaled to the receiver. The

-9receiver utilizes the soft pilot symbols to obtain an initial estimation of signal parameters such as the channel taps and the correlation matrix. After a first demodulation, decided symbols may be utilized as effective pilots in a second pass of parameter estimation. By limiting the decided soft pilot symbols to a lower modulation than the remaining symbols in the sequence, their decisions are reliable enough to make them useful pilots. The soft pilots are different than traditional fixed pilots in that some data throughput is carried by these soft pilot symbols. Thus, replacing traditional fixed pilots with soft pilots improves data throughput.

FIG. 1 is a flow chart illustrating the steps of an exemplary embodiment of the method of the present invention. At step 11, a radio signal is transmitted with some symbols having higher reliability (for example, with a lower order modulation) than other transmitted symbols. At step 12, the radio signal is received and the higher reliability symbols are demodulated first to form soft pilot symbols. At step 13, the soft pilots are utilized as known symbols for channel estimation and demodulation of the higher order modulation symbols. At step 14, data is extracted from both the soft pilot symbols and the higher order modulation symbols.

An exemplary embodiment of the present invention specifies the modulation type and the location (time/frequency/code) of the soft pilot symbols within the data sequence. According to one embodiment of the invention, the constellation points of the soft pilots are taken as a subset of the higher order modulation constellation for the data transmission, such as 16QAM or 64QAM. The transmitter may utilize a specified lower order modulation for the pilot symbols such as Binary Phase Shift Keying (BPSK) or Quadrature Phase Shift Keying (QPSK). For the rest of the symbol sequence, the transmitter may utilize a higher order modulation (for example, 16QAM or 64QAM). These specified soft symbol locations and the modulation type(s) are known by the receiver. The receiver may know the information a priori or through signaling.

Thus the present invention transmits lower order modulation symbols inserted among higher order modulation symbols, and the receiver performs associated actions to exploit the lower order modulation symbols as effective pilots. A symbol can carry a range of number of bits m: m=Q bit corresponds to a pure pilot; m=1 bit corresponds to BPSK; m=2 bits corresponds to QPSK; and so on, up to the maximum number M (=6 for 64QAM). If it is assumed for simplicity that all symbols have the same energy, then the bit energy and the bit reliability decrease with m. Thus, the symbols can be used as pilots of various levels of reliability, and the receiver can perform parameter estimations in multiple passes.

-10FIG. 2 shows the data bit mapping to points in the constellation for 16QAM in one exemplary embodiment of the present invention. The four corner points of the 16QAM constellation (shown in the figure as starred points) are taken as the constellation for the soft pilots. Two features of this embodiment can be readily recognized. First, the soft pilot constellation is equivalent to a scaled QPSK constellation. It thus offers the benefits of constant envelope and higher average power. Second, the soft pilot constellation points can be easily addressed within the higher-order constellation by keeping a subset of the bit labels fixed. In the example shown in FIG. 2, the soft pilot constellation points are those with the last two bit labels fixed at “11”.

As noted, the use of soft pilot symbols causes the transmitted 16QAM or 64QAM symbols to have a higher average power. For example, if one in ten symbols for one channelization code is a soft pilot symbol, the average power is increased by 0.15 dB for 16QAM and by 0.54 dB for 64QAM. Alternatively, if there are fifteen channelization codes, and one in ten symbols for one of the fifteen channelization codes is a soft pilot symbol, the average power is increased by only 0.02 dB for 16QAM and by 0.04 dB for 64QAM. In practice, the transmitted power may have to be reduced by these amounts when utilizing soft pilots. It has been seen, however, that the net system performance is improved by the use of soft pilots.

FIG. 3 shows the data bit mapping to points in the constellation for 16QAM in another exemplary embodiment of the present invention. In this embodiment, the soft pilot constellation size is enlarged to allow higher capacity for carrying data. However, the soft pilot constellation provides a constant quadrature amplitude feature which may be utilized to derive an amplitude reference. The soft pilot constellation points are addressed within the higher-order constellation by fixing the last bit label to “1It is clear to those skilled in the art that an alternative soft pilot constellation may be specified by fixing the third bit label to “1”, providing constant in-phase amplitude.

The introduction of soft pilots reduces the number of channel coded bits that can be carried by the transmission signal. The reduction in channel coded bits can be implemented by two different approaches. In a first approach utilizing high-level puncturing, the reduction in channel coded bits is explicitly handled by the entire channel coding chain. This approach may be adopted when designing a new communications system or protocol. However, backward compatibility is an important factor to consider when introducing soft pilot symbols into existing systems. For backward compatibility, it may be preferred to adopt a second approach utilizing low

-11level puncturing such that the majority of the channel coding chain is affected by the new feature. In the following, the HSPA examples are utilized to illustrate the two approaches in detail.

Soft Pilot Generation in HSPA:

FIG. 4 illustrates the existing channel coding chain for the High Speed Downlink Shared Channel (HS-DSCH). In a first high-level puncturing approach for implementing the reduction in channel coded bits, the behavior of the overall channel coding chain is changed similarly to the one for the HS-DSCH. The impact, however, is not simply a different number of coded bits to be output by the “physical-layer HARQ functionality”, but rather a significant redesign and redefinition of several inter-connect and intricate physical-layer procedures in “physical-layer HARQ functionality”, “physical channel segmentation”, “HS-DSCH interleaving”, and “Constellation rearrangement”. Such significant redesign of the critical channel coding chain will render most of the existing implementation obsolete and will be difficult to co-exist with new and legacy equipment in a network.

FIG. 5 illustrates the channel coding chain for the HS-DSCH in an exemplary embodiment of the present invention. In a second, preferred approach for implementing the reduction in channel coded bits, the soft pilot symbols are preferably generated by low-level puncturing of channel coded bits before the “physical channel mapping” stages of the channel coding chain. The preferred embodiment thus makes the presence of soft pilot symbols transparent to the “physical-layer HARQ functionality”, “physical channel segmentation”, “HS-DSCH interleaving”, and “constellation rearrangement” stages.

FIG. 6 is a flow diagram illustrating an overview of a soft pilot generation process in an exemplary embodiment of the present invention. In HSDPA, the bit collection procedure in physical-layer HARQ functionality and the HS-DSCH channel interleaving are designed to map systematic turbo-coded bits, if present, to the first bit labels of the 16QAM or 64QAM as much as possible. The purpose of this design is to ensure the important systematic turbo-coded bits are transmitted over the channel with higher reliability. As shown in FIG. 6, this is accomplished in the channel interleaver by utilizing pair-by-pair bit multiplexing and independent rectangular interleavers. When the data modulation is based on QPSK, only the first rectangular interleaver branch is active. When the data modulation is based on 16QAM, the first and the second rectangular interleaver branches are active. All three branches are active when the

-12data is carried by 64QAM. Coupled with the constellation labeling specified in 3GPP, “Technical Specification Group Radio Access Network; Spreading and Modulation (FDD),” TS 25.213 v8, the bits in the first branch are transmitted over the channel with highest reliability. The bits in the third branch are transmitted with lowest reliability. Hence, in initial transmissions, the systematic bits are normally transmitted through the first branch as much as possible. For initial transmissions, the HARQ parameters are generally set such that the “constellation rearrangement” is effectively by-passed. It should be obvious to those skilled in the art that soft pilot symbols can be inserted right after the channel interleaving. For retransmissions, HARQ parameters can be used to instruct the “constellation rearrangement” to effectively retransmit channel coded bits with different reliability. Soft pilot symbols may be inserted into the signal after the “constellation rearrangement” procedure.

FIG. 7 is a flow diagram illustrating a soft pilot generation process for the HSDSCH in an exemplary embodiment of the present invention. The coded bit inputs are denoted by r_P:k and the outputs are denoted by r’_P:k. Normally, the input bits are passed to the output without modification: γ'_ρΛ = r_P:k. If a scaled QPSK soft pilot symbol (such as that shown in FIG. 2) is inserted to replace a 16QAM data symbol, then r’_P:k = r_p,_k, r’_Ptk+1= r_p,_k+i, r’_p,_k+2 = 1, and r’_p,_k+3 = 1. If a scaled QPSK soft pilot symbol is inserted to replace a 64QAM data symbol, then r’_p,_k = r_P:k, r’_P:k+1 = r_p,_k+1, r_pk+2 = 1, r’_p,_k+3 = 1, r’_p,_k+4 = 1, and t p,k+5 ⁼ 1

If a soft pilot symbol with constant quadrature amplitude (such as that shown in FIG. 3) is inserted to replace a 16QAM data symbol, then r’_P:k = r_p,_k, r’_P:k+1 = r_Ptk+1, r’_pk+2 = r_P:k+2, and r’_P:k+3 = 1. If a soft pilot symbol with constant quadrature amplitude is inserted to replace a 64QAM data symbol, then r’_p,_k = r_P:k, r’_P:k+1 = r_P:k+1, r’_P:k+2 = r_P:k+2, r’_P:k+3 = 1, r’_P,_k+4 = r_P:k+4, and r’_P:k+5 = 1. If a soft pilot symbol with constant in-phase amplitude is inserted to replace a 16QAM data symbol, then r’_P:k = r_P:k, r’_P:k+1 = r_P:k+1, r’_P:k+2 = 1, and r’_pk+3 = r_P:k+3. If a soft pilot symbol with constant in-phase amplitude is inserted to replace a 64QAM data symbol, then r’_p,_k = r_P:k, r’_P:k+1 = r_P:k+1, r’_P:k+2 = 1, r’_P:k+3 = r_P:k+3, r’_P:k+4 = 1, and r_p,_k+5 = r_pk+3.

Soft Pilot Generation for the Enhanced Dedicated Channel (E-DCH):

FIG. 8 is a flow diagram illustrating a soft pilot generation process for the EDCH in an exemplary embodiment of the present invention. To accomplish reliability identification similar to that in HS-DSCH, the bit collection procedure in physical-layer HARQ functionality and the channel interleaving are designed to map systematic turbo

-13coded bits, if present, to the first bit labels of the 4PAM as much as possible. According to the preferred embodiment, the soft pilot symbols are generated after the E-DCH channel interleaving.

FIG. 9 is a functional block diagram of an exemplary embodiment of an interleaver structure for the E-DCH. The channel interleaving is facilitated by two rectangular interleaver branches when the data is carried by 4PAM. The coded bit inputs to the “soft pilot generation” are denoted by ν_ρΛ and the outputs are denoted by v’_P:k. Normally, the input bits are passed to the output without modification: v’_pk = v_P:k. If a scaled BPSK soft pilot symbol is inserted to replace a 4PAM data symbol, then v’_P:k =

V p,k+1 ⁼ 1

According to the preferred embodiment, the soft pilot symbols are generated by puncturing channel coded bits at fixed locations (in terms of time and code/frequency). On the receiver side, the soft values corresponding to the punctured bits are set to zero. With this, the use of the soft pilot symbols introduces no changes to the core ratedematching and channel decoder implementation.

Note also that, according to this embodiment, the soft pilot symbols are generated by puncturing channel coded bits that are mapped to the least reliable bit labels. Since the soft values corresponding to these low-reliability bits are normally very small, setting them to zero introduces negligible impact to the overall channel coding performance.

Location of Soft Pilot Symbols:

Soft pilot symbols may be imbedded on the same code, on a single separate code, on different antennas in Multiple-Input-Multiple-Output (ΜΙΜΟ) systems, and the like. The placement may be coordinated so that the soft pilot symbols either coincide or do not coincide on different codes and/or antennas.

The soft pilot symbols can be inserted into the signal in several practical ways:

1. HSPA - one code assigned to the HSPA user utilizes soft pilot symbols while other codes assigned to the same user utilize a higher order modulation.

2. HSPA - certain data symbols within each code assigned to the HSPA user are soft pilot symbols while the remaining symbols in the codes are conventional data symbols. For example, symbols 0 through N-1 on code A, N through 2N-1, on code B, and so on may be soft pilot symbols.

3. HSPA - symbols Λ/ through 2N are soft pilot symbols on all codes assigned to the HSPA user while the remaining symbols in the codes assigned to the same user are conventional data symbols.

4. Long Term Evolution (LTE) - replace demodulation pilots with soft pilot symbols for some (or all) of the embedded demodulation pilots.

The following embodiments are designed with further consideration of (a) supporting time-varying channels, (b) minimizing coding performance impact, and (c) reducing impact on peak-to-average ratio (PAR).

FIG. 10 illustrates a first exemplary embodiment of soft pilot symbol location. The soft pilot symbols are spread out in time to provide a more reliable reference for time-varying channels. The exact locations of the symbols may be specified by periodic patterns. To allow for averaging for estimation noise reduction, the soft pilot symbols may be present in more than one code at the same spread-out locations. In contrast to concentrating the soft pilot symbols into only one (or very few codes), the spread-out pattern across codes minimizes the impact on overall channel decoding performance.

FIG. 11 illustrates a second exemplary embodiment of soft pilot symbol location. The embodiment previously illustrated in FIG. 10 is suitable only if the soft pilot symbols do not contribute to substantial increase in PAR. If the PAR increase is of concern, the embodiment of Figure 11 can be adopted. The soft pilot symbol locations between different codes are offset to reduce PAR increase.

The use of soft pilot symbols provides several benefits. First, the soft pilot symbols are more robust than the surrounding symbols, thereby providing reliable decision-directed parameter estimation. Second, the soft pilot symbols may still carry some data, contrasted to fixed pilot symbols, which allow no data throughput for the symbol. Third, by making the soft pilot symbols “constant envelope” modulation symbols inserted among higher order modulation symbols, the soft pilot symbols become particularly helpful in establishing the amplitude reference essential for demodulating the higher order modulation symbols.

The use of soft pilot symbols is applicable to any wired or wireless communication system. Soft pilots provide higher data throughput than traditional pilotaided schemes, and do not sacrifice performance as most blind estimation schemes do. The soft pilot approach requires that the receiver use a data-aided approach. However, as opposed to traditional data-aided approaches, the present invention specifies the modulation and location (in time/code/frequency) of the soft pilot symbols so that the receiver will know that there are certain high-quality symbols that can be

-15used in a data-aided approach. Receiver estimation algorithms based on such symbols are less error-prone and provide consistently good parameter and/or signal quality estimates.

An HSPA receiver that can utilize such soft pilots is fully described below in an exemplary embodiment consisting of a data-aided Generalized Rake (G-Rake) receiver. By way of background, the G-Rake receiver receives and processes WCDMA signals experiencing interference in dispersive channels. This interference is composed of self-interference (intersymbol interference), multiple access interference (interference due to non-zero code cross correlation), and other cell (downlink) or other user (uplink) interference. This interference must be suppressed in order to achieve good HSDPA throughput. In addition, the enhanced throughput requirements set by 3GPP for type 2 (single antenna terminal) and type 3 (dual antenna terminal) receivers cannot be met without interference suppression.

Linear methods for suppressing interference generally fall into the categories of chip level or symbol level equalization. Symbol level equalization follows the traditional Rake architecture where the received chip-level data is despread at multiple delays, and then the multiple images are combined. Chip level equalization reverses the order of these operations; the received chip data is first combined using a linear filter and then despread at a single delay. These techniques are generally equivalent from a performance perspective.

FIG. 12 is a functional block diagram of a G-Rake receiver 20 which may be modified to utilize the present invention. The receiver may be implemented, for example, in a mobile terminal or other wireless communication device. Spreadspectrum signals are transmitted through a radio channel and are received at one or more antennas of the receiver. A radio processor (not shown) generates a series of digitized baseband signal samples 21 from the received signal and inputs them to the G-RAKE receiver. In turn, the G-Rake receiver 20 demodulates the received signal samples to produce soft values or bit estimates 22. These estimates are provided to one or more additional processing circuits (not shown) for further processing, such as forward-error-correction (FEC) decoding and conversion into speech, text, or graphical images, and the like. Those skilled in the art will recognize that the particular information type(s) carried by the received signal and the particular processing steps applied by the receiver 20 are a function of its intended use and type.

-16A complete description of a G-Rake receiver suitable for use with the soft pilot symbols of the present invention is provided in co-owned U.S. Patent Publication No. 2005/0201447.

Turning first to symbol level equalization, the G-Rake combining weights perform the coherent combining as well as interference suppression. The combining weights are given by:

w = R;¹h, (1) where R„ is the impairment covariance matrix and h is a vector of net channel coefficients. It should be noted that the term “impairment” includes both interference and noise while the term “net channel coefficient” refers to a channel coefficient that includes the effects of the transmit and receive filters as well as the fading channel.

There are two general methods for implementing a G-Rake receiver. These methods are generally known as nonparametric and parametric. The nomenclature here focuses on the approach taken to obtain the impairment covariance matrix. Nonparametric method(s) are blind, and estimate R_M directly from observed data. The parametric method assumes an underlying model, and computes R from model parameters. Examples of both methods are provided below.

There are two ways that one can obtain a nonparametric estimate of the impairment covariance matrix. The first approach uses the pilot channel to estimate the slot-based quantities:

ι·ΥΣν(^· \ ^k N_{p i} (2) =T—7 Σ(^χΧ^α·)^λ* -h)(x_p(k>* -h^ p ¹ k=o

Using these quantities, the impairment covariance matrix can be obtained from:

= · Ο)

Another approach for generating a nonparametric estimate of the impairment covariance matrix involves the use of unoccupied traffic codes as described in coowned U.S. Patent No. 8,098,715. The despread values for these codes contain

-17impairment samples only. These impairment samples can be used to directly estimate R as follows:

7V_c-17V_r-l k = V7F Σ (4)

C¹ T q=0 k=o

Here, x,_raffic(k) is a despread vector of traffic symbols for the qth code during the kth symbol interval, N_T is the number of symbols per code, and N_c is the number of codes.

The parametric approach for generating the impairment covariance matrix depends upon a model for the interference as described in co-owned U.S. Publication No. 2005/0201447. This model depends upon the radio channel(s) between the UE and the modeled base station(s). Assuming a single serving base station and J interfering base stations, the model for the impairment covariance matrix is given by:

R„ = E_c(0)Rrfe,)+ iXOW' (i_;)+(5)

7=1 where:

L-l L-loo ^dl ) = ΣΣ 8j Σ^ΛΡ (^J1 ^mTc ~ ^Tk (t))^Rp (^d2 - ^mTc - ^Tk («)) /=0 n=0m=-oo

L-l L-loo = ΣΣ^(^)Σ^ι ~^mTc-T^K{^d2-mT_c-T_k(nf) /=0 n=0m=-oo

R-n vA ’ ) ^— Rp vA ~ ^2/ (6)

Here, E_c(j) is the total chip energy for base station g_y is a vector of radio channel (medium) coefficients for the channel between the UE and the jth base station, R_p(&) represents the convolution of the transmit and receive pulse shape filters evaluated at θ, τ_; is a vector of L channel delays corresponding to the channel between the UE and the jth base station, T_c is the chip time, and d_k is the delay of the kth finger employed by the UE.

-18Chip equalization is discussed in G. Klutz et al., “Sparse Chip Equalizer for DSCDMA Downlink Receivers”, IEEE Communication Letters, vol. 9, no. 1, pp. 10-12, 2005. According to Klutz, the received signal at the chip level is given by:

r = He + v. (7)

Here, r is a N + L-1 block of received chips, H is the (n + L-V)xN) sized Toeplitz convolution matrix whose columns are time shifted versions of the channel impulse response h with delay spread L (chip or sub-chip spaced version of the net channel coefficients), v represents white Gaussian noise due to neighboring base stations and thermal noise, and c is the transmitted chip sequence. The chip equalizer filter f that suppresses the interference in equation (7) is the solution to:

f = Ab, (8) where

A = A'{XX} b = £jx“C“p)

X = CR

C_p = NxS sized pilot scrambling and spreading matrix p = pilot chip sequence

Note that it is assumed that there are S pilot symbols per data block and that the columns of matrix R are time-shifted versions of the chip level received signal r.

Similar to G-Rake, there are several ways to generate the chip equalizer filter. One may use a parametric approach, a nonparametric approach, and a direct adaptation approach. The parametric and nonparametric forms differ (primarily) in how the A matrix is calculated. The nonparametric form uses the received chip data directly to calculate the A matrix via:

A®-------R^hR (9)

N + L - 1

-19In contrast, the parametric form works instead with the channel impulse response and the powers of the serving base station and the white Gaussian noise. The entries of the A matrix for the parametric form can be written as:

j) = I_or^h*(n)i(n + τ_{ -7.) + I_oc$(i - j), (10) n

where r_k is the l·? chip equalizer tap delay, I_or is the serving base station power, and I_oc is the white Gaussian noise power. The direct adaptation approach treats the equalization problem as an adaptive filtering problem. It uses the common pilot signal as a known reference to train the filter taps using any of the common adaptive filter algorithms (LMS, RLS, etc.).

The existing parametric and non-parametric equalization approaches have different strengths and weaknesses. The strengths and weaknesses of the G-Rake parametric/nonparametric approaches are discussed below. It is assumed that these strengths/weaknesses hold for chip equalization as well.

The strength of the parametric approach is that performance (BER, BLER, or throughput) is relatively insensitive to UE speed. The main weakness of the parametric approach is that it relies on channel information developed by the path searcher/delay estimator. If this information is incorrect, then the effective color of the impairment will be mis-modeled leading to performance degradation.

The strength of the nonparametric approach is that it is a blind technique. There is no specific model for interference, so all interference is captured by the estimation approach. This blind approach, however, is also indirectly a weakness. Blind approaches typically need a significant amount of “training” data to perform well. The pilot channel has only 10 symbols per slot, so the pilot-based approach to covariance estimation requires significant smoothing (filtering) to work well. Smoothing limits the effectiveness of the approach to low speed. The unused code approach is highly effective if a set of unused codes can be identified. However, identifying unused codes in the downlink is quite problematic.

It is noted that there is a further weakness inherent in existing equalization techniques. There appears to be an irreducible error floor (i.e., performance ceiling) for practical receiver implementations based on the existing standard. No such phenomenon occurs for a genie receiver. In order to increase the peak data rates offered in practice, a practical receiver must more closely mimic the performance of the

-20genie receiver. It is contemplated that WCDMA release 9 will add more pilot symbols so that nonparametric and/or direct adaptation receivers perform better. The present invention offers an alternative to this approach, which reduces the peak throughput only marginally yet still achieves close to genie receiver performance.

In the two-pass G-Rake receiver of the present invention, the first pass computes a set of “approximate” or “rough” combining weights. These combining weights are used to coherently combine the symbols from one or more traffic codes The combined values are rescaled to some target constellation power and hard symbol decisions are made (i.e., no decoder involvement). The hard symbol decisions are then used as demodulation pilots and the impairment covariance matrix is recalculated nonparametrically using these demodulation pilots. From the recalculated impairment covariance matrix, a set of second pass combining weights are computed. These combining weights are used to coherently combine all the traffic data. When utilizing soft pilot symbols, the receiver operation is the same except that the first pass combining weights are only applied to the soft pilot symbols.

FIG. 13 is a flow chart illustrating an exemplary embodiment of a processing method performed by the two-pass G-Rake receiver of the present invention. At step 31, first pass combining weights are created. At step 32a, despread values for one or more codes are coherently combined using the first pass combining weights. Alternatively, the process may move to step 32b where despread values corresponding to soft pilot symbols are coherently combined to create symbol estimates. At step 33, the symbol estimates are rescaled to some target constellation power. At step 34, hard symbol decisions are made on the rescaled symbol estimates given the constellation used for transmission. At step 35, the hard symbol decisions are utilized to nonparametrically estimate the impairment covariance matrix. At step 36, second pass combining weights are computed utilizing the estimated impairment covariance matrix. At step 37, all traffic data is combined utilizing the second pass combining weights.

This process can be realized in different ways depending upon the scenario. For single stream SISO/SIMO/MIMO scenarios, there are two variants. Similarly for the dual stream ΜΙΜΟ scenario, there are at least two variants. Each variant is described in an alternative embodiment below.

First, a single stream SISO/SIMO symbol level embodiment will be described. For the first pass of demodulation, combining weights are computed via:

Wfirst ⁼ R_M,firsth >

(11)

-21(12) where

<img file="IL234990A_D0001.tif" />

In the above equation, x_p(n,m) represents a vector of despread common pilot values corresponding to the m^th pilot symbol interval during the n^th slot, x^ct(n,k) represents a vector of despread traffic values corresponding to the l·? traffic symbol interval during the n^th slot for the c^th code, N is the number of common pilot symbols per slot, Nc is the number of traffic codes used for estimation, and N_t is the number of data symbols per slot.

It is assumed that a single traffic code is used to create symbol estimates (note: what follows could easily be extended to multiple traffic codes). The first-pass combining weights w_fm._t are applied to traffic code f to create symbol estimates via:

£(Z:) = wX_stx_f^/(z2,Z:). (13)

These symbol estimates are translated to hard symbol decisions by normalizing the energy of the symbol estimates to some target constellation power (e.g. unity) and then selecting the constellation point closest to each symbol estimate. This procedure can be described mathematically as:

= (14) y Δ

Jmin = arg min|z(fc)- V e S

-22where /&lt;(./) is the value of the f^h constellation point taken from the set of constellation points S . The hard decisions are then used to construct a more accurate estimate of the impairment covariance matrix via:

h_f = ^-^x{(n,k)s*(k) k=0

K = -y- (n, k)(x/ (n, k$ (15) k=o

Λ Λ Λ Λ _T1

Ρ — R _ h h⁷⁷ second ^{— U}i^Ut

The more accurate estimate of the impairment covariance matrix is then used to compute the second pass combining weights:

Wsecond second, (Ί θ) and the second pass combining weights are used to coherently combine all the despread traffic data.

Another embodiment is the single stream SISO/SIMO chip level/symbol level embodiment. This embodiment is identical to the symbol level embodiment except that the matrix R„ _frst used to compute first pass combining weights is computed from chip level data. A nonparametric method for realizing this is described above in the prior art section. Specifically, the method of equation (9) is adopted, where the columns of matrix R are time-shifted versions of the chip level received signal r. The setting R_Mfirst = A is made and then the first pass combining weights are calculated. The remainder of the embodiment takes place at the symbol level and is identical to the single stream SISO/SIMO symbol level embodiment.

Another embodiment is the dual stream ΜΙΜΟ symbol level embodiment. This description assumes the D-TxAA ΜΙΜΟ transmission scheme standardized in WCDMA release 7 is utilized, although the invention is general enough to cover other 2x2 ΜΙΜΟ schemes. For the first pass of demodulation, combining weights are computed via:

-23where ht ^W first,1

W_first&gt;2 (18) m=0

N_p-l (19)

In the above equation, x_p(n,m) represents a vector of despread common pilot values corresponding to the m^th pilot symbol interval during the n^th slot, x^ct(n,k) represents a vector of despread traffic values corresponding to the V^th traffic symbol interval during the n^th slot for the c^th code, N is the number of common pilot symbols per slot, Nc is the number of traffic codes used for estimation, N_t is the number of data symbols per slot, s/m) is the m^th pilot symbol transmitted from antenna 1, s2(m) is the m^th pilot symbol transmitted from antenna 2, and b, and b2 are the columns of precoding matrix B used to transmit streams 1 and 2 (i.e. B = [b_x b₂J).

We assume that a single traffic code is used to create symbol estimates (note: what follows could easily be extended to multiple traffic codes). The first-pass combining weights w_firstl and w_flrst2 are applied to traffic code f to create symbol estimates via:

(20)

These symbol estimates are translated to hard symbol decisions by normalizing the energy of the symbol estimates to some target constellation power and then

-24selecting the constellation point closest to each symbol estimate. This procedure can be described mathematically as:

&amp; k=Q y Δι

A™ = ^arg ^min|zi (k) - V x(j) e S ίι(^) = /γ(Χ_η) ^=|ΣΙυ«Γ κ ?ϊ\ λ/ Δ 2

Λ™ = arg min|z₂ (k) - ic(jf V x(j) e 5 ^2 (^)= Jinin) (21) where /&lt;-(./) is the value of the f^h constellation point taken from the set of constellation points S .

The hard decisions are then used to construct a more accurate estimate of the impairment covariance matrix via:

<img file="IL234990A_D0002.tif" />

(22)

The more accurate estimate of the impairment covariance matrix is then used to compute the second pass combining weights ^W second, 1 ^x,l (^1 ) ^W second,2 ⁼ ®·χ,2^ί# (^2 ) (23)

-25and the second pass combining weights are used to coherently combine all the despread traffic data for both streams.

Note: for the first receiver pass, R may be obtained using a parametric GRake formulation. There is a significant advantage to this approach if a QAM modulation is employed.

Another embodiment is the dual stream ΜΙΜΟ chip level/symbol level embodiment. This embodiment is identical to the symbol level embodiment except that the matrix R_x used to compute first pass combining weights

Wfirst.1 w_first&gt;2 =κ;Υ_#(ΐ)₂) is computed from chip level data. A nonparametric method for realizing this is described above. Specifically, the method of equation (9) is adopted, where the columns of matrix R are time-shifted versions of the chip level received signal r. The setting R = A is made, and then the first pass combining weights are calculated. The remainder of the embodiment takes place at the symbol level and is identical to the dual stream ΜΙΜΟ symbol level embodiment.

As will be recognized by those skilled in the art, the innovative concepts described in the present application can be modified and varied over a wide range of applications. Accordingly, the scope of patented subject matter should not be limited to any of the specific exemplary teachings discussed above, but is instead defined by the following claims.