Electrical power multiplication

ES 2 647 014 T3

DESCRIPTION

Multiplication of electrical energy

Cross reference with related cases

The present application is a continuation application in part of US Patent Application Number 11 / 062,035 filed February 18, 2005 entitled ELECTRICAL POWER MULTIPLICATION. This application is also related to the US patent application pending together with the present one entitled Use of Electrical Power Multiplication for Power Smoothing in Power Distribution filed on the same date as the present one and with the application number 11 / 062,179.

Background

Power multiplication may be desirable for many applications that require considerable power resources that cannot be provided economically or physically considering the current state of power technology. For example, attempts have been made to use conventional mechanical flywheel arrangements and capacitive storage for energy storage and energy multiplication. However, such approaches are frequently unsuitable due to decay in power output amplitude and / or frequency as stored energy is extracted or released.

Power multiplication can also be accomplished electrically by using an electromagnetic path pattern to accumulate electrical energy and increase or magnify actual AC power. Such technology has been described by Tischer, FJ, Resonance Properties of Ring Circuits, IEEE Transactions on Microwave Theory and Techniques, vol. MTT-5, 1957, pp. 51-56. The energy multiplier suggested by Tischer allows to obtain a practical energy multiplication of 10 to 500 times the output energy level of a given generator. The multiplication of energy is obtained without appreciable decay in amplitude or frequency.

However, the energy multiplier suggested by Tischer works at relatively short wavelengths where the physical circumference of the device is on the order of an integral number of wavelengths of free space because the electrical length of the electromagnetic path suggested by Tischer it is equal to an integer multiple of the wavelength of a traveling wave multiplied by it. At such short wavelengths, the physical size of the electromagnetic path is such that it can be practically constructed. However, energy multiplication using an approach suggested by Tischer is not practical at lower energy frequencies, such as 60 Hz with relatively long wavelengths since the size of the electromagnetic path would be on the order of several hundred kilometers.

In today's electrical distribution systems, such as the North American power grid, it is often the case where services experience severe mismatches between peak and average load demands. This can result in brownouts and blackouts in the system. In addition, the capacity of the North American power grid is reaching the limit. Consequently, it may be the case that brownouts and blackouts can initiate chain reactions in the power grid that result in a reliable power loss.

US 3049 679 describes a microwave energy multiplier.

US 4686 407 describes a split mode traveling wave ring resonator.

US 3171 086 describes a traveling wave amplifier and an oscillator with tunnel diodes.

Furthermore, another problem faced by energy markets is that intermediate charging points, such as cities, frequently separate electricity generating stations from remote electrical loads. During times of heavy load, the demanded capacity cannot be transmitted from power generating stations to remote loads around intermediate cities.

Brief description of the various views of the drawings

The invention can be understood with reference to the following drawings. Components in drawings are not necessarily to scale. Furthermore, in the drawings, like reference numerals designate corresponding parts throughout the various views.

Fig. 1 is a drawing of an energy multiplier according to the prior art;

Fig. 2 is a drawing of a directional coupler of the power multiplier of Fig. 1;

Fig. 3 is a drawing of an impractical power multiplier with respect to a geographic map illustrating a 2

ES 2 647 014 T3 problem of carrying out energy multiplication using an energy multiplier illustrated in Fig. 1 at energy frequencies of relatively small wavelengths;

Fig. 4A is a block diagram of the power transmission line from a power generator to an electrical load;

Fig. 4B is a diagram of an equivalent impedance per length of the transmission line of Fig. 4;

Fig. 5 is a drawing of alternative transmission lines that could be used as the power transmission line of Fig. 4A and having an equivalent impedance that can be modeled by the scheme of Fig. 4B;

Fig. 6A is a schematic of a T network used in an energy multiplier according to an embodiment of the present invention;

Fig. 6B is a schematic of a π network used in an energy multiplier according to an embodiment of the present invention;

Fig. 7A is a schematic of one embodiment of the network T of Fig. 6A;

Fig. 7B is a schematic of one embodiment of the π network of Fig. 6B;

Fig. 8 is a diagram of an energy multiplier network according to an embodiment of the present invention;

Fig. 9 is a schematic of a phase shifter used in the energy multiplier of Fig. 8 according to an embodiment of the present invention;

Fig. 10 is a schematic of a directional coupler used in the power multiplier of Fig. 8 according to an embodiment of the present invention;

Fig. 11 is a schematic of a second power multiplier according to the embodiment of the present invention;

Fig. 12 is a schematic diagram of a power multiplier coupled to a power distribution network according to an embodiment of the present invention; Y

Fig. 13 is a schematic diagram of multiple power multipliers coupled to a power distribution network according to one embodiment of the present invention.

Detailed description

Referring to Fig. 1, an energy multiplier 100 according to the prior art is shown. The energy multiplier 100 includes an energy multiplier waveguide 103 and a launch waveguide 106. Both the energy multiplier waveguide 103 and launch waveguide 106 are conventional transmission lines such as hollow pipes, coaxial cables, parallel cable transmission lines. Launch waveguide 106 is coupled to energy multiplier waveguide 103 using a directional coupler 109.

An electromagnetic signal generator 113 is coupled to launch waveguide 106 and generates an excitation traveling wave 116 which is launched into launch waveguide 106. Directional coupler 109 includes two slits 119 that are separated by a distance D. Distance D is approximately equal to% of the wavelength of driving forward wave 116. In this way, the electromagnetic signal generator 113 generates the driving wave 116 at a predefined frequency having a wavelength Aw that is approximately four times the electrical distance D / Aw. The launch waveguide 106 terminates in a coupled charge 123. The total length of the energy multiplier waveguide 103 is an integer multiple of the wavelength Aw of the driving forward wave 116. In the case where the energy multiplier waveguide 103 is a closed circle or a closed ring as shown, the total length of the energy multiplier waveguide is equal to its circumference.

To operate the power multiplier 100, the electromagnetic signal generator 113 generates the driving wave 116 of excitation which is fired into the launch waveguide 106. When the driving forward wave 116 reaches directional coupler 109, a portion of the driving forward wave 116 engages the energy multiplier waveguide 103, thereby creating a traveling wave 126 that propagates along the line. energy multiplier waveguide 103. Directional coupler 109 couples part of driving forward wave 116 to energy multiplier waveguide 103 so that forward wave 116 travels in a single direction around energy multiplier waveguide 103. Specifically, because the distance D between the slits 119 is approximately equal to% of the wavelength Aw of the driving forward wave 116, all the energy coupled to the energy multiplier waveguide 103 propagates in a single direction. as will be further described with reference to subsequent figures.

ES 2 647 014 T3

Furthermore, because the length of the energy multiplier waveguide 103 is an integer multiple of the wavelength A_w of the forward excitation wave 116, the forward wave 126 is especially synchronized with the forward wave 116 excitation. Under these conditions, the portion of driving forward wave 116 that is continuously coupled to power multiplier waveguide 103 is boosted or added to traveling wave 126. Consequently, the energy of traveling wave 126 can be quite large in magnitude. That is, the energy flux of the Poyntlng vector, ¹Z Re {ExH *} is pumped into the energy multiplier waveguide, which is a distributed, passive, and linear energy storage structure. The mean energy of the traveling wave 126 is distributed in the sense that it is uniformly distributed along the entire length of the energy multiplier waveguide 103.

Once deflated, the accumulation of the traveling wave 126 energy within the energy multiplier waveguide 103 will continue until the losses around the energy multiplier waveguide 103 plus the loss in the coupled load 123 terminating the launch waveguide 106 is equal to the energy generated by the electromagnetic signal generator 113. The M energy amplification and the C coupling_oroptimal pt can be calculated as follows:

]

GA²)

Cop, = 1-A 'where A is the field propagation decay for a single path of the multi-energy waveguide 103. The amount of CoPt is the coupling value for which amplification is maximized.

The reciprocating coupler has the property that the energy escaping from the energy multiplier waveguide 103 back to the launch waveguide 106 is reduced in magnitude. Furthermore, the energy leak back into the launch waveguide 106 propagates only in a single direction towards the coupled load 123 and, because that energy is in the correct phase, cancels the energy propagating from the generator 113 electromagnetic signals to the coupled load 123. Consequently, when driving wave 126 and traveling wave 126 are in phase, the coupled charge 123 dissipates little or no energy. Convenient nomograms for engineering design of lossy power multipliers operating at ultra-high frequencies are described in Tomlyasu, K., Attenuatlon in a Resonant Ring Circuit, IEEE Transactions on Microwave Theory and Techniques, vol. MTT-8, 1960, pp. 253-254.

Referring now to FIG. 2, a drawing of a portion of the energy multiplier waveguide 103 and a portion of the launch waveguide 106 is shown. Drecclonal coupler 109 is also shown. The drawing of Fig. 2 is provided to better explain the function of the reciprocating coupler 109. To explain the operation of the reciprocating coupler 109, the driving forward wave 116 is thrown into the launch waveguide 106 and approaches the first slit 119a. A part of the forward excitation wave 116 enters the multi-energy waveguide 103 through the first slit 119a and propagates in both directions into the interior of the multi-energy waveguide 103 as a wave part Wi and a part W₂ cool. The portion of driving forward wave 116 that does not pass through first slit 119a advances along launch waveguide 106 until it reaches second slit 119b. At this point, a second part of the driving forward wave 116 enters the energy multiplier waveguide 103 through the second slot 109b and propagates in both directions in the energy multiplier waveguide 103 as a part W3. wave and a wave part W4. IF the distance D between the slits is equal to% of the wavelength A_w of driving forward wave 116, as shown, then wave part W3 cancels wave part W1. Also, part W₂ Wave reinforces the wave part W4, thus resulting in wave 126 progressive. As a consequence of the cancellation of the wave parts W1 and W3, and the reinforcement of the W parts₂ and wave W4, the traveling wave 126 travels in a single direction around the power multiplier waveguide 126. Because excitation traveling wave 116 and traveling wave 126 are in phase or spatially synchronized, the portion of excitation traveling wave 116 that is coupled to the energy multiplier waveguide 103 is continuously added to wave 126. progressive wave, thus multiplying the energy of the progressive wave 126. The energy of the traveling wave 126 is a real energy. This means that there is no reactive component.

Referring now to Fig. 3, there is shown a drawing of a map 133 of the United States illustrating the problem preventing the operation of the power multipliers 100 at low frequencies, such as power frequencies. Suppose, for example, that the operating frequency is 60 Hz, which represents the

ES 2 647 014 T3 frequency of the United States power generation system. Assuming that the speed of light is approximately 300,000 km / s, at 60 Hz, the wavelength of both excitation moving wave 116 and moving wave 126 is calculated as:

<img file="ES2647014T3_D0001.tif" />

300,000 km / sec

Hz

5,000 km.

Thus, the length or circumference of a hypothetical energy multiplier waveguide 100a would have to be approximately 5,000 kilometers. Consequently, a corresponding hypothetical transmission line 101 employed in the energy multiplier waveguide 100a would be approximately 5,000 kilometers in length. Obviously, due to the size involved, the creation of such an energy multiplying waveguide 100a is not physically practical and its cost is prohibitive.

Turning now to Fig. 4A, a discussion of power transmission lines is emphasized. In FIG. 4A, a power generator 153 is electrically coupled to an electrical load 156 by a power transmission line 159. Such a transmission line 159 may be traditionally used, for example, to distribute power to homes and businesses, as can be appreciated by those of ordinary skill in the art.

Referring now to FIG. 4B, an equivalent circuit 163 is shown illustrating the equivalent impedance per unit length of transmission line 159 (FIG. 4A). Specifically, each unit length of transmission line 159 includes a series inductance Lt and a series resistor Rt. Furthermore, between the conductors of the transmission line 159 there is a capacitance Ct in parallel and a conductance Gt in parallel. Consequently, the equivalent impedance per unit length of transmission line 159 can be expressed in terms of an inductance Lt in series, a resistance Rt in series, a capacitance Ct in parallel, and a resistance Rt in parallel.

The equivalent circuit 163 reflects the fact that transmission lines 159 direct the propagation of field energy. The field energy propagating along a transmission line 159 is stored in the magnetic fields and electric fields associated with the structure of the transmission line 159 itself. Mode by mode, the magnetic field energy stored in a transmission line 159 can be equated with the magnetic field energy stored in an equivalent distributed inductance. Furthermore, the energy stored in the electric fields of the line can be equated with the energy stored in an equivalent distributed capacitance. Field energy losses per unit length of transmission line 159 can be equated with equivalent series resistive and conductive losses per unit length.

Referring now to Fig. 5, various embodiments of transmission line 159 (Fig. 4A) are shown for which the equivalent impedance can be expressed using the equivalent circuit 163 (Fig. 4B) described above. For example, transmission line 159 may comprise, for example, a parallel transmission line 159a including parallel conductors 166. Alternatively, transmission line 159 may comprise a coaxial transmission line 159b that includes an inner conductor 169 and an outer conductor 173. In yet another alternative, transmission line 159 may comprise an electrical structure 159c that includes a conductor 176 of predefined geometry located with respect to a ground plane 179. Alternatively, conductor 176 may be located relative to such a second conductor instead of ground plane 179. The predefined geometry of conductor 176 can be, for example, a helix or other geometry. In yet another alternative, the transmission line 159 may comprise an electrical structure 159d comprising a single conductor 181 in the shape of a helix or other appropriate shape. In addition, transmission line 159 may comprise other types of transmission lines and electrical structures such as, for example, conductive tracks ("striplines"), fiber optic cables, etc., as can be appreciated by those of ordinary skill in the art. matter.

Assuming that it were indeed possible to create power multiplier 100a at power frequencies such as 60 Hz, such power multiplier 100a would involve the use of a transmission cable in one of the configurations described above. In this sense, the impedance of said transmission cable can be calculated and the equivalent impedance in terms of the inductance Lt in series (Fig. 4B), the resistance Rt in series (Fig. 4B), the capacitance Ct in parallel (Fig. 4B) and the conductance Gt in parallel (Fig. 4B) can be determined.

With reference to Figs. 6A and 6B, a 183T network and a 186π network are shown that can be employed in accordance with the various embodiments of the present invention. In this sense, the 183 T network includes the impedance Zi in series and the impedance Z₂ serially. The 183 T network also includes the impedance Z₃ in parallel. The impedance Zo characteristic of a shown symmetric 183 T network can be calculated as follows:

ES 2 647 014 T3

The network 186 π includes impedances Za and Zb in parallel. The 186 π network also includes a serial or mean impedance Zc.

The impedance Zo characteristic of a symmetric 186 π network can be calculated as follows:

or

Z_c + 2Z_to)

For a further description of both the 183T network and the 186π network, reference is made to Terman, FE, Radio English Handbook. McGraw-HIH, 1943, pp. 172-178, 191-215, which is incorporated herein by reference in its entirety. The 183T network and / or the 186π network may be employed, for example, in the construction of an energy multiplier according to various embodiments of the present invention, as will be described. In particular, the impedance represented by the 183T network and / or the 186π network are forms of the equivalent circuit 163 (Fig. 4B).

With reference below to Flgs. 7A and 7B, an exemplary scheme of a 183a T network and a 186a π network are shown that may be employed in various embodiments of the present invention. In this sense, the 183a T network includes a serial L / 2 conduit shown as two separate serial L / 2 conductors. In addition, the 183a T network also includes a capacitance C in parallel. The network 183a T includes a series of loss resistors R and a conductance G in parallel that are Inherent in the conductors that make up the L / 2 conductors, the capacitance C and the electrical cable that connects these components.

The 186a π network includes an L inductance in series and C / 2 capacitances in parallel. For multiple 186a π networks that are coupled to each other in series, the adjacent parallel capacitances C / 2 can be added to become capacitance C. The 186a π networks also include a series resistor R and a parallel conductance G that are Inherent in the conductors that make up the L inductance, the C / 2 capacitances and the electrical cable that connects these components. The 183a T network and the 186a π network illustrate more particular embodiments of the 183 T networks or the 186 π networks.

With reference below to Flg. 8, an example of an energy multiplier 200 according to one embodiment of the present invention is shown. The power multiplier 200 includes a power multiplier network 203 and a launch network 206. Launch network 206 also includes a dual-channel coupler 209 that couples launch network 206 to multi-power network 203. A power source 213 is coupled to launch network 206. Furthermore, the launch network 206 terminates in a coupled load Rl.

In one embodiment, the power multiplier network 203 is a multi-connected, speed inhibit circuit constructed from a series of concentrated elements 216. As contemplated herein, the term network is defined as an interconnected structure of electrical elements. The expression multiply connected is a mathematical expression that describes the existence of a closed path in a resonator, waveguide, or other electrical structure that cannot be reduced to a point without part of the closed path passing through reglons that are external. to the geometric limits of the resonator, waveguide, or other electrical path. The power multiplier network 203 is "speed inhibitor" since the electrical structure of the power multiplier network 203 results in a reduced propagation speed of an electromagnetic wave through the power multiplier network 203 relative to the speed of an electromagnetic wave through empty space, which is the speed of light

Furthermore, the term "concentrate" is defined herein as effectively concentrated in a single location. Thus, the term "concentrated elements" refers to discrete, two-terminal, concentrated electrical elements, such as capacitance, inductances, resistance, and / or conductance. Thus, the concentrated elements, as described herein, can comprise discrete inductors, capacitors or resistors. Furthermore, as contemplated herein, the concentrated elements may also comprise diodes, transistors, and other semiconductors that can be described, for example, as non-linear resistors or conductors having a resistance or conductance that is controlled by the polarity of applied voltages or currents, etc. Furthermore, the concentrated elements may also comprise Inherent capacitors, conductors, resistors or conductances of various electrical structures such as helixes, parallel plates or other structure, as will be described later. Similar to the power multiplier network 203, the reciprocating coupler 209 is also constructed using concentrated elements.

The power multiplier network 203 is a speed inhibitor circuit that results in a slower propagation speed of an electrical disturbance such as a traveling wave. In this sense, the power multiplier network 203 has an electrical length that is equal to an integer multiple of the wavelength of the operating frequency of the power source 213. Due to the speed-inhibiting nature of the power multiplier network 203, its size is quite compact compared to the operating frequency wavelength of the power source 213. Furthermore, the dlrecclonal coupler 209 causes a phase shift that is equal to one quarter

ES 2 647 014 T3 of the length of an excitation traveling wave generated by the power source 213 at the operating frequency, as will be described later.

In one embodiment, the power multiplier network 203 is constructed from concentrated elements 216 such as, for example, L conductors and C capacitances as shown in Fig. 8. In one embodiment, L conductors they can be real inductors and the capacitances C can be real capacitors that are commercially available or can be built as needed. For example, the multilayer energy network 203 can be characterized as a ring of networks 183a T (Fig. 7A) or networks 186a π (Fig. 7B) interconnected, although networks 183a T (Fig. 7A) or networks 186a π (Fig. 7B) interconnected may be arranged in a multiply connected structure other than a ring. Each of the networks 183a T or the networks 186a π can be considered a section of the power multiplier network 203. In this sense, supposing that the power multiplier network 203 comprises a number of networks 183a T, then each L / 2 conductors can be divided into two L / 2 conductors in series that constitute the L / 2 conductors in series, such as it is described in network 183a T (Fig. 7A). Similarly, assuming that the power multiplier network 203 comprises a number of networks 186a π, each capacitance C can also be considered as a pair of capacitances C / 2 in parallel, where each of the capacitances C / 2 in parallel constitutes one of the capacitances C / 2 in parallel of the network 186a π (Fig. 7B). Regardless of whether 183a T networks or 186a π networks are used to create the sections of the power multiplier network 203, each of the 183a or 186a networks results in an offset Φ_ε predefined phase.

Supposing that 183a T networks or 186a π networks are to be used to construct the power multiplier network 203 at a certain frequency f and a certain quality factor Q, then the values ​​for the elements 216 concentrated such as the inductances L and the capacitances C u other concentrated elements are determined. The quality Q factor is conventionally defined as

<img file="ES2647014T3_D0002.tif" />

Said values ​​can be calculated from the known characteristic impedance Zo and the constant and complex propagation of the transmission line of a predetermined part of the hypothetical transmission line 101 (Fig. 3) of the hypothetical power multiplier 100a. In this sense, the characteristic impedance Zo and the constant and complex propagation of the transmission line can be calculated for a predefined unit length of the hypothetical transmission line 101 as follows:

Z = R_T + jeaL_T,

Y = Gf + jfííCj,

Z „= 7z / 7 = J (R_T+ j0L_TX (G_TíiSc ^, yy - TZV = 4R_T+&gt; L, j (G77j (oC_T), where Z is the serial impedance per unit length of the transmission line, Y is the parallel admittance per unit length of the transmission line. In the case of low loss (that is, Rt = 0 and G- = 0), the characteristic impedance is reduced to

Furthermore, the velocity of propagation can be calculated as

To determine the values ​​of R_T, Lt, Gt and Ct for a given section of transmission line 159, various references that provide such information can be consulted, such as, for example, Terman, FE, Radio Englneerlng Handbook, McGraw-HIII, 1943, pp. 172-178, 191-215, or other references, as can be appreciated.

Once the characteristic impedance Zo is known for a predefined part of the hypothetical transmission line 101,

ES 2 647 014 T3 then the complex electrical length Θ of the predefined part of the hypothetical transmission line 101 is calculated as

Θ = γ1 where I is the physical length of the predefined part of the hypothetical transmission line 101. Given the characteristic impedance Zo, the constant and complex propagation of the transmission line and the electrical length Θ of the predefined part of the hypothetical transmission line 101, the impedance Zi and Z₂ in series, and the Z pedal₃ in parallel of the 183 T network (Fig. 6A) can be calculated as follows:

A = tgh (9/2), and

Z₃ = Z₀/ senh (o),

Alternatively, the impedances Za and Zb in parallel, and the mean impedance Zc of the 186 π network can be calculated as follows:

Z_TO = Z_AND = Z_Q Coth (0/2), and Z_c = Z₀ senhfc).

Once the Zi and Z pedaloses are known₂ in series, and the Z pedal₃ in parallel of the 183 T network, or the parallels Za and Zb in parallel, and the mean impedance Zc of the 186 π network, then the corresponding values ​​for L and C can be determined. Assuming, for example, that the impedances Za and Zb in parallel, and the mean impedance Zc of the 186 π network, then the interduct L associated with the mean impedance Zc can be calculated from them, where

Z_G = Γ 4 jroL,

Furthermore, the capacitance C associated with the impedances Za and Zb in parallel can be calculated, where

<img file="ES2647014T3_D0003.tif" />

There may be a case where L and C are too large to be practically represented as a concentrated element 216. IF this is the case, then an Inverse calculation or Inverse mapping can be performed using known values ​​for L and C to determine which part of the hypothetical transmission line 101 can be represented by a given 183T network or 186π network. In this sense, it can be determined how many 183 T networks or 186 π networks can necessarily be used in a given power multiplier network 203. In this sense, values ​​for L and C can be chosen in view of the calculated values ​​for L and C identified above.

Assuming that the impedances Zi and Z₂ in series, and the Z pedal₃ in parallel network 183 T are calculated from predetermined values ​​for L and C, then the characteristic impedance Zo and the constant and complex propagation of the transmission line can be calculated as follows:

Z_or = yz, (z, + 2ZJ. y

<img file="ES2647014T3_D0004.tif" />

Alternatively, assuming that the impedances Za and Zb in parallel, and the mean impedance Zc of the 186 π network are calculated from default values ​​for L and C, then the characteristic impedance Zo and the propagation constant y complex transmission line v can be calculated as follows:

ES 2 647 014 T3

7=7 _ ^c. ⁰ Mz, + 2Z_? γ = ArcT g

Jz_c(z_c + 2Z J

z. + z_c

Once the length I of the hypothetical transmission line 101 that is represented by a 183T network or a specific 186π network is known, then it can be determined how many 183T networks or similar 186 π networks are necessary to simulate the impedance of the entire hypothetical transmission line 101. In this way, by performing the forward and backward calculations described above, the overall values ​​for the inductances L and capacitances C of the power multiplier network 203 can be determined.

Furthermore, the power multiplier network 203 further comprises a phase shifter 219. The phase shifter 219 comprises a circuit constructed from concentrated elements that is combined in series with a portion of the reciprocating coupler 209 to form the L channel of the specific section within which the reciprocating coupler 209 is located.

The power multiplier network 203 also includes a diverter 223 that couples the power multiplier network 203 to a load 226. Diverter 223 is defined herein as an electrical circuit or element that can be used to divert or redir all or a portion of a traveling wave from the power multiplier network 203 to the load 226. In this regard, diverter 223 may comprise, for example, a Switch, Relay, Solid State Switch, Plasma Switch, or other device with similar capability. Diverter 223 may also be a circuit having an electrical window that is biased using a predefined control voltage or current to divert energy within a traveling wave to load 226, depending on the state of the control voltage or current. , etc.

During operation, the power source 213 is used to launch a traveling wave of excitation into the launch network 206. The driving wave of excitation can be, for example, a sine wave or other suitable shape. The reciprocating coupler 209 couples at least a portion of the driving wave of excitation from the launch network 206 to the power multiplier network 203, resulting in a traveling wave that propagates within the power multiplier network 203. Because the electrical length of the power multiplier network 203 is an integer multiple of the wavelength of the power source 213 and that the direct coupler 209 is equal to% of the wavelength of the power source 213, then the traveling wave propagating within the power multiplier network 203 is continually boosted by the portion of the excitation traveling wave coupled to the power multiplier network 203. Furthermore, the traveling wave propagates in only one direction around the power multiplier network 203. This results in an energy magnification M of the traveling wave energy by a predefined factor that can be many times greater than the energy of the energy source 213, depending on the losses and tolerances of the concentrated elements 216 and other factors. .

Both the driving wave of excitation thrown into the launch network 206 and the traveling wave that propagates around the power multiplier network 203 can be AC ​​power signals such as electrical power signals generated at 50 Hz, 60 Hz, 400 Hz or any other power frequency that may be found in electrical generating systems in the United States and in countries around the world. In any case, however, the frequency of the driving traveling wave, traveling wave, and power source 213 can be any possible frequency, although they typically correspond to frequencies with wavelengths for which the closed path length of the power multiplier network 203 is approximately 1/10 of the wavelength or less of the traveling wave.

When the driving wave of excitation is applied to the launch network 206, the energy of the traveling wave increases continuously with time until it reaches a maximum energy. The maximum energy is reached when the losses in the power multiplier network 203 plus the losses in the coupled load Rl are equal to the energy supplied by the power source 213. When maximum power is reached, diverter 223 may be actuated to direct the traveling wave from power multiplier network 203 to electrical load 226. In a typical situation, it may take up to about a dozen cycles to reach the maximum power in the multi-power network 203, although it is possible that the maximum power will be reached in more or fewer cycles. Alternatively, diverter 223 may be actuated to direct the traveling wave from power multiplier network 203 at any time deemed appropriate such as, for example, when accumulated energy in power multiplier network 203 reaches any threshold value. predefined, etc.

The power multiplier 200 provides significant advantages in that it facilitates true power multiplication at lower power frequencies, such as the operating frequencies of power distribution systems.

ES 2 647 014 T3 electrical power from around the world operating, for example, at 50 Hz, 60 Hz, 400 Hz or other low frequencies. The speed-inhibiting nature of the power multiplier network 203 facilitates the creation of a power multiplier 200 that can operate at such low power generation frequencies with surprising size reduction. That is, where previous theory taught that power multipliers operating at conventional power generation frequencies would require a hypothetical waveguide extending thousands of kilometers as described with reference to Fig. 3, can now be created in a compact size that can be housed, for example, in a small room.

The velocity of traveling wave propagation through the energy multiplier network 203 relative to the velocity of a traveling wave through empty space is described herein as the velocity factor. The natural speed inhibitory force of the power multiplier network 203 allows for speed factors that are on the order of 1 / 1,000,000, although still smaller speed factors can be achieved.

In addition, the energy multiplier 200 may further include a number of launch networks 206, wherein each launch network 206 is coupled to the energy multiplier network 203 via a directional coupler 209. Such a configuration would facilitate a corresponding increase in the rate at which traveling wave energy accumulates during operation of the energy multiplier 200.

In an alternative embodiment, the traveling wave may be a lone wave propagating around the energy multiplier network 203. In order to propagate a lone wave around the energy multiplier network 203, the energy multiplier network 203 is constructed to include non-linear elements such as, for example, diodes, transistors, or other active components so that it is non-linear. linear and dispersive. Thus, non-linear components are defined herein as components that provide an output that has an amplitude that is not linearly proportional to the input, as can be appreciated by those of ordinary skill in the art. By constructing the energy multiplier network 203 from a suitable network of non-linear elements and / or a combination of linear and non-linear elements, a lone wave can propagate around the energy multiplier network 203. In this sense, the energy source 213 would be a pulse generator that generates and launches a traveling wave of excitation to the launch network 206. To achieve energy multiplication, a lone excitation traveling wave would have to be spatially synchronized with the lone traveling wave. In addition, the launch network 206, directional coupler 209, and phase shifter 219 can be constructed to include elements that are non-linear and dispersive in nature to facilitate the propagation of solitary waves therethrough.

It should be appreciated that as the gain of the power multiplier network 203 increases, its quality Q factor increases and its bandwidth BW narrows around the operating frequency. In one embodiment, this may be a desirable feature for a strictly monochromatic system. If wider bandwidths BW are desired, the electrical bandwidth BW of the power multiplier network 203 can be tailored to the specific application. For example, low loss power multiplier networks 203 with wider passbands and controlled shapes can be constructed following various electrical filter designs. See, for example, Matthaei, GL, L. Young, and EMT Jones, Microwave Filters, Impedance Matching Networks, and Coupling Structures, McGrawHill, 1964; and Fano, RM, Theoretical Limitations on Broadband Matching of Arbitrary Impedances, Journal of the Franklin Institute, vol. 249, 1950, pp. 53-83 and 129-155.

In another embodiment, the energy multiplier 200 described above can also be constructed incorporating so-called Tracking Filter design techniques so that the electrical passband of the energy multiplier 200 can be dynamically and automatically controlled to consistently track variations. of frequency and phase of the power source 213 while maintaining the desired operating properties described above. In the implementation of an energy multiplier 200 with a dynamic electrical passband, the frequency of the energy source 213 is controlled and compared to the resonant frequency of the energy multiplier network 203. An error signal can be generated from such a comparison and can be used in a feedback loop to dynamically modify the parameters of the ring components such as the concentrated elements of the energy multiplier network 203 to tune it to the spectral variations of the signal. source 213 of energy. In that case, the concentrated elements described above can be parametrically dynamic with variable parameters as can be seen.

Referring now to FIG. 9, a schematic is shown providing an example of the phase shifter 219 in accordance with one aspect of the present invention. The phase shifter 219 comprises a lattice 183a T (Fig. 7A), although a lattice 186a π may also be employed. In this sense, the phase shifter 219 includes inductances Lt in series and a capacitance Ct in parallel. In this sense, the phase shifter 219 is constructed from concentrated elements as part of the energy multiplier network 203.

The inductances Lt in series and the capacitance Ct in parallel are specified to result in a phase shift Φ5. Inductances Lt in series and / or capacitance Ct in parallel (assuming a 186a T network is used) 10

ES 2 647 014 T3 can be variables to allow the displacement Φ_ε phase is adjusted as necessary to compensate for any inaccuracies in the displacements Φ_ε phase of each section and on the phase shift Θ of the directional coupler 209. This is done to ensure that the total phase shift presented by the power multiplier network 203 is an integer multiple of 360 degrees for the wavelength of the power source 213. Next, the specific calculations that are performed to determine the values ​​of the inductances Lt and the capacitance Ct in parallel will be described.

Referring to FIG. 10, there is shown a schematic illustrating an example of the directional coupler 209 in accordance with one aspect of the present invention. Directional coupler 209 comprises a series of concentrated elements. Said directional coupler 209 ensures that the traveling wave propagates in a single direction along the power multiplier network 203 and that the traveling wave boost is achieved with the portion of the excitation traveling wave propagating through the launch network 206.

With the above description of the power multiplier network 203, directional coupler 209, and phase shifter 219, the total phase shift presented by power multiplier network 203 can be determined as follows:

Í&gt; PMW - &lt;^ (Ν-ΐ) -ί-φ -? - θ_τ where N is equal to the number of sections in the power multiplier network 203.

Furthermore, the diverter (Fig. 8) can be constructed similarly to the directional coupler 209 in that the values ​​of the coupling capacitances are used to control the rate at which power exits the power multiplier network 203.

Referring now to FIG. 11, a schematic of an energy multiplier 250 according to another embodiment of the present invention is shown. The power multiplier 250 includes a power multiplier network 253 that is constructed from a toroldal propeller as shown, or any of its variants comprising Left-hand, right-hand, or Left-hand and left-hand propeller overlays. right hand as described in Canadian Patent 1,186,049, US Patent 4,622,558 and US Patent 4,751,515, each of these references filed by James F. Corum, The full text of each of these references being incorporated by reference herein. In this sense, the toroldal helix includes the L inductances (Fig. 8) by virtue of its construction. Furthermore, the Impedance presented by the toroldal helix includes capacitances that can be appreciated by people of ordinary skill in the art (see Krause, John D., Antennas, McGraw-HIII, 1^to edition, 1950, Fig. 7.2). The power multiplier 250 includes a launch network 256 that is coupled to the power multiplier network 253 via a directional coupler 259. Power multiplier 250 also includes diverter 223 that couples an output from power multiplier 250 to a load 226 as shown. Power source 213 is coupled to launch network 256 and launches a driving wave of excitation to launch network 256 in a manner similar to that described with reference to power multiplier 200. Similarly, launch network 256 terminates on a coupled Rm load.

Directional coupler 259 can be, for example, a helix section or even a 186π lattice (Fig. 7B) as shown. Directional coupler 259 imposes a phase shift of% wavelength of the driving traveling wave in a manner similar to that described above.

The operation of the power multiplier 250 is substantially similar to that described with reference to the power multiplier 200 of FIG. 8. Power multiplier 250 illustrates the fact that power multiplier network 253 may comprise one or more electrical structures such as a toroldal helix, two or more criss-cross helixes, a helix wound in opposite directions, or other electrical structures that include capacitances and Inherent conductors that act as the concentrated elements 216 (Fig. 8) such as the L conductors (Fig. 8) and capacitances C (Fig. 8).

With reference again to Fig. 8, once the values ​​for the inductances L and the capacitances C have been determined for each section of the power multiplier 200 comprising 183 T networks (Fig. 6A) or 186 π networks (Fig. 6B ), then the actual energy amplification that can be achieved by the resulting energy multiplier 200 can be determined, given the values ​​for the concentrated elements (i.e., capacitances C in parallel and inductances L in series). Specifically, the concentrated elements are specified to achieve a predefined phase shift for each section at the predefined operating frequency.

The following describes the sequence of calculations that are performed to determine the values ​​for the concentrated elements 216, such as the capacitances C and the inductances L of the power multiplier 200. In the following calculations, it is assumed that each section of the power multiplier network 203 comprises networks 186 π (Fig. 6B). For 11

ES 2 647 014 T3 start, the operating frequency f of the energy multiplier 200 is specified. In addition, both the inductance L and the capacitance C of each section of the energy multiplier network 203 are specified based on the values ​​for said elements, identified above. Furthermore, a quality factor Q is specified for the inductances L of each section of the energy multiplier network 203. The frequency, in terms of radians / s, is calculated as ω = 2τϊί radians ^,

Also, the resistance across each of the inductances L is calculated as r = - Ohms.

Q

Next, the impedance Zc is calculated as follows:

Z_c = r + ÍüL Ohms, where i represents V-1 as known to those of ordinary skill in the art. Given the capacitances C specified above, the impedances Za and Zb in parallel are calculated as follows:

Z_TO -Z_B Ohms.

ioC

Then the characteristic impedance Zo is calculated as follows:

<img file="ES2647014T3_D0005.tif" />

^{Zs = Z}^ '(Z_c.,. 2Z_to)

Ohms.

Characteristic impedance is defined as the ratio of the forward wave voltage to the forward wave current. In this sense, a physical measurement of the characteristic impedance of each section can be made and it can be compared with the calculated characteristic impedance Zo to verify its precision.

In addition, the constant y of propagation for each section is calculated as follows:

And "atgh yz_c(Z_{c +} 2Zj

The attenuation constant α for each section and the phase constant β for each section are defined as ^αί «χ; ηπ = Μΐ) Nepers / section, and β ^ = Im (?) radians / section

The phase shift for each section can be calculated as φ - (57.296 Grad / Rad) Ji_saction Degrees.

The velocity of the traveling wave in sections per second propagating through the energy multiplier network 203 is calculated as v = —---- sections / second.

β section

Next, the electrical circumference Ca of the energy multiplier network 203 is specified in terms of 12

ES 2 647 014 T3 wavelengths at operating frequency in degrees as

C_Grad= C, {360 Degrees / wavelength) Degrees.

Next, the number of sections N (either T networks or π networks) is calculated as

Φ

Once the number N of sections is known, then the loss resistance Rc around the closed path of the power multiplier network 203 can be calculated as

R_c = Nr Ohms.

where r is as defined above. The field propagation decay A for a single path of the power multiplier network 203 can be calculated as₌ g.^-orseDtion^W

The attenuation A® around the power multiplier network 203 is calculated as

TO_d0 = -2Clog (A).

The Impulse duration T of a peripheral disturbance is calculated as - seconds.

The energy increase M of the energy multiplier 200 at optimal coupling is calculated as

M- ¹ (I ^ A)

The increase Mjb of energy expressed in decbels is calculated as

M_dü = 10Jog (M).

Coupling C_opt optimal is calculated as dpi = 1-AA

Coupling C_opt Optimum is calculated in declines (dB) as = Wlog (C_opl) dB.

In addition, a useful reference that can be consulted to determine the various elements of the direcclonal coupler 209 and phase shifter 219 is Matthael, GL, L. Young, and Ε. Μ. T. Jones, Mlcrowave Fllters, Impedance Matching Networks, and Coupling Structures, McGraw-HIII, 1964, (see Chapter 14). Although specific circuit designs can be described herein that may be employed such as the 209 digital coupler and the

ES 2 647 014 T3 phase shifter 219, it is understood that other circuit designs and circuit structures may also be employed, such alternative designs being included within the scope of the present invention.

Referring now to FIG. 12, the power multiplier 200 is shown coupled to a power distribution network 300 in accordance with one embodiment of the present invention. Although the energy multiplier 200 used by the energy multiplier network 203 is shown in Fig. 12, it is understood that other embodiments of energy multipliers may be employed as described herein such as energy multiplier 250, where energy multiplier 200 and energy multiplier network 203 are described herein merely as an example.

The power distribution network 300 may be, for example, an electrical network such as the North American electrical network or other electrical networks anywhere in the world. As shown in FIG. 12, launch network 206 is coupled to power distribution network 300. The diverter outlet 223 is also coupled to the power distribution network 300.

Diverter 223 receives a load feedback 303 which may comprise, for example, a load feedback signal generated based on a current electrical load in the power distribution network 300. Directional coupler 209 can be selectively coupled to launch network 206, or launch network 206 can be selectively coupled to power distribution network 300 to facilitate controlled energy input to power multiplier network 203 from the network. 300 of energy distribution, thus resulting in the storage of energy in the energy multiplier network 203 of the energy multiplier 200. Alternatively, directional coupler 209 may be configured to control the rate at which power is fed to power multiplier network 203. By virtue of the fact that launch network 206 and diverter 223 are coupled to power distribution network 300, power multiplier network 203 can be used to store energy from power distribution network 300 and to supply power. to power distribution network 300 as desired.

Diverter 223 may be configured to control the output of power multiplier network 203 in response to load feedback 303. In this sense, the energy stored in the power multiplier network 203 can be supplied, for example, to the power distribution network 300 to provide power when there is an abrupt increase in the electrical load associated with the power distribution network 300. Energy.

Because facilities supplying power to power distribution networks 300 can experience severe mismatches between peak and average load demands, power multiplier network 203 can be advantageously employed to perform power smoothing. For example, the power multiplier network 203 can be used in locations local to electrical loads that may be remote from power generation stations to “smooth out voltage drops and blackouts due to installations with high peak load demands. a-average. In this sense, the energy multiplier network 203 can be coupled to various locations of the energy distribution networks 300 to provide a local controlled smooth transition between load states, allowing temporary energy storage that can be harnessed as needed. .

This can reduce electromechanical stress on existing power generation equipment at power generation stations. Specifically, when large load swings and transients occur in power distribution systems 300, significant electromechanical stresses can occur on rotating machinery used in power generation. For example, the individual occurrence of a large transient or the repeated occurrences of smaller transients over time can result in the catastrophic failure of shafts and other mechanical components of electrical generators. Additionally, electrical wiring failures can occur at generators and at other points in electrical distribution systems. In addition, oscillations and load transients can affect the frequency and phase stability of electrical generators as they react to changes in electrical loads. The power multiplier network can be used to eliminate such stresses on power generation and distribution equipment, and can guarantee frequency and phase stability in existing power distribution networks 300.

In circumstances where there is an intermediate electrical charging point such as a city between electric generating stations and a remote load, it is possible that, during times of heavy load, the capacity demanded cannot be transported from the electric generating station to remote charging through the intermediate electrical charging point. In this way, an energy multiplier 200 including the energy multiplier network 203 can be employed, for example, to address the problem of electrical traffic congestion during peak hours around said intermediate charging point. For example, the power multiplier network 203 may be coupled to the power distribution network 300 near the intermediate load point to allow the storage of energy that can be accessed in such times of heavy traffic, thereby smoothing the demand. and preventing loss of service in remote charging.

ES 2 647 014 T3

To facilitate effective power smoothing in a given power distribution network 300, one or more power multiplier networks 203 may be coupled to parts subjected to stress due to the demand of a given power distribution network 300. As described above, such parts stressed due to the demand of a power distribution network 300 may be in locations close to cities or other large loads that experience high peak-to-average load demands. Furthermore, said parts subjected to stress due to demand may be close to intermediate electrical charging points. Furthermore, as will be appreciated, other locations of various power distribution networks 300 may be stressed due to demand.

The various embodiments of the energy multipliers described herein, including the energy multiplier 200 that the energy multiplier network 203 employs, are ideal for power smoothing in a power distribution network 300 since the energy stored in these networks of multiplication of energy is available almost instantaneously. Accordingly, the power multiplier network 203 can be used, for example, to supply power when the generating equipment in the power distribution network 300 cannot react fast enough to compensate for abrupt changes, such as increases in load. electrical. In this sense, one or more power multiplier networks 203 may be used, for example, to supply power to the power distribution network 300 for periods of time to facilitate the adjustment of power generation systems coupled to the distribution network. of energy to supply energy to the increased electrical load after the occurrence of the abrupt increase.

Referring to FIG. 13, a number of power multipliers 200/250 are shown employing power multiplier networks 203/253 coupled to power distribution network 300 in accordance with another embodiment of the present invention. Although power multiplier networks 203/253 are shown, as can be appreciated, other embodiments of the power multiplier networks may be employed. An energy multiplier control system 206 is provided with outputs that are electrically coupled to each of the diverters 223 of the respective energy multipliers 200/250.

The power multiplier control system 206 generates control outputs that are applied to the diverters 223 to control the release of energy from each of the power multiplier networks 203/253 to the power distribution network 300 in response to the load feedback from power distribution network 300. In one embodiment, the power multiplier control system 206 is configured to apply power from each of the power multiplier networks 203/253 to the power distribution network 300 in a sequential order. In this sense, the period of time during which the energy distribution network 300 can receive energy from the energy multiplier networks 203/253 increases as a function of the number of energy multiplier networks 203/253 used. In this regard, multiple 203/253 power multiplier networks can be employed to provide adequate time for the generator set to adjust to changing electrical loads without stressing the mechanical and electrical components of the generator set. Alternatively, energy stored in multiple grids of the power multiplier grids 203/253 can be applied to the power distribution grid 300 simultaneously to satisfy extreme load increases.

In addition, the elements that are used to construct the various embodiments of the 203/253 energy multiplier networks described herein can be constructed using low loss dielectrics and high permittivity in capacitances, and low loss conductors in inductances (such as inductance coils). Such low loss conductors can be, for example, cryogenic conductors and / or superconductors. Such low loss conductors allow for much greater storage capacity with extremely high efficiencies. Specifically, because energy storage will increase in energy multiplier networks 203/253 as described herein until the losses experienced in energy multiplier networks 203/253 equal the energy input, where a Given energy multiplier network is built with extremely low loss conductors, it follows that large amounts of energy can be stored.