ES2631906T3

Apparatus and method for generating audio subband values and apparatus and method for generating time-domain audio samples

Abstract

An apparatus intended to generate complex audio subband values in audio subband channels, comprising: an analysis window generator (110) for generating a window of a frame (120) of temporary domain audio input samples, which are arranged in a time sequence that extends from an early sample to a late sample, using a analysis window function (190) comprising a sequence of window coefficients in order to obtain window samples, said analysis window function (190) being composed of a first group (200) of window coefficients, which comprise a first part of the sequence of window coefficients, and a second group (210) of window coefficients, which it comprises a second part of the window coefficient sequence, the first part comprising less window coefficients than the second part, so that a global energy value of the window coefficients in the first part is greater than a global energy value of the window coefficients of the second part, the first group of window coefficients being used to generate the late sample window in the temporal domain and using the second group of window coefficients to generate the window of early samples in the temporal domain, configuring the analysis window generator to function following the equation: ** Formula ** where zi, n is a sample in the real value window corresponding to the block index i and the sample index n of a frame, the value xi, n being an audio input sample in the time domain of value real corresponding to the block index i and the sample index n, where w contains real value window coefficients of the window analysis function, and where N is a parameter that designates the number of samples in a block; and a calculator (170) for calculating the audio subband values based on the window samples, said calculator (170) comprising a time / frequency converter adapted to generate the audio subband values, so that all subband values based on a frame (150) of window samples form a spectral representation of the window samples of the frame (150) of window samples, and adapting said time / frequency converter to generate audio subband values of complex value, and configuring said calculator to function governed by the equations ** Formula ** where k, an index of spectral coefficients or band index, an integer in the range 0 <= k <N, where XReal, i, k and Ximag, i, k represent the real part and the imaginary part of an audio subband value of complex value corresponding to the block index i and the index of spectral coefficients k, and being not a parameter representing an index option, and so that the intended apparatus To generate the audio subband values of complex value in the audio subband channels is a bank of analysis filters.

ES2631906T3, drawing sheet 1 — Sheet 1 of 33

Term

1.1 yearsto projected expiry

Projected expiry 23 October 2027, counted from filing; an application has no term until it is granted.

5 claims: 4 independent, 1 dependent

1
ES 2 631 906 T3 Reivindicaciones 1. Aparato destinado a generar valores de subbanda de audio de valor complejo en canales de subbanda de audio, que comprende:un generador de ventanas de análisis (110) para generar una ventana de una trama (120) de muestras de entrada de audio de dominio temporal, que se disponen en una secuencia temporal que se extiende desde una muestra temprana a una muestra tardía, utilizando una función de ventana de análisis (190) que comprende una secuencia de coeficientes de ventana a fin de obtener muestras en ventana, dicha función de ventana de análisis (190) estando compuesta de un primer grupo (200) de coeficientes de ventana, que comprenden una primera parte de la secuencia de coeficientes de ventana, y de un segundo grupo (210) de coeficientes de ventana, que comprende una segunda parte de la secuencia de coeficientes de ventana, la primera parte comprendiendo menos coeficientes de ventana que la segunda parte, de modo que un valor de energía global de los coeficientes de ventana en la primera parte es superior a un valor de energía global de los coeficientes de ventana de la segunda parte, utilizándose el primer grupo de coeficientes de ventana para generar la ventana de muestras tardías en el dominio temporal y utilizándose el segundo grupo de coeficientes de ventana para generar la ventana de muestras tempranas en el dominio temporal, configurándose el generador de ventanas de análisis para funcionar rigiéndose por la ecuación: w(10N - 1 - n) x Jn for 0 n 10 · N , siendo z¡, „ una muestra en ventana de valor real correspondiente al índice de bloque I y al índice de muestra n de una trama, siendo el valor x¡, „ es una muestra de entrada de audio en el dominio temporal de valor real correspondiente al índice de bloque I y al índice de muestra n, siendo w contiene coeficientes de ventana de valor real de la función de análisis de ventana, y siendo N es un parámetro que designa el número de muestras en un bloque;y una calculadora (170) para calcular los valores de subbanda de audio basándose en las muestras en ventana, comprendiendo dicha calculadora (170) un convertidor tiempo/frecuencia adaptado para generar los valores de subbanda de audio, de modo que todos los valores de subbanda basados en una trama (150) de muestras en ventana forman una representación espectral de las muestras en ventana de la trama (150) de muestras en ventana, y adaptando dicho convertidor de tiempo/frecuencia para generar valores de subbanda de audio de valor complejo, y configurando dicha calculadora para funcionar rigiéndose por las ecuaciones: siendo k, un índice de coeficientes espectrales o índice de banda, un número entero en el rango 0 k N, donde XR ea i, ¡ ,k y X¡ ma g, ¡, k representan la parte real y la parte imaginaria de un valor de subbanda de audio de valor complejo correspondiente al índice de bloque i y al índice de coeficientes espectrales k, y siendo no un parámetro representando una opción de índice, y de modo que el aparato destinado a generar los valores de subbanda de audio de valor complejo en los canales de subbanda de audio es un banco de filtros de análisis.
2
Aparato (300) destinado a generar muestras de audio de valor real en el dominio temporal, que comprende:una calculadora (310) para calcular una secuencia (330) de muestras intermedias en el dominio temporal a partir de valores de subbanda de audio en canales de subbanda de audio, dicha secuencia comprendiendo muestras tempranas intermedias en el dominio temporal y muestras tardías en el dominio temporal, dicha calculadora (310) comprendiendo un convertidor tiempo/frecuencia adaptado para generar la secuencia de muestras intermedias en el dominio temporal, de modo que los valores de subbanda de audio proporcionados a la calculadora (310) forman una representación espectral de la secuencia de muestras intermedias en el dominio temporal, y dicho convertidor tiempo/frecuencia estando adaptado para generar la secuencia de muestras intermedias en el dominio temporal basándose en valores de subbanda de audio de valor complejo;ES 2 631 906 T3 configurándose la calculadora para que funcione rigiéndose por la ecuación: z j - Re al, i, k k=Q (π / / 1 — (η + n n I k + — °\ 2 hn agj.ti ' í 1 sin —(n + n n ) k + — N °l 2 \\ para 0 n 10 N, siendo x’¡, „ una muestra Intermedia en el dominio temporal de una trama correspondiente al índice de muestra n y al índice de bloque I, siendo N es un número entero que designa la longitud de un bloque, de modo que XR ea i,¡,ky X¡mag,¡,k representan la parte real y la parte Imaginarla de un valor de subbanda de audio de valor complejo correspondiente al índice de bloque I y al índice de coeficientes espectrales k, siendo no un parámetro que representa una opción de índice, y siendo N un parámetro que designa el número de muestras por bloque;un generador de ventanas de síntesis (360) para generar la ventana de la secuencia (330) de muestras Intermedias en el dominio temporal utilizando una función de ventana de síntesis (370) que comprende una secuencia de coeficientes de ventana a fin de obtener muestras en ventana Intermedias en el dominio temporal, dicha función de ventana de síntesis (370) estando compuesta de un primer grupo (420) de coeficientes de ventana, que comprenden una primera parte de la secuencia de coeficientes de ventana, y de un segundo grupo (430) de coeficientes de ventana, que comprende una segunda parte de la secuencia de coeficientes de ventana, la primera parte comprendiendo menos coeficientes de ventana que la segunda parte, de modo que un valor de energía global de los coeficientes de ventana en la primera parte es superior a un valor de energía global de los coeficientes de ventana de la segunda parte, utilizándose el primer grupo de coeficientes de ventana para generar la ventana de muestras tardías en el dominio temporal y utilizándose el segundo grupo de coeficientes de ventana para generar la ventana de muestras tempranas en el dominio temporal, configurándose el generador de ventanas de síntesis para funcionar rigiéndose por la ecuación: z x ¿ i,n w(ri x' in for 0 n 10 · N en la que z’¡, n es el valor de una muestra Intermedia en el dominio temporal en ventana correspondiente al índice de muestra n y al índice de bloque I de una trama, y en la que w(n) es un coeficiente de ventana de valor real de la función de ventana de síntesis ;y una etapa de salida de un adlclonador de superposición (400) para procesar las muestras Intermedias en ventana en el dominio temporal a fin de obtener las muestras en el dominio temporal, dicha etapa de salida de un adlclonador de superposición estando configurada para funcionar rigiéndose por la ecuación: OU ^i,n ~ Z i,n + Z i-\,n + N +Z i-2,u + 2N+ Z i-3.n + 3N +Z ¡-4,„ + 4Ν + Ζ »-5, ii+SN + Z' : ΐ-β,η + βΝ + Z ',-2. íi + TN i-S.n + SN +z'. ¡-9.H+9N para 0 η N en la que out¡, n representa una muestra de valor real en el dominio temporal correspondiente al índice de muestra n y al índice de bloque I, y de modo que el aparato destinado a generar las muestras de audio en el dominio temporal de valor real es un banco de filtros de síntesis.
3
Método destinado a generar valores de subbanda de audio de valor complejo en canales de subbanda de audio, que comprende:la generación de una ventana de una trama de muestras de entrada de audio en el dominio temporal, que se disponen en una secuencia temporal que se extienden desde una muestra temprana hasta una muestra tardía, utilizando una función de ventana de análisis que está compuesta de un primer grupo (200) de coeficientes de ventana, que comprende una primera parte de la secuencia de coeficientes de ventana, y de un segundo grupo (210) de coeficientes de ventana, que comprende una segunda parte de la secuencia de coeficientes de ventana, la primera parte comprendiendo menos coeficientes de ventana que la segunda parte, de modo que un valor de energía global de los coeficientes de ventana en la primera parte es superior a un valor de energía global de los coeficientes de ventana de la segunda parte, utilizándose el primer grupo de coeficientes de ventana (200) para generar la ventana de muestras tardías en el dominio temporal y utilizándose el segundo grupo de coeficientes de ventana (210) para generar la ventanas de muestras tempranas en el dominio temporal, de modo que la generación de ventana se basa en la ecuación: z ín = w(10N - 1 - n) · x in for 0 n 10 · N , siendo z¡, n una muestra en ventana de valor real correspondiente al índice de bloque I y al índice de muestra n de una trama, siendo x¡, n una muestra de entrada de audio en el dominio temporal de valor real ES 2 631 906 T3 correspondiente al índice de bloque i y al índice de muestra n, donde w contiene coeficientes de ventana de valor real de la función de análisis de ventana, y siendo N un parámetro que designa el número de muestras en un bloque;calculándose los valores de subbanda de audio utilizando las muestras en ventana, el cálculo comprendiendo un convertidor tiempo/frecuencia, que convierte los valores de subbanda de audio, de modo que todos los valores de subbanda basados en una trama (150) de muestras en ventana forma una representación espectral de las muestras en ventana de la trama (150) de muestras en ventana, y de modo que el convertidor tiempo/frecuencia, que convierte los valores de subbanda de audio, comprende la generación de la secuencia de muestras intermedias en el dominio temporal basándose en valores de subbanda de audio de valor complejo, y el cálculo basándose en las ecuaciones: / siendo k, un índice de coeficientes espectrales o un índice de banda, un número entero en el rango 0 k N, de modo que XR ea i,¡,k y X¡ ma g,¡,k representan la parte real y la parte imaginaria de un valor de subbanda de audio de valor complejo correspondiente al índice de bloque i y al índice de coeficientes espectrales k, siendo no un parámetro que representa una opción de índice, y de modo que el método destinado a generar los valores de subbanda de audio de valor complejo en los canales de subbanda de audio se implementa mediante un banco de filtros de análisis.
4
Método destinado a generar muestras de audio de valor real en el dominio temporal, que comprende:el cálculo de una secuencia de muestras intermedias en el dominio temporal a partir de valores de subbanda de audio en canales de subbanda de audio, dicha secuencia comprendiendo muestras tempranas intermedias en el dominio temporal y muestras tardías intermedias en el dominio temporal;el cálculo comprendiendo una conversión tiempo/frecuencia de los valores de subbanda de audio, de modo que los valores de subbanda forman una representación espectral de la secuencia de muestras intermedias en el dominio temporal;y de modo que la conversión tiempo/frecuencia comprende la generación de la secuencia de muestras intermedias en el dominio temporal basándose en valores de subbanda de audio de valor complejo;el cálculo basándose en la ecuación: X , para 0 n 10 N, siendo x’¡, n una muestra intermedia en el dominio temporal de una trama correspondiente al índice de muestra n y al índice de bloque i, siendo N un número entero que designa la longitud de un bloque, de modo que XR ea i,¡,k y X¡mag,¡,k representan la parte real y la parte imaginaria de un valor de subbanda de audio de valor complejo correspondiente al índice de bloque i y al índice de coeficientes espectrales k, siendo nO un parámetro que representa una opción de índice, y siendo N un parámetro que designa el número de muestras por bloque;la generación de la ventana de la secuencia de muestras intermedias en el dominio temporal utilizando una función de ventana de síntesis que comprende una secuencia de coeficientes de ventana a fin de obtener muestras en ventana intermedias en el dominio temporal, dicha función de ventana de síntesis estando compuesta de un primer grupo (420) de coeficientes de ventana, que comprende una primera parte de la secuencia de coeficientes de ventana, y de un segundo grupo (430) de coeficientes de ventana, que comprende una segunda parte de la secuencia de coeficientes de ventana, la primera parte comprendiendo menos coeficientes de ventana que la segunda parte, de modo que un valor energético global de un coeficiente de ventana en la primera parte es superior a un valor energético global de un coeficiente de ventana de la segunda parte, utilizándose el primer grupo de coeficientes de ventana para la generación de la ventana de las muestras tardías en el dominio temporal, y utilizándose el segundo grupo de coeficientes de ventana (420) para la generación de la ventana de las muestras tempranas intermedias en el dominio temporal, de modo que la generación de ventana se basa en la ecuación: z' ln = ' x 'i,n f° r 0 n 10 · Λ7 ES 2 631 906 T3 en la que ζ’Ι,η es el valor de la muestra Intermedia en ventana en el dominio temporal que corresponde al índice de muestra n y al índice de bloque I de una trama, y en la que w(n) es un coeficiente de ventana de valor real de la función de ventana de síntesis ;y una adición de superposición de las muestras Intermedias en ventana en el dominio temporal a fin de obtener las muestras en el dominio temporal, dicha adición de superposición basándose en la ecuación: I+Z': ,+z'. ί-Ι,η + Ν^ i-2,n + 2N ' i-l.ii+lN ' i-4,» + 4N + Z i-6,n+6N+ Z i-1.n + TN +Z i- ,ii + ZN +Z i-9,n + 9N +z', ¡-5.H + 5N para 0 η N en la que outl.n representa una muestra de valor real en el dominio temporal correspondiente al índice de muestra n y al índice de bloque I, y de modo que el método destinado a generar las muestras de audio en el dominio temporal de valor real se ¡mplementa mediante un banco de filtros de síntesis.
5
Programa con un código de programación para la Implementación, si se ejecuta en un procesador, de un método según la reivindicación 3 o según la reivindicación 4. ES 2 631 906 T3 FIG 1 mo 220 ES 2 631 906 T3 FIG2A 300 ES 2 631 906 T3 FIG 2B tzj~ 440 ES 2 631 906 T3 FIG 3 ES 2 631 906 T3 FIG 4 ES 2 631 906 T3 FIG 5 ES 2 631 906 T3 ’CO LO Ω_ O C ’ω O O +-» D D O O O S O Ll O O - Π5 Q. •I J ? D O _O O S O to C5 ES 2 631 906 T3 FIG 7A ES 2 631 906 T3 FIG 7B 160-2 160-4 160-6 160-8 160-10 160-1 ( 160-3 ( 160-5 ( 160-7 ( 160-9 ( ES 2 631 906 T3 FIG 7C ES 2 631 906 T3 FIG8A 100 ES 2 631 906 T3 FIG 8B 101 ES 2 631 906 T3 FIG 8C 102 ES 2 631 906 T3 FIG8D 103 ES 2 631 906 T3 FIG 9A S400 S410 S420 S430 S440 S450 Ξ460 S470 S480 función [y, estado] = ldfb80 (x, estado) % búfer actualización estado (640-64+(1:64)} = x;% ventana aplicación win_ana = ldfb80„win;% asume las últimas muestras alineadas a la derecha del búfer x_win_orig - state. W w¡n_ana;% preparar bloque de x_stack = Igualar (X WID Or¡g,128,5), % cambio de signo x_stack(:,2:2:4)) = -x_stackf: I 2;2:4»;% colapsar bloque de x_stack = sum(-x_stack(end:-1:1 ,:),2)’;% FFT impar de datos con ventanas temp = fft( x_stack. *exp(-1 ΐ *p i * (0:128-1 )/128)};% posterior giro nt = (64+1)/2 y = 2conj (temp (1:64) ?exp(-2¡*pl”((0:64-1) +0.5)*m/128))·, % búfer de desplazamiento — (1:640-64)= (64+(1:640-64));104 ES 2 631 906 T3 FIG 9B S40Ó' S412 S420 función [y, estado] = ldfb80 (x, estado) i % búfer actualización State (320-32+(1:32)) = x;% ventana aplicación win_ana = ldfb80jv¡n();win_ana = (win_ana (1:2:end) + w¡n_ana (2:2:end))/2;% asume las últimas muestras alineadas a la derecha del búfer S430’ x _ Wfn _ orf 0 wm ana: S44D % preparar bloque de x.stack = igualar ( X _ W ¡ n _orig ,64,5)j % cambio de signo 545 θ x_stack(:, (2:2:4)) = -x_stack{:, (2:2:4));X % colapsar bloque de xstack = sum(-x_stack(end 1:1,:) r 2}’: % FFT impar de datos con ventanas tenp = fft( x_slack.*exp(-1 ¡*p¡ * (0:64-1 )/64));S470’ % posterior giro m = (32+1)/2 $ 4 80* 7 = 2*conj (temp (1:32) .*exp(-2i*p¡*((0:32-1)+0.5)*nV64));% búfer de desplazamiento ·«·· (1:320-32)= ·«·· (32+(1:320-32)): 105 ES 2 631 906 T3 FIG10A S500 S510 función [y, estado] = ldfb80 (x, estado) % previo giro m = (64+1)/2;. temp = 0.5*conj (x).*exp( (2i*p¡*((0:64-1)+0.5)*m/128));% simetría impar temp = [temp conj (temp (64:-1:1))): S520 S53O S540 S550 S560 % FFT impar y_J nl = real (¡ffl (temp) .* exp(l*p¡*(0:128-1 )/128)): % expandir dato;alternar cambio de y_knl = -y_knl (128:-1:1);tmp = [y_knl -y_knl y_knl -y_Rni y_knl] % ventana síntesis r w¡n_ana - ld1b80_win;w¡n syn = win_ana(end:-1:1);% ventana aplicación tmp = tmp.*winsys;% búfer actualización ' (640-64+(1:64)) = 0;+ tmp;estado — estado S570 S580 % obtener salida y — ~..d. % búfer de desplazamiento estad ° (1:640-64) = estad ° (64+1:640);106 ES 2 631 906 T3 FIG10B S500' S510’ S520’ S530' S542 S550 S560' S570* S580 1 función [y, estado] = ldfb80 (x, estado) % previo giro m = (32+1)/2: temp = 0.5*conj (x).‘exp( (2¡*pi*((0:32-1)+0.5)*m/64));% simetría impar temp = [temp conj (temp (32:-1:1 ))j;% FFT impar y_knl - real {¡fft {temp) .* exp í *pi* (0:64-1 )/64)): % expandir dato;alternar cambio de y_knl = -yjnl {64:-1:1): tmp = [y_knl -y_knl y_knl -y^knl y_knlj’¡ % ventana síntesis wínana = ldfbSQ_w¡n;wín^syn = win_ana(end:-1:1);w¡n_syn = (win_syn(1:2:end)+wfn_syn(2:2:end))/2;% ventana aplicación tmp = tmp,*wín_sys;% búfer actualización — ¢320-32+(1:32)) = 0: estado = -J- tfílp;% obtener salida y= · !, ' ίο (1:32): % búfer de desplazamiento estad» (1:320-64) = estad ° (32+1:320): 107 ES 2 631 906 T3 u_ O _l E o O (Z) 3 C (Λ (0 (0 c C (0 (0 +-» +-» c c D D f Comparación: Ventana del banco de filtros con retardo bajo modulado complejo (CMLDFB) con respecto a la ventana sinusoidal muestras 108 ES 2 631 906 T3 O σ ro σ o o ’c? o σ cü o c o o /) o o o c (ü X2 ó σ (ü c (ü +J c o o (ü FIG12 ω fO ω D E o CM r— 109 ES 2 631 906 T3 FIG13 prniugeiAi c '2 ’o Ό O E o o o o '3 LO O c '2 Ό (Λ O Q. O Q. ω o .c o o o r^. O C Ό o £ r*o 0 P*110 ES 2 631 906 T3 FIG 14A Codee Fuente de retardo retardo [muestras 1 retardo [ms] AAC-LD +SBR MDCT/IMDCT/ superposición QMF SBR doble frecuencia 960.2 577 384 40 12 8 2881 60 AAC-ELD LD-MDCT/LD-IMDCT/ superposición CLDFB SBR doble frecuencia 720.2 64 0 30 1 1504 31 . . FIG 14B retardo [muestras] retardo [ms] Retardo global [ms] LD-MDCT + LD-IMDCT 720*2 muestra 30 ms QMF 2*288= 576 muestras 12 ms 42 ms CLDFB 2*32= muestras 1.3 ms 31.3 ms 111 ES 2 631 906 T3 FIG 15A Respuesta de la frecuencia 800 112 ES 2 631 906 T3 FIG 15B Respuesta de la frecuencia 113 ES 2 631 906 T3 Comparación: Distintas funciones de ventana con retardo global de 127 Omega/pi 114 ES 2 631 906 T3 FIG16B Omega/pi 115 ES 2 631 906 T3 Comparación: Simétrico con respecto a prototipo con retardo global de 383 Omega/pi 116 ES 2 631 906 T3 O Q. Ξ oo O c o D_ 117 ES 2 631 906 T3 Señal temporal original - (castagnettes -s¡02) Muestras temporales - índice de muestreo 44100 kHz 118

Independent claims5

1,637 paragraphs in 49 sections, as filed

ES 2 631 906 T3

DESCRIPTION

Apparatus and procedure for the generation of audio subband values, apparatus and procedure for the generation of audio samples in the time domain Technical field

[0001] The embodiments of the present invention relate to an apparatus and a method for generating audio subband values, an apparatus and a method for generating audio samples in the time domain, and systems comprising any of the aforementioned apparatuses. above, which can be implemented for example in the field of current audio coding, audio decoding or other applications related to audio transmission.

[0002] Current digital audio processing is typically based on encoding plans that allow a significant reduction in bit rates, bandwidths, and memory space compared to direct transmission or storage of the corresponding audio data. .

This is achieved by encoding the audio data on the transmitter side and decoding the encoded data on the receiver side in advance, for example by providing the decoded audio data to a listener or subjecting it to further signal processing.

[0003] Such digital audio processing systems can be implemented with respect to a wide range of parameters that normally influence the quality of the transmitted audio data, or be processed in some other way, on the one hand, and the efficiency calculation, bandwidths and other performance-related parameters, on the other hand. Very often, obtaining higher qualities requires higher bit rates, greater computational complexity, and higher storage requirements for the corresponding encoded audio data. Therefore, depending on the intended application, with a suitable and achievable quality a compromise must be reached between factors such as allowed bit rates, acceptable computational complexity and acceptable amounts of data.

[0004] Another parameter, which is particularly important in real-time applications, such as bidirectional or monodirectional communication, the delay imposed by the different coding plans can also play an important role. As a consequence, the delay imposed by audio encoding and decoding places an additional restriction on the aforementioned parameters by balancing the needs and costs of different encoding plans that have a specific field of application. . Since such digital audio systems can be applied in many different fields of applications ranging from ultra-low quality transmission to high-end transmission, different parameters and different limitations are often imposed on the corresponding audio systems. In some applications, a higher delay may require, for example, a higher bit rate and therefore a higher bandwidth compared to an audio system with a higher delay, as a comparable level of quality.

[0005] However, in many cases, trade-offs may have to be made taking into account different parameters such as bit rate, computational complexity, memory requirements, quality and delay.

[0006] Enhanced MPEG-4 Low Delay AAC - Low Bitrate High Quality Communication, AES Convention Paper 6998, May 5, 2007 discloses low delay Advanced Audio Coding (AAC.) That reduces transfer speed demand of bits by 25-33%. This is achieved by using a delay-optimized version of the spectral band replication tool and using a dedicated low-delay filter bank featuring the same cosine modulation functions as traditional MDCT.

[0007] EP 1,199,711 A1 discloses encoding of the audio signal using bandwidth expansion. The broadband and narrowband acoustic band signals are encoded in such a way that the perceived sound quality of the corresponding reconstructed signals is improved. An enhancement estimation unit perceptually amplifies a reconstructed acoustic source using an enrichment spectrum. An asymmetric window function is arranged over a relevant entire frame and extends over at least a part of at least the previous frame.

[0008] The Use Of Asymmetric Windows For Reducing The Time Delay In Real-Time Spectral Analysis, D. Floréncio, ICRSP - 91, Volume 5, April 14, 1991 discloses the idea of presenting windows

Asymmetric ES 2 631 906 T3 as a way of reducing the delay time in spectral analysis. Specifically, the application to the encoding of LP speech and, particularly, CELP and LD-CELP is described.

Summary

[0009] An embodiment of an apparatus for generating complex value audio subband values is defined in claim 1. An embodiment of an apparatus for generating real value time domain audio samples is defined in claim 2.

The corresponding procedures are defined in claims 3 and 4.

Brief description of the drawings

[0011] In the following, embodiments of the present invention will be described with reference to the accompanying drawings.

Figure 1 is a schematic diagram of an embodiment of a device for generating audio subband values;

Figure 2a is a schematic diagram of an embodiment of a device for generating time-domain audio samples;

Figure 2b represents a functional principle according to an embodiment of the present invention in the form of a device intended to generate samples in the time domain;

Figure 3 represents the concept of window coefficient interpolation according to an embodiment of the present invention;

Figure 4 represents the interpolation of the window coefficients in the case of a sinusoidal window function;

Figure 5 is a schematic diagram of an embodiment of the present invention comprising an SBR decoder and an SBR encoder; Figure 6 represents the delay sources of an SBR system;

Figure 7a is a flow chart of one embodiment of a method for generating audio subband values;

figure 7b represents a stage of the embodiment of the method represented in figure 7a;

Figure 7c is a flow chart of an exemplary method for generating audio subband values;

Figure 8a is a flow chart of one embodiment of a method for generating samples in the time domain;

Figure 8b is a flow chart of an exemplary procedure for generating samples in the time domain;

Figure 8c is a flow chart of a further embodiment of a method for generating samples in the time domain;

Figure 8d is a flow chart of a further example of a procedure for generating samples in the time domain;

Figure 9a depicts a possible implementation of an embodiment of a method for generating audio subband values;

Figure 9b depicts a possible implementation of an embodiment of a method for generating audio subband values;

Figure 10a depicts a possible implementation of an embodiment of a method for generating time domain samples;

Figure 10b depicts a further possible implementation of an embodiment of a method for generating time domain samples;

Figure 11 depicts a comparison of a synthesis window function according to an embodiment of the present invention and a sinusoidal window function;

Figure 12 depicts a comparison of a synthesis window function according to an embodiment of the present invention and a SBR QMF filter function;

Figure 13 represents the different delays caused by the window function and the prototype filter function shown in Figure 12;

FIG. 14a is a table illustrating the various delay contributions of a conventional AAC-LD + SBR codec and an AAC-ELD codec comprising an embodiment of the present invention;

Figure 14b is a further table comprising details related to the delay of different elements of different codecs;

Fig. 15a depicts a comparison of a frequency response of a device that is based on a window function according to an embodiment of the present invention and a device that is based on a sinusoidal window function;

Figure 15b shows a close-up of the frequency response depicted in Figure 15a;

Figure 16a represents a comparison of the frequency response of 4 different window functions;

Figure 16b shows a close-up of the frequency responses depicted in Figure 16a;

Figure 17 depicts a comparison of the frequency response of two different window functions, an asymmetric window function and a window function that is a symmetric window function; and Figure 18 schematically represents the general temporal neutralization property of the human ear;

Fig. 19 depicts a comparison of an original audio time signal, a time signal generated based on the HEAAC codec, and a time signal based on a codec comprising an embodiment of the present invention.

Detailed description of the embodiments

Figures 1 to 19 represent functional diagrams and additional diagrams that describe the properties and functional characteristics of different embodiments of the devices and procedures for generating audio subband values, of devices and procedures for generating samples and domain systems temporary devices comprising at least one of the aforementioned devices or procedures. However, before describing a first embodiment of the present invention in more detail, it should be noted that the embodiments of the present invention can be implemented in hardware and software. Therefore, the implementations described using functional diagrams of hardware implementations of the corresponding embodiments can also be considered as flow diagrams of a suitable embodiment of a corresponding procedure. Furthermore, a flow chart describing an embodiment of the present invention can be considered a functional diagram of a corresponding hardware implementation.

[0013] Next, filter bank implementations will be described, which can be implemented as an analysis filter bank or a synthesis filter bank. An analysis filter bank is a device intended to generate audio subband values in audio subband channels based on audio samples in the time domain (input) that are in a time sequence that extends from an initial sample to a late sample. That is, the term analysis filter bank can be used synonymously for an embodiment of the present invention in the form of a device for generating audio subband values. Accordingly, a synthesis filter bank is a filter bank for generating time domain audio samples from audio subband values in audio subband channels. That is, the term synthesis filter bank can be used synonymously for an embodiment according to the present invention in the form of a device intended to generate audio samples in the time domain.

ES 2 631 906 T3

[0014] Both an analysis filter bank and a synthesis filter bank, collectively also referred to as filter banks, can be implemented, for example, as modulated filter banks. Modulated filter banks, the examples of which will be described in more detail below, are based on oscillations having frequencies based on, or derived from, center frequencies of the subbands in the frequency domain. The term "modulated" in this context refers to the fact that the aforementioned oscillations are used in conjunction with a window function or a prototype filter function, depending on the specific implementation of said modulated filter bank. Modulated filter banks can, in principle, be based on real value oscillations such as harmonic oscillation (sinusoidal oscillation or cosine oscillation) or corresponding complex value oscillations (complex exponential oscillations). Therefore, modulated filter banks are called real modulated filter banks or modulated filter banks with complex filters, respectively.

Embodiments of the present invention in the form of low-delay complex modulated filter banks and low-delay real modulated filter banks and the corresponding procedures and software implementations will be described in more detail in the following description. One of the main applications of such low-delay modulated filter banks is an integration into a low-delay spectral band (SBR) replication system, which is currently based on the use of a complex QMF filter with a symmetric prototype filter ( QMF = quadrature mirror filter).

[0016] As will become apparent within the framework of the present description, an implementation of low-delay filter banks according to embodiments of the present invention will provide the advantage of an improved balance between delay, frequency response, dispersion. temporal noise and reconstruction quality.

The aforementioned improved compensation, in particular between delay and reconstruction quality, is based on a method of using so-called zero delay techniques in order to extend the filter impulse response of the corresponding filter banks without introducing an additional delay. Lower delay can be achieved with a predefined quality level, better quality with a predefined delay level, or simultaneous improvement of both delay and quality by using an analysis filter bank or a synthesis filter bank according to a shape. carrying out the present invention.

[0018] The embodiments of the present invention are based on the discovery that these improvements can be achieved using a new window function for either of the two filter banks described above. That is, the quality and / or delay in the case of an analysis filter bank can be improved by using an analysis window function comprising a sequence of window coefficients, comprising a first group comprising a first consecutive part of the sequence of window coefficients and the second group of window coefficients comprising a second consecutive part of the sequence of window coefficients. The first part and the second part comprise all the window coefficients of the window function. Furthermore, the first part comprises fewer window coefficients than the second part but an overall energy value of the window coefficients in the first part is higher than an overall energy value of the window coefficients of the second part. The first group of window coefficients is used to display windows of late time domain samples and the second group of window coefficients is used for early time domain samples displayed in windows. This form of the window function provides the opportunity to process time domain samples with window coefficients that have higher energy values earlier. This is a result of the described distribution of window coefficients for the two parts and its application to the sequence of audio samples in the time domain. As a consequence, the use of such a window function can reduce the delay introduced by the filter bank to a constant quality level or allow an improved quality level based on a constant delay level.

Accordingly, in the case of an embodiment of the present invention in the form of a device for generating audio samples in the time domain and a corresponding method, a synthesis window uses a synthesis window generator function comprising a sequence of window coefficients ordered correspondingly into a first (consecutive) portion and a second (consecutive) portion. Also, in the case of a synthesis window function, an overall energy value of a window coefficient in the first part is greater than an overall energy value of a window coefficient in a second part, the first part comprising less window coefficients than the second part. Due to said distribution of the window coefficients between the two parts as the synthesis windowing generator uses the first part of windowing coefficients to display subsequent time domain sample windows and the second part of window coefficients to display Previous time domain windows, the effects and benefits described above can also be applied to a filter bank

ES 2 631 906 T3 of synthesis or a corresponding embodiment of a process.

The detailed descriptions of the synthesis window functions and the analysis window functions used within the framework of the embodiments of the present invention will be described in more detail below. In many embodiments of the present invention, the sequence of window coefficients of the synthesis window function and / or the analysis window function comprises exactly the first group and the second group of window coefficients. Furthermore, each of the window coefficients of the sequence of window coefficients belongs to exactly one of the first group and the second group of window coefficients.

Each of the two groups comprises exactly one part of the sequence of window coefficients in a consecutive manner. In the present description, a part comprises a consecutive set of window coefficients according to the sequence of the window coefficients. In the embodiments according to the present invention, each of the two groups (first and second group) comprises exactly one part of the sequence of window coefficients as explained above. The groups of corresponding window coefficients do not comprise any window coefficients that do not belong to the exact part of the corresponding group. That is, in many embodiments of the present invention, each of the first and second group of window coefficients comprises only the first part and the second part of window coefficients without comprising other window coefficients.

[0022] Within the framework of the present description, a consecutive part of the sequence of window coefficients is to be understood as a connected set of window coefficients in the mathematical sense, in which the set is not devoid of window coefficients in comparison with the sequence of window coefficients, which would be in an interval (eg index range) of the window coefficients of the corresponding part. As a consequence, in many embodiments of the present invention, the sequence of window coefficients is divided exactly into two connected parts of window coefficients, each forming the first or second groups of window coefficients. In these cases, each window coefficient comprised in the first group of window coefficients is arranged before or after each of the window coefficients of the second group of window coefficients with respect to the overall sequence of window coefficients.

That is, in many embodiments according to the present invention the sequence of window coefficients is divided exactly into two groups or parts without leaving any window coefficient. According to the sequence of window coefficients, which also represents an order thereof, each of the two groups or parts comprises all the window coefficients up to (but excluding) or starting from (including) an edge window coefficient . As an example of this, the first part or group may comprise window coefficients that have indices between 0 and 95 and between 96 and 639 in the case of a window function that includes 640 window coefficients (that have indices between between 0 and 639). In this case, the edge window coefficient would be the one corresponding to the index 96. Of course, other examples are also possible (for example, between 0 and 543 and between 544 and 639).

The detailed example of implementation of an analysis filter bank that will be described below provides a filter length that covers 10 blocks of input samples while causing a system delay of only 2 blocks, which is the corresponding delay introduced by an MDCT (Modified Discrete Cosine Transform) or an MDST (Modified Discrete Sine Transform). One difference is because the longer filter length covers 10 input sample blocks compared to an MDCT or MDST implementation where the overlap increases from 1 block in the case of MDCT and MDST to an overlap of 9 blocks . However, other implementations can also be made that cover a different number of input sample blocks, which are also called audio input samples. Furthermore, other advantages and disadvantages can be considered and applied as well.

[0025] Figure 1 represents a functional diagram of an analysis filter bank 100 as an embodiment of an apparatus for generating audio subband values in audio subband channels. The analysis filter bank 100 comprises an analysis window generator 110 for displaying windows of a frame 120 of time domain audio input samples. Frame 120 comprises T blocks 130-1, ..., 130-T blocks of audio samples in the time domain (input), in which T is a positive integer and equal to 10 in the case of the form of embodiment shown in FIG. 1. However, frame 120 may also comprise a different number of blocks 130.

Both frame 120 and each of blocks 130 comprise time domain audio input samples in a time sequence extending from an early sample to a late sample along a timeline as indicated by arrow 140 in figure 1. That is, in the illustration as shown in figure 1, the further to the right the sample of

ES 2 631 906 T3 time domain audio, which in this case is also a time domain audio input sample, the later is the corresponding time domain audio sample with respect to the time domain audio sample sequence .

[0027] The analysis windowing generator 110 generates, based on the sequence of time domain audio samples, windowed samples in the time domain, which are arranged in a frame 150 of windowed samples. According to the frame 120 of the audio input samples in the time domain, also the frame of the windowed samples 150 comprises blocks T of the windowed samples 160-1, ..., 160-T. In preferred embodiments of the present invention, each of the windowed sample blocks 160 comprises the same number of windowed samples as the number of audio input samples in the time domain of each input sample block 130. audio in the time domain. Thus, when each of the blocks 130 comprises N time domain input audio samples, the frame 120 and the frame 150 each comprise the T N samples. In this case, N is a positive integer, which can, for example, have the values 32 or 64. In the case T = 10, each of the frames 120, 150 comprise 320 and 640, respectively, in the previous case.

The analysis window 110 is coupled to a calculator 170 for calculating the audio subband values based on the windowed samples provided by the analysis window generator 110. The audio subband values are provided by the calculator 170 as a block 180 of audio subband values, each of the audio subband values corresponding to an audio subband channel. In a preferred embodiment, also the audio subband value block 180 comprises N subband values.

Each of the audio subband channels corresponds to a characteristic center frequency. The center frequencies of the various audio subband channels can be found, for example, evenly distributed or evenly spaced with respect to the frequency bandwidth of the corresponding audio signal as described in the audio input samples of time domain provided to analysis filter bank 100.

The analysis window generator 110 is adapted to display time-domain audio input sample windows of frame 120 based on an analysis window function comprising a sequence of window coefficients to obtain the windowed samples. plot 150. The analysis window 110 is adapted to display windows of the time domain audio sample frame 120 by multiplying the values of the time domain audio samples with the window coefficients of the analysis window function. That is, the windowing comprises an element multiplication of the time domain audio samples with the corresponding window coefficient. Since both the time domain audio sample frame 120 and the window coefficients comprise a corresponding sequence, the element multiplication of the time domain audio samples and window coefficients is performed according to the corresponding sequences. , for example, indicated by a sample coefficient index and window.

[0031] In some embodiments of the present invention in the form of a bank of analysis filters 100, as represented in Figure 1, the analysis window function, as well as the synthesis window function in the case of a synthesis filter bank, it comprises only real-value window coefficients. That is, each of the window coefficients attributed to a window coefficient index is a real value.

[0032] The window coefficients together form the corresponding window function and an example thereof is represented in FIG. 1 as analysis window function 190. As indicated above, the sequence of window coefficients that form the analysis window function 190 comprises a first group 200 and a second group 210 of window coefficients. The first group 200 comprises a first consecutive and connected part of the window coefficients of the sequence of window coefficients, while the second group 210 comprises a second consecutive and connected part of a window coefficient. Together with the first part of the first group 200, they form the complete sequence of window coefficients of the analysis window function 190. Furthermore, each window coefficient of the sequence of window coefficients belongs to either the first part or the second part of window coefficients, such that the analysis window function 190 is composed of the window coefficient of the first part and the second part. The first part of window coefficients is therefore identical to the first group 200 of window coefficients and the second part is identical to the second group 210 of window coefficients, as indicated by the corresponding arrows 200, 210 in the Figure 1.

The number of window coefficients in the first group 200 of the first part of window coefficients is less than the number of window coefficients in the second group of the second part of window coefficients. However, an overall energy value of the window coefficients of the first

ES 2 631 906 T3 group 200 is greater than an overall energy value of the window coefficients of the second group 210. As will be described later, an energy value of a set of window coefficients is based on a sum of the squares of the absolute values of the corresponding window coefficients.

In embodiments according to the present invention, the analysis window function 190 as well as a corresponding synthesis window function is therefore asymmetric with respect to the sequence of window coefficients or to an index of a window coefficient. Based on a set of definitions of indexes of window coefficients on which the analysis window function 190 is defined, the analysis window function 190 is asymmetric, when for all real numbers there is no additional real number, so so that the absolute value of the window coefficient corresponding to the window (no - n) is not equal to the absolute value of the window coefficient corresponding to the window coefficient index (no + n), when (n0 - n) and (n0 + n) belong to the first set of definitions.

[0035] Furthermore, as also schematically represented in FIG. 1, the analysis window function 190 comprises sign changes in which the product of two consecutive window coefficients is negative. More details and other features of possible window functions according to embodiments of the present invention will be described in more detail with reference to Figures 11 to 19.

As noted above, the window sample frame 150 comprises a similar block structure with individual blocks 160-1, ..., 160-T as frame 120 of individual time domain input samples. Since the analysis windowing generator 110 is capable of windowing the audio input samples in the time domain by multiplying those values with the window coefficients of the analysis window function 190, the windowed sample frame 150 is it is also in the temporal domain. The calculator 170 calculates the audio subband values or, more accurately, the audio subband value block 180 using the windowed sample frame 150 and performs a transfer from the time domain to the frequency domain. Thus, the calculator 170 can be thought of as a time / frequency converter, which can provide the block 180 of audio subband values as a spectral representation of the windowed sample frame 150.

[0037] Each audio subband value of block 180 corresponds to a subband having a characteristic frequency. The number of audio subband values comprised in block 180 is also sometimes referred to as the band number.

In many embodiments according to the present invention, the number of audio subband values in block 180 is identical to the number of audio input samples in the time domain of each of the blocks 130 of frame 120 . In the event that the frame 150 of windowed samples comprises the same block structure as the frame 120 such that each of the blocks 160 of windowed samples also comprises the same number of windowed samples as the block of said windows. audio input samples in time domain 130, block 180 of audio subband values naturally also comprises the same number as block 160.

[0039] Optionally, frame 120 can be generated based on a block of recent time domain audio input samples 220 by shifting blocks 130-1, ..., 130- (T-1) one block in the opposite direction to the arrow 140 that indicates the direction of time. In this way, A time domain audio input sample frame 120 is generated which is processed by shifting the most recent blocks (T-1) of a directly preceding frame 120 of one-block time domain audio samples toward the domain audio samples previous time domain and adding the recent block 220 of recent audio samples in the time domain as the new block 130-1 comprising the last time domain audio samples of the present frame 120. In Figure 1 this is also indicated by a series of broken arrows 230 indicating the displacement of blocks 1301, ..., 130 - (T-1) in the opposite direction to arrow 140.

Due to this displacement of the blocks 130 in the opposite direction of time, as indicated by the arrow 140, a current frame 120 to be processed comprises the block 130- (T-1) of the directly preceding frame 120 as new block 130 -T. Therefore, the blocks 130- (T-1), ..., 130-2 of the present frame 120 to be processed are equal to the block 130- (T-2), ..., 130-1 of the frame directly previous 120. Block 130-T of frame 120 directly above is discarded.

As a consequence, each time domain audio sample of the new block 220 will be processed T times in the frame of T consecutive processing of T consecutive frames 120 of time domain audio input samples. Therefore, each recent block time domain audio input sample 220 contributes not only to T distinct frames 120, but

ES 2 631 906 T3 likewise to T distinct frames 150 of windowed samples and T 180 blocks of audio subband values. As indicated above, in a preferred embodiment according to the present invention, the number of T blocks in frame 120 equals 10, such that each time domain audio sample provided to the analysis filter bank 100 contributes to 10 distinct blocks 180 of audio subband values.

[0042] Initially, before a single frame 120 is processed by the analysis filter bank 100, the frame 120 may be initialized to a small absolute value (lower than a predetermined threshold value), for example the value 0. As will be described in more detail below, the shape of the analysis window function 190 comprises a center point or a center of gravity, which typically corresponds to, or lies between, two window coefficient indices of the first group 200.

As a consequence, the number of recent blocks 220 to be inserted into frame 120 is small, before frame 120 is filled at least to a point such that the parts of frame 120 are occupied by non-missing values (that is, non-zero values) that correspond to window coefficients that make a significant contribution to their energy values. Typically, the number of blocks to enter into frame 120 before significant processing begins is between 2 and 4 blocks depending on the shape of the analysis window function 190. Therefore, the analysis filter bank 100 can provide blocks 180 faster than a corresponding filter bank employing, for example, a symmetric window function. As the recent blocks 220 are normally provided to the analysis filter bank 100 as a whole, each of the recent blocks corresponds to a recording or sampling time, which is provided substantially by the length of the block 220 (i.e., the number of audio input samples in the time domain comprised in block 220) and the sampling rate or sampling frequency. Therefore, the analysis window function 190, incorporated in an embodiment of the present invention, provides a reduced delay before the first and subsequent blocks 180 of audio subband values can be provided or output by the bank. of filters 100.

[0044] As an additional option, the device 100 may be capable of generating a signal or incorporating information relative to the analysis window function 190 used in the generation of frame 180 or regarding a synthesis window function that is will be used within the framework of a synthesis filter bank. Thus, the analysis filter function 190 may, for example, be a time reversal or index version of the synthesis window function to be used by the synthesis filter bank s.

[0045] Figure 2a is a schematic diagram of an embodiment of a device 300 for generating time domain audio samples based on the block of audio subband values. As described above, an embodiment of the present invention in the form of apparatus 300 for generating audio samples in the time domain is also often referred to as a synthesis filter bank 300 when the device can generate audio samples in the time domain. time domain, which in principle can be reproduced, based on audio subband values comprising spectral information relative to an audio signal. Thus, synthesis filter bank 300 can synthesize time domain audio samples based on audio subband values, which can be generated, for example, by a corresponding analysis filter bank 100.

[0046] Figure 2a represents a functional diagram of the synthesis filter bank 300 comprising a calculator 310 to which a block 320 of audio subband values (in the frequency domain) is provided. The calculator 310 can calculate a frame 330 comprising a sequence of time domain intermediate samples from the audio subband values in block 320. The time domain intermediate sample frame 330 comprises in many embodiments according to the present invention also a similar block structure such as for example the window sample frame 150 of the analysis filter bank 100 of FIG. 1.

In these cases, frame 330 comprises blocks 340-1, ..., 340-T of intermediate samples of the time domain.

The sequence of intermediate samples in the time domain of frame 330, as well as each block 340 of intermediate samples in the time domain, comprises an order according to the time indicated by an arrow 350 in FIG. 2a. As a consequence, frame 330 comprises an early time domain sample in block 340-T and a late time domain sample in block 340-1, representing the first and last intermediate time domain samples for frame 330. , respectively. Also, each of the blocks 340 comprises a similar order. Consequently, in synthesis filter bank embodiments, the terms frame and sequence can often be used synonymously.

ES 2 631 906 T3

[0049] The calculator 310 is coupled to a synthesis window generator 360 in which the frame 330 of intermediate samples of the time domain is arranged. The synthesis window generator is capable of windowing the time domain intermediate sample sequence using a synthesis window function 370 schematically depicted in FIG. 2a. As an output, the synthesis windowing generator 360 provides a frame 380 of intermediate samples of the windowed time domain, which may also comprise a block structure of blocks 390-1, ..., 390-T.

Frames 330 and 380 may comprise T blocks 340 and 390, respectively, where T is a positive integer. In a preferred embodiment according to the present invention in the form of a synthesis filter bank 300, the number of blocks T is equal to 10. However, in different embodiments, different numbers of blocks may be found in a of the frames. More precisely, in principle the number of blocks T may be greater than or equal to 3, or greater than or equal to 4, depending on the circumstances of the implementation and the embodiments described above for the embodiments according to the present invention comprising a frame block structure for both the synthesis filter bank 100 and the synthesis filter bank 300.

The synthesis windowing generator 360 is coupled to an output stage of an overlay adder 400, to which the windowed time domain intermediate sample frame 380 is provided. The output phase of an overlay adder 400 may process the intermediate time domain samples into windows to obtain a block 410 of time domain samples. In this way, block 410 of the time domain samples (output) can be provided, for example, to other elements for further processing, storage or transformation into audible audio signals.

[0052] The calculator 310 for calculating the sequence of time domain samples comprised in frame 330 can transfer data from the frequency domain to the time domain. Thus, calculator 310 may comprise a frequency / time converter capable of generating a time domain signal from the spectral representation comprised in block 320 of audio subband values. As explained in connection with the calculator 170 of the analysis filter bank 100 shown in FIG. 1, each of the audio subband values in block 320 corresponds to an audio subband channel having a characteristic center frequency.

[0053] On the other hand, the intermediate samples of the time domain comprised in frame 330 represent in principle information in the time domain. The synthesis window generator 360 is capable and capable of providing windows of the sequence of intermediate samples of the time domain comprised in frame 330 using a synthesis window function 370 shown schematically in FIG. 2a. The synthesis window function 370 comprises a sequence of window coefficients also comprising a first group 420 and a second group 430 of window coefficients, as explained above in relation to the window function 190 with a first group 200 and a second group 210 of window coefficients.

The first group 420 of window coefficients of the synthesis window function 370 comprises a first consecutive part of the sequence of window coefficients. Similarly, the second group 430 of coefficients comprises a second consecutive part of the sequence of window coefficients, in which the first part comprises fewer window coefficients than the second part and in which an overall energy value of the window coefficients in the first part is greater than the corresponding global energy value of the window coefficients in the second part. Other characteristics and properties of the synthesis window function 370 may be similar to the corresponding characteristics and properties of the analysis window function 190, as schematically represented in FIG. 1. As a consequence, reference is made to the corresponding description in the framework of the analysis window function 190 and the further description of the window functions with respect to Figures 11 to 19, with the first group 200 corresponding to the first group 420 and the second group 210 corresponding to the second group 430.

[0055] For example, the parts comprised in the two groups 420, 430 of window coefficients normally each constitute a consecutive and connected set of window coefficients comprising each other all of the window coefficients of the sequence of window coefficients of the window function 370. In many embodiments according to the present invention, the analysis window function 190 as depicted in FIG. 1 and the synthesis window function 370 as depicted in FIG. 2a are based on each other. For example, the analysis window function 190 may be a time-reversed or index-reversed version of the synthesis window function 370. However, other relationships between the two window functions 190, 370 may also be possible. It may be advisable to use a synthesis window function 370 within the framework of the synthesis window generator 360, which is related to the window function of analysis 190, to be used during generation (optionally prior to further modifications) of block 320 of audio subband values provided to synthesis filter bank 300.

ES 2 631 906 T3

[0056] As indicated in connection with FIG. 1, the synthesis filter bank 300 of FIG. 2a can be optionally adapted such that the input block 320 can comprise additional signals or additional pieces of information relating to the window functions. By way of example, block 320 may comprise information related to the analysis window function 190 used to generate block 320 or regarding the synthesis window function 370 to be used by the synthesis window generator 360. For example, therefore, the filter bank 300 can be accepted to isolate the corresponding information and to provide the same to the synthesis window generator 360.

The output phase of an overlay adder 400 can generate block 410 of time domain samples by processing intermediate time domain samples in windows comprised in frame 380. In various embodiments according to the present invention, the output phase of an overlay adder 4000 may comprise a memory for temporarily storing frames 380 previously received from intermediate samples of the time domain in windows. Depending on the implementation details, the output phase of an overlay adder 400 may comprise, for example, T different storage locations comprised in memory for storing a total number of T 380 frames of windowed time domain intermediate samples. However, also a different number of storage locations may be comprised in the output phase of an overlay adder 400 if appropriate. Furthermore, in various embodiments according to the present invention, the output stage of an overlay adder 400 may be capable of providing block 410 of time domain samples based solely on a single frame 380 of intermediate time domain samples. The embodiments of the various synthesis filter banks 300 will be described in more detail below.

Figure 2b represents a functional principle according to an embodiment of the present invention in the form of a synthesis filter bank 300. The block 320 of audio subband values is first transferred from the frequency domain to the domain. time using calculator 310, which is represented in Figure 2b by arrow 440. The frame 320 resulting from intermediate samples of the time domain comprising blocks 340-1, ..., 340-T of intermediate samples in the time domain is thus performed in windows by means of the synthesis window generator 360 (not represented in figure 2b) multiplying the sequence of intermediate samples of the time domain of frame 320 by the sequence of window coefficients of the synthesis window function 370 to obtain the frame 380 of intermediate samples of the windowed time domain.

The frame 380 again comprises blocks 390-1, ..., 390-T of intermediate samples of the time domain in windows, which together form the frame 380 of intermediate samples of the time domain in windows.

In the embodiment represented in Figure 2b of a synthesis filter bank of the present invention 300, the output phase of an overlay adder 400 can thus generate block 410 of output samples from the domain. temporal by adding for each index value of the audio samples of the time domain of block 410, the intermediate samples of the time domain of a block 390 of different frames 380. As shown in Figure 2b, the time domain audio samples of block 410 are obtained by adding for each audio sample index an intermediate sample of the window time domain of block 390-1 of frame 380, they are processed by the synthesis window generator 360 in the current cycle and as described above, the corresponding time domain intermediate sample from the second block 390-2 of a frame 380-1 is processed immediately before frame 380 and stored in a storage location in the output phase of an overlay adder 400. As shown in Figure 2b, the corresponding time domain intermediate samples of additional blocks 390 can be used (for example, block 390-3 of frame 380-2, block 390-4 of frame 380-3 , block 390-5 of frame 380-4) processed by synthesis filter bank 300. Frames 380-2, 380-3, 380-4, and optionally other frames 380, have been processed by bank of synthesis filters 300 in previous cycles. Frame 380-2 was processed immediately before frame 380-1, and therefore frame 380-3 was generated immediately before frame 380-2, and so on.

[0060] The output phase of an overlay addition 400, as used in the embodiment, may sum for each index of block 410 of samples of the time domain (output) T different blocks 390-1, ... , 390-T of intermediate samples of the time domain in windows from T different frames 380, 380-1, ..., 380- (T-1). Therefore, apart from the first T processed blocks, each of the time domain samples (output) of block 410 is based on T blocks 320 different from audio subband values.

As in the case of the embodiment of the present invention, the analysis filter bank 100 described in Figure 1, due to the shape of the synthesis window function 370, the synthesis filter bank 300 offers the ability to quickly provide block 410 of time domain (output) samples. This is also a consequence of the shape of the window function 370.

ES 2 631 906 T3

Since the first group 420 of window coefficients corresponds to a higher energy value and comprises fewer window coefficients than the second group 430, the synthesis window generator 360 can provide significant frames 380 of windowed samples when the window is filled. frame 330 of intermediate time domain samples such that at least the window coefficients from the first group 420 contribute to frame 380. The window coefficients of the second group 430 present a lower contribution due to their lower energy value.

Therefore, when starting, the synthesis filter bank 300 is initialized to 0, the supply of blocks 410 can, in principle, be started when only a few blocks 320 of audio subband values have been received by synthesis filter bank 300. Therefore, also the synthesis filter bank 300 allows a significant delay reduction compared to the synthesis filter bank exhibiting, for example, a symmetric synthesis window function.

[0063] As indicated above, the calculators 170 and 310 of the examples shown in Figures 1 and 2a can be implemented as real-value calculators that generate or can process real audio subband values from blocks 180 and 320 , respectively. In these cases, the calculators can, for example, be implemented as real-value calculators based on harmonic oscillating functions such as the sinusoidal function or the cosine function. In some embodiments, complex value calculators are implemented such as calculators 170, 310. In these cases, the calculators can be implemented for example based on complex exponential functions or other harmonic complex value functions. The frequency of real value or complex value swings typically depends on the index of the audio subband value, which is sometimes also called the band index or subband index of the specific subband. Furthermore, the frequency can be identical or depend on the center frequency of the corresponding sub-band. For example, the frequency of the oscillation can be multiplied by a constant factor, offset from the center frequency of the corresponding subband or it can depend on a combination of both modifications.

[0064] A complex value calculator 170, 310 based on real value calculators can be constructed or implemented. For example, for a complex value calculator, an efficient implementation can be used in principle for both the cosine and sine modulated part of a filter bank representing both the real and the imaginary part of a complex value element. This means that it is possible to implement both the cosine modulated part and the sine modulated part based, for example, on the modified structures of DCT-IV and DST-IV. Furthermore, other implementations could employ the use of an FFT (FFT = fast Fourier transform) optionally implemented together for both the real part and the complex modulation part of the calculators using an FFT or instead using a separate stage of the FFT. for each transform.

Mathematical description

[0065] In the following sections an example of the embodiments of an analysis filter bank and synthesis filter bank will be described with a plurality of 8-block overlaps with the part, which do not cause additional delays, such as explained above, and one block to the future, which causes the same delay as an MDCT / MDST structure (MDCT = modified discrete cosine transform, MDST = modified discrete sine transform). That is, in the following example, the parameter T equals 10.

[0066] First, a description will be made of a complex modulated low delay analysis filter bank. As depicted in FIG. 1, the analysis filter bank 100 comprises the steps of presenting an analysis window performed by the analysis window generator 110 and a modulation of the analysis performed by the calculator 170. The analysis window is based on the equation zin = w (lQN - 1 - n) xin for 0 <n <10 · N, (1) where z¡, n is the windowed sample (real value) corresponding to block index i and sample index n of frame 150 represented in figure 1.

The value x, n is the time input sample (real value) corresponding to the same block index i and sample index n. The analysis window function 190 is represented in equation (1) by its real value window coefficients w (n), where n is also the window coefficient index in the range indicated in equation (1). As already described above, the parameter N is the number of

ES 2 631 906 T3 samples in a block 220, 130, 160, 180.

[0067] From the arguments of the analysis window function w (10N-1-n) it can be seen that the analysis window function represents an inverse version or an inverted version in time of the window function of synthesis, which is actually represented by the coefficient w (n).

The modulation of the analysis carried out by means of the calculator 170 in the embodiment represented in figure 1, is based on the two equations

Y

2 / Vl + 87V siní ~ (n + nn / k + - 2 (3;

ii = -8W V JY V 2.

so that the index of the spectral coefficient or band index k being an integer in the range of

The values XReal, i, k and X¡mag, ¡, x represent the real part and the imaginary part of the audio subband value of the complex value corresponding to the index of block i and to the index of the spectral coefficient k of block 180. The parameter n0 represents an index option, which is equal to n0 =-77/2 + 0.5 . (5)

The corresponding complex modulated low delay synthesis filter bank comprises the transformation steps of a synthesis modulation, a synthesis window and an overlay addition as will be described later.

The synthesis modulation is based on the equation

“+ N ^ k + sin ^ (n + £ n <10 · N. (6) in which x'¡, n is an intermediate sample of the time domain of frame 330 corresponding to the sample index ny to the block index i. Again, the parameter N is an integer that indicates the length of the block 320, 340, 390, 410, which is also called the transform block length or, due to the block structure of the frames 330, 380, move to the previous block. Other variables and additional parameters have also been introduced, such as the index of the spectral coefficient k and the displacement n0

The synthesis window performed by the synthesis window generator 360 in the embodiment represented in Figure 2a is based on the equation z 'yes = w (n) x 'iin for 0 <n <10N, (7) where z'i,n is the value of the intermediate sample of the window time domain that corresponds to the sample index ny to the block index i of frame 380.

ES 2 631 906 T3 (8)

[0073] The mark of the transform of the superposition addition is based on the equation

+ Ζ'ί-6, η + ίΎ+ Ζ'/ -7, Β + 7 //+ Ζ,/ -8, Λ + 8ν+ Ζ'/ -9, Λ + 9 ^, for 0 £ n <N where out¡, n represents the time domain sample (output) corresponding to sample index n and block index i. Thus, equation (8) represents the overlay addition operation as performed in the output phase of an overlay adder 400 as depicted at the bottom of FIG. 2b.

[0074] By way of comparison, for example, equations (2) and (6) with regard to the cosine part, the cosine contribution of the analysis modulation and the synthesis modulation present a comparable structure when it is considered that of an MDCT. Although the design procedure allows in principle an extension of the MDCT in both directions with respect to time, in this case only a backward extension of the E (= T-2) blocks is applied, in which each of the T blocks comprise N samples.

The coefficient of the frequency Xi, k of band k and block i in an N-channel or N-band analysis filter bank can be summarized by the

(n + | -

(9) index of the spectral coefficient k as defined in equation (4).

Again, n is a sample index and wa is the analysis window function.

[0075] For the sake of completeness, the mathematical description provided above about the complex modulated slow slow delay analysis filter bank can be presented in the same summarized way as equation (9), exchanging the cosine function for the exponential function complex value. More accurately, with the definition and variables provided above, equations (1), (2), (3), and (5) can be summarized and expanded according to

X'¡.k = -2 Σ j

2N-I

J — fn + WXk + i ») (10) in which, unlike equations (2) and (3), the extension of 8 blocks backwards has been replaced by the variable E (= 8).

The stages of synthesis modulation and synthesis window display, as described in the case of complexes in equations (6) and (7), can be summarized in the case of a bank of real value synthesis filters. The frame 380 of intermediate samples of the window time domain, also referred to as a demodulated vector, is given by

Z i, n

(11) in which Z'i, n is the intermediate sample of the time domain in windows corresponding to the band index i and the sample index n. The sample index n is again an integer in the interval of y

ES 2 631 906 T3 <η <N (2 + E) = N · T

Ws (n) is the synthesis window, which is compatible with the analysis window wto (n) from equation (9). Thus, the stage of the superposition transform is given by

J = (E + 1) (13) where x 'i,n is the reconstructed signal, or rather a sample of the time domain of block 410 as supplied by the output phase of an overlay adder 400 depicted in FIG. 2a.

In the case of the complex value synthesis filter bank 300, equations (6) and (7) can be summarized and generalized with respect to the extension of E (= 8) blocks to the trajectory according to

(14)

it is also valid for the case of complex values.

As the direct comparison of equation (14) with equation (7) shows, the window function W (n) of equation (7) is the same function of the synthesis window as Ws (n) from equation (14). As indicated above, the similar comparison of equation (10) with the function coefficient of the analysis window wto (n) with equation (1) shows that the analysis window function is the time-reversed version of the synthesis window function in the case of equation (1).

Like both, an analysis filter bank 100 as represented in Figure 1 and a synthesis filter bank 300 as represented in Figure 2a represent a significant improvement in terms of a trade-off between On the one hand the delay and on the other the quality of the audio processing, the filter banks 100, 300 are often called low delay filter banks. The complex value version of it is sometimes called a low delay complex filter bank, abbreviated CLDFB.

As has been shown in the previous analysis of the mathematical foundations, the framework used to implement the proposed low-delay filter banks uses a structure of the MDCT or IMDCT type (IMDCT = inverse MDCT), known from the standard MPEG-4, using an extended overlay. Additional regions of overlap can be attached en bloc to the left as well as the right of the MDCT-like core. In this case, only the extension to the right is used (for the synthesis filter bank), which works only from previous samples and therefore does not cause any additional delay.

Observing equations (1), (2) and (14) it has been shown that the processing is very similar to that of MDCT or IMDCT. By only slight modifications comprising a modified analysis window function and a synthesis window function, respectively, the MDCT or IMDCT is extended to a modulated filter bank that can handle multiple overlays and is very flexible with respect to its delay. For example, with equations (2) and (3) it has been shown that the complex version is obtained in principle simply by adding a sinusoidal modulation to the given cosine modulation.

Interpolation

[0083] As indicated in relation to Figures 1 and 2a, both the analysis window generator

ES 2 631 906 T3

110 such as the synthesis window generator 360 or the corresponding filter banks 100, 300 are able to display windows of the corresponding frames of time domain samples by multiplying each of the corresponding time domain audio samples by an individual window coefficient. That is, each of the time domain samples is multiplied by a (individual) window coefficient, as for example shown with equations (1), (7), (9), (10), (11 ) and (14). As a consequence, the number of window coefficients of the corresponding window function is normally identical to the number of audio samples in the corresponding time domain.

[0084] However, under certain implementation circumstances, it may be advisable to implement a window function that has a second larger number of window coefficients compared to the actual window function that has a smaller first number of coefficients, which is actually used during window display of the corresponding frame or sequence of audio samples in the time domain. This may be, for example, advisable where the memory requirements of a specific implementation may be more valuable than the computational efficiency. Another scenario in which a sampling of the window coefficients could be useful is the so-called dual method, which is used, for example, in the framework of SBR systems (SBR = spectral band replication). The SBR concept will be described in more detail with reference to Figures 5 and 6.

In this case, the analysis window generator 110 or the synthesis window generator 360 can be further adapted such that the corresponding window function used to display windows of the time domain audio samples provided to the generator of corresponding windows 110, 360 It is obtained by interpolating the window coefficients of the largest window function that has a second largest number of window coefficients.

The interpolation can be performed, for example, by linear, polynomial or spline interpolation. For example, in the case of linear interpolation, but also in the case of an interpolation based on a polynomial or spline, the corresponding window generator 100, 360 may be able to interpolate the window coefficients of the window function used to display windows based on two consecutive window coefficients of the larger window function according to a sequence of the window coefficients of the larger window function to obtain a coefficient of window function window.

[0087] Especially in the case of an even number of audio samples in the time domain and window coefficients, an implementation of an interpolation as described above, a significant improvement of the audio quality can be obtained. For example, in the case of an NT even number of time domain audio samples in one of the 120,330 frames, which does not use an interpolation, for example linear interpolation, will cause severe aliasing effects during post-processing of the corresponding audio samples in the time domain.

[0088] Figure 3 is an example of a linear interpolation based on a window function (an analysis window function or a synthesis window function) to be used in relation to frames comprising audio samples in the time domain NT / 2. Due to memory limitations or other implementation details, the window coefficients of the window function itself are not stored in a memory, but rather a larger window function comprising NT window coefficients that is stored in the appropriate memory or is are available in some other way. Figure 3 represents in the upper graph, the corresponding window coefficients c (n) as a function of the indices of window coefficients n in the interval between 0 and N T-1.

[0089] Based on a linear interpolation of two consecutive window coefficients of the window function that have the highest number of window coefficients, as represented in the upper graph of Figure 3, an interpolated window function is calculated based on in the equation ci [n] = | (c [2n] + c [2n + l]) for Q <n <NT / 2 (15)

The number of interpolated window coefficients ci (n) of the window function to be applied to a frame presenting NT / 2 audio samples in the time domain comprises half the number of window coefficients.

[0091] To further illustrate this, in Figure 3 the window coefficients 450-0, ..., 450-7 are shown at the top of Figure 3 corresponding to a window coefficient c (0),. .., c (7). Based on said window coefficients and on the additional window coefficients of the window function, an application of equation (15) allows to obtain the window coefficients ci (n) of the function

ES 2 631 906 T3 of interpolated window shown in the lower part of Figure 3. For example, based on window coefficients 450-2 and 450-3, window coefficient 460-1 is generated based on equation (15 ), as represented by arrows 470 in Figure 3. Accordingly, the window coefficient 460-2 of the interpolated window function is calculated based on the window coefficient 450-4, 450-5 of the window function depicted in the upper part of Figure 3. Figure 3 shows generating additional window coefficients ci (n).

[0092] To illustrate the cancellation of aliasing that can be achieved by reducing the interpolated sample of the window function, Figure 4 represents the interpolation of the window coefficients in the case of a sinusoidal window function, which can be used, for example, in an MDCT. For the sake of simplicity, the left half of the window function and the right half of the window function are drawn over each other. Figure 4 represents a simplified version of a sinusoidal window, comprising only 2 4 window coefficients or points for an MDCT having a length of 8 samples.

[0093] Figure 4 shows four window coefficients 480-1, 480-2, 480-3 and 480-4 of the first half of the sinusoidal window and four window coefficients 490-1, 490-2, 490-3 and 390-4 of the second half of the sinusoidal window. The window coefficient 490-1, ..., 490-4 corresponds to the window coefficient indices 5, ..., 8. The window coefficients 490-1, ..., 490-4 correspond to the second half of the length of the window function, so N '= 4 must be added to the given indices to obtain the actual indices.

[0094] To reduce or even to achieve cancellation of the effects of aliasing as described above, the window coefficient should satisfy the condition v (n) · (Wl - n) = w (W + n ) · W (2W-l - n) (16) as best as possible. The better the relation (16) is satisfied, the better the aliasing or aliasing is.

Assuming the situation that a new window function having half the number of window coefficients must be determined for the left half of the window function, the following problem arises. Because the window function comprises an even number of window coefficients (numbered with sample reduction), without using an interpolation scheme as indicated in Figure 3, the window coefficients 480-1 and 480-3 or 480-2 and 480-4 correspond to only an aliasing value of the original window function or original filter.

This causes an unbalanced proportion of spectral energy and an asymmetric redistribution of the center point (center of gravity) of the corresponding window function occurs. Based on the interpolation equation (15) for the window coefficient w (n) of Figure 4, the interpolated values I1 and I2 satisfy the alias generation relation (16) much better and therefore a significant improvement regarding the quality of processed audio data.

However, by using an even more elaborate interpolation scheme, for example a spline or other similar interpolation scheme, one could even obtain window coefficients that satisfy relation (16) even better. A linear interpolation is, in most cases, sufficient and allows a fast and efficient implementation.

[0098] In the situation where a typical SBR system using an SBR-QMF filter bank (QMF = quadrature mirror filter), it is not required to implement a linear interpolation scheme or other interpolation scheme since the prototype SBR filter -QMF comprises an odd number of prototype filter coefficients. This means that the prototype SBR-QMF filter comprises a maximum value relative to which sample reduction can be implemented in such a way that the symmetry of the prototype SBR-QMF filter remains intact.

[0099] In Figures 5 and 6, a possible application of the embodiments according to the present invention will be described in the form of both an analysis filter bank and a synthesis filter bank. An important field of application is an SBR system or an SBR tool (SBR = spectral band replication).

[0100] However, other applications of the embodiments according to the present invention may come from other fields, where there is a need for spectral modifications (eg gain modifications or equalizations), such as spatial coding of objects audio, low-delay parametric stereo coding, spatial / envelope coding, frame loss concealment, echo cancellation or other corresponding applications.

ES 2 631 906 T3

[0101] The basic idea behind SBR is the observation that there is normally a great correlation between the characteristics of a high frequency range of a signal, which will be called a high band signal, and the characteristics of the band frequency range. low band, such as so-called low band or low band signals, of the same signal present. Thus, a good approximation for the representation of the original high-band input signal can be achieved by transposing the low-band to the high-band.

[0102] In addition to transposition, the reconstruction of the high band incorporates spectral envelope shaping, which comprises a gain adjustment. This process is normally controlled by a transmission of the high-band spectral envelope of the original input signal. The additional orientation information sent from the encoder controls other synthesis modules, such as an inverse filter, a noise addition, and sinusoidal to cope with the audio material when transposition alone may prove insufficient. The corresponding parameters include the high band noise parameters for the noise addition and the high band parameter of tonalities for the sinusoidal addition. This guidance information is commonly called SBR data.

[0103] The SBR process can be combined with any conventional codec or waveform by pre-processing on the encoder side and post-processing on the decoder side. The SBR encodes the high frequency part of an audio signal at a very low cost, while the audio codec is used to encode the lower frequency part of the signal.

[0104] On the encoder side, the original input signal is analyzed, the high-band spectral envelope and its characteristics relative to the low-band are encoded, and the resulting SBR data is multiplexed with a codec bit stream for the low band. First, the SBR data is demultiplexed on the decoder side. The decoding process is generally organized in stages. First, the core decoder generates the low band, and second, the SBR decoder functions as a post processor that uses the decoded SBR data to guide the spectral band replication process. This is followed by a full bandwidth output signal.

[0105] In order to obtain as high a coding efficiency as possible and to maintain low computational complexity, SBR-enhanced codecs are often implemented as so-called double-rate systems. Double speed means that the band-limited core codec is operating at half the external audio sample rate. Instead, the SBR part is processed at the full sample rate.

[0106] Figure 5 shows a schematic functional diagram of an SBR 500 system. The SBR 500 system comprises, for example, an AAC-LD encoder (AAC-LD = advanced audio codec - low delay) 510 and an SBR 520 encoder to which the audio data to be processed in parallel is provided. The SBR 520 encoder comprises an analysis filter bank 530, which is shown in FIG. 5 as a QMF analysis filter bank. The analysis filter bank 530 can provide audio subband values of corresponding to subbands based on the audio signals provided to the SBR 500 system. Next, said audio subband values are supplied to an SBR 540 parameter extraction module, which generates the SBR data as described above, comprising for example the spectral envelope for the high band, the band noise parameter high and high band tonality parameter. This SBR data is then provided to the AAC-LD 510 encoder.

[0107] The AAC-LD 510 encoder is depicted in Figure 5 as a double rate encoder. That is, the encoder 510 operates at half the sample rate compared to the sample rate of the audio data supplied to the encoder 510. To facilitate this, the AAC-LD encoder 510 comprises a sample reduction stage 550, which may optionally comprise a low pass filter to avoid distortions caused, for example, by a violation of the NyquistShannon Theorem. The sample reduction audio data as output by the sample reduction stage 550 is then supplied to an encoder 560 (analysis filter bank) in the form of an MDCT filter bank. The signals provided by encoder 560 are then quantized and encoded in quantization and encoding step 570. Furthermore, the SBR data provided by the SBR 540 parameter extraction module is also encoded to obtain a bit stream, which will then be output by the ACC-LD 510 encoder. The quantization and encoding stage 570 can, for example, quantize the data according to the properties of the human ear.

[0108] The bit stream is then supplied to an AAC-LD 580 decoder, which is part of the side of the decoder to which the bit stream is transported. The AAC-LD decoder comprises a decoding and dequanting stage 590, which extracts the SBR data from the bit stream and the dequantized or re-quantized audio data in the frequency domain representing the low band. The data from

ES 2 631 906 T3 low band are then supplied to a synthesis filter bank 600 (reverse MDCT filter bank). In the reverse MDCT stage (MDCT-1) 600 the signals provided to the reverse MDCT stage are converted from the frequency domain to the time domain to provide a temporal signal. Said time domain signal is then supplied to the SBR decoder 610, which comprises an analysis filter bank 620, which is represented in Figure 5 as a QMF analysis filter bank.

[0109] Analysis filter bank 620 performs spectral analysis of the temporal signal supplied to analysis filter bank 620 representing the low band. Said data is then supplied to a high frequency generator 630, which is also called an HF generator. Based on the SBR data provided by the AAC-LD encoder 580 and its decoding and dequanting stage 590, the HF generator 630 generates the high band based on the low band signals provided by the analysis filter bank 620. Both the low-band and high-band signals are supplied to a synthesis filter bank 640, which transfers the low-band and high-band signals from the frequency domain to the time domain to provide an audio output signal from the time domain of the SBR 500 system.

[0110] For the sake of completeness, it should be noted that in many cases the SBR 500 system, as depicted in Figure 5, is not implemented in this way. More accurately, the AAC-LD 510 encoder and the SBR 520 encoder are typically implemented on the encoder side, which is typically implemented separately from the decoder side comprising the AAC-LD 580 decoder and the SBR 610 decoder. That is, the system 500 shown in Figure 5 substantially represents the connection of two systems, in particular an encoder comprising the above-mentioned encoders 510, 520 and a decoder comprising the above-mentioned decoders 580, 610.

[0111] The embodiments according to the present invention in the form of analysis filter banks 100 and synthesis filter banks 300 can be implemented for example in the system 500 represented in figure 5 in replacement of the analysis filter bank 530 , analysis filter bank 620 and synthesis filter bank 640. That is, the synthesis or analysis filter banks of the SBR elements of the system 500 can be replaced, for example, by the corresponding embodiments according to the present invention. In addition, the MDCT 600 and the reverse MDCT 560 can also be replaced by low-delay analysis and synthesis filter banks, respectively. In this case, if all the described replacements have been implemented, the so-called low-delay enhanced AAC codec (codec = codec-decoder) will be used.

[0112] The enhanced low-delay AAC (AAC-ELD) aims to combine the low-delay characteristics of an AAC-LD (Advanced Audio Codec - Low Delay) with the high coding efficiency of HE-AAC (Audio Codec). High Performance Advanced Audio) using SBR with AAC-LD. The SBR 610 decoder in this case acts as a post processor, which is supplied after the core decoder 580 comprises a complete analysis filter bank and a 640 synthesis filter bank. Therefore, the components of the SBR 610 decoder add an additional decoding delay, which is represented in Figure 5 by shading the elements 620, 630, 540.

[0113] In many SBR 500 system implementations, the lower frequency portion or low band typically ranges from 0 kHz to typically 5-15 kHz and is encoded using a waveform encoder, called a core codec. The core codec can be, for example, one of the MPEG family of audio encoders. Furthermore, a reconstruction of the high frequency or high band part is achieved by a low band transition. The combination of SBR with a core encoder is implemented in many cases as a double rate system, in which the underlying AAC encoder / decoder is operated at half the sample rate of the SBR encoder / decoder.

[0114] Most of the control data is used in spectral envelope representation, which features variable frequency and temporal resolution in order to control the SBR process in the best possible way with the least possible bit rate overhead. The other control data is primarily intended to control the ratio of high band tone to noise.

[0115] As depicted in Figure 5, the output of the underlying AAC decoder 580 is typically analyzed with a 32-channel QMF filter bank 620. The HF generator module 630 then re-creates the high band by connecting QMF sub-bands from the existing low band to the high band. In addition, the reverse filter is performed by a subband, based on the control data obtained from the bit stream (SBR data). The envelope adjuster modifies the spectral envelope of the regenerated high band and adds additional elements such as noise and sinusoids are added according to the bitstream control data. Since all operations are performed in the domain

ES 2 631 906 T3 frequency (also known as QMF or subband domain), the final stage of decoder 610 is a synthesis of QMF 640 to retain a time domain signal. For example, in the case where the encoder-side QMF analysis is performed on a 32-QFM subband system for 1024 time-domain samples, in the high-frequency reconstruction 64-QMF subbands are obtained over which performs the synthesis producing 2048 time domain samples, so that an oversampling by a factor of 2 is achieved.

[0116] Additionally, the 510 core encoder delay doubles when operating at half the original sample rate in double rate mode, resulting in additional sources of delay in both the encoder and the decoder process. an AAC-LD in combination with SBR. These delay sources will then be examined and their associated delay will be minimized.

[0117] Figure 6 shows a simplified functional diagram of the system 500 depicted in Figure 5. Figure 6 concentrates on delay sources in the encoder / decoder process that use SBR and low delay filter banks for encoding. Comparing Figure 6 with Figure 5, the MDCT 560 and the reverse MDCT 600 have been replaced by delay optimized modules, the so-called MDCT 560 '(LD MDCT) and the low delay MDCT 600' (LD MDCT). In addition, the 630 HF generator has also been replaced by a 630 'optimized delay module.

[0118] Apart from the low-delay MDCT 560 'and the low-delay reverse MDCT 600', a modified SBR transmission and a modified HF generator 630 'are used in the system shown in Fig. 6. In order to avoid the delay through the different transmission of a 560, 600 core encoder / decoder and the corresponding SBR modules, the SBR transmission is suitable to accommodate the transmission length of 480 or 512 samples of the AAC-LD. Furthermore, the variable temporal grating of the HF generator 630, involving 384 delay samples, is restricted with respect to broadcasting SBR data over adjacent AC-LD frames. Therefore, the only remaining sources of delay in the SBR module are filter banks 530, 620, and 640.

[0119] According to the situation represented in figure 6, which represents a partial implementation of the AAC-ELD codec, some delay optimizations have already been implemented, including the use of a low delay filter bank in the AAC-LD core and removing a previously mentioned SBR overlay. For further delay improvements, the remaining modules should be investigated. Figure 6 depicts the delay sources in the encoder / decoder process using SBR and the low-delay filter banks called in this case LD-MDCT and LD-IMDCT. Compared to Figure 5, in Figure 6 each box represents a delay source, in which the delay optimization modules are outlined. So far similar modules have not been optimized for low delay.

[0120] Figure 7a represents a flow chart comprising a pseudo-code in C or C ++ to illustrate an embodiment according to the present invention in the form of an analysis filter bank or a corresponding method for generating audio subband values in subband audio channels. To be even more precise, Figure 7a represents a flow chart of a complex value analysis filter bank for 32 bands.

[0121] As noted above, the analysis filter bank is used to divide the time domain signal, eg the core encoder output into N = 32 subband signals. The filterbank output, subband samples, or audio subband values, in the case of a complex value analysis filterbank, are valued and therefore oversampled by a factor of 2 compared to a real value filter bank. Filtering involves and comprises the following steps, in which an x (n) matrix comprises exactly 320 samples in the time domain. The higher the index of the n samples in the array, the older the samples will be.

[0122] After starting the process embodiments in step S100, first, the matrix samples x (n) are shifted 32 positions in step S110.

The oldest 32 samples are discarded and 32 new samples are stored at positions 31 to 0 in step S120. As shown in FIG. 7a, the incoming time domain audio samples are stored at positions corresponding to a decreasing index n in the range of 31 to 0. This causes a temporal inversion of the samples stored in the corresponding frame or vector, in such a way that the inversion of the index of the window function has already been taken into account to obtain the analysis, the window function being based on the window function synthesis (equally long).

[0123] During a step S130, the window coefficients ci (j) are obtained by a linear interpolation of the coefficients c (j) based on equation (15). The interpolation is based on a block size (block length or number of subband values) of N = 64 values and is based on a frame comprising T = 10 blocks. Therefore, the index of the window coefficients of the interpolated window function is in the range between 0 and 319 according to equation (15). The

ES 2 631 906 T3 window coefficients c (n) are indicated in the table in Annex 1 of the description. However, depending on the details of the implementation, to obtain the window coefficients based on the values given in the tables in Annexes 1 and 3, the changes of additional signs with respect to the corresponding window coefficients must be taken into account. to the indices 128 to 255 and 384 to 511 (multiplication with factor (-1)).

[0124] In these cases, the window coefficients w (n) or c (n) to be used can be obtained according to

Mn) = wtable(n) s (n) (16a) with the sign change function s (n) according to s (n) = for 128 <n <255 and 384 <n <51Í else (16b) for n = 0 to 639, where wtable (n) are the values provided in the tables in the Annexes.

[0125] However, the window coefficients are not required to be implemented according to the table in Annex 1 to obtain, for example, the delay reduction described above. To achieve said delay reduction, while maintaining the quality level of the processed audio data or, to achieve other compensation, the window coefficients c (n) for the window coefficient index n in the range comprised between 0 and 639, they can satisfy one of the sets of relationships listed in one of Annexes 2 to 4. Furthermore, it should be noted that other window coefficients c (n) may also be employed in embodiments according to the present invention. Of course, other window functions comprising a different number of window coefficients of 320 or 640 may also be implemented, although the tables in Annexes 1 to 4 only apply to window functions that have 640 window coefficients.

[0126] The linear interpolation according to S130 causes a significant improvement in the quality and the reduction or cancellation of the aliasing effects in the case of a window function comprising an even number of window coefficients.

It should also be noted that the complex unit is not j as in equations (1), (2) and (16), but is indicated by i =^~ 1

[0127] In step S140, the samples of the matrix x (n) are multiplied by elements by the coefficients ci (n) of the interpolated window.

[0128] In step S150, the windowed samples are summed according to the equation given in the flow chart of Fig. 7a to create the 64 element matrix u (n). In step S160, 32 new subband samples or audio subband values W (k, 1) are calculated according to the matrix operation Mu, in which the element of the matrix M is given by

where exp () indicates the complex exponential function and, as mentioned above, i is the imaginary unit. Before the loop of a flow chart ends with step S170, each of the subband values W (k, 1) (= W [k] [1]), corresponding to the subband sample, can be output 1 in the sub-band with the k index. In other words, each loop of the flowchart shown in Figure 7a yields 32 subband values of complex values, each representing the output of a subband from a filter bank.

[0129] Figure 7b represents step S150 in which the frame 150 of windowed time domain audio samples is collapsed comprising 10 blocks 160-1, ..., 160-10 of windowed time domain audio samples z (n) towards the vector u (n) by a sum of 5 times two blocks of frame 150 each. Collapse or retraction is performed based on elements such that the time domain audio samples are added in windows corresponding to the same sample rate within each of the blocks 160-1, 160-3, 160-5, 160 -7 and 160- 9 to get the value

ES 2 631 906 T3 corresponding in the first blocks 650-1 of the vector u (n). Accordingly, based on blocks 160-2, 160-4, 160-6, 160-8 and 160-10, corresponding elements of vector u (n) are generated in block 160-2 in step S150.

[0130] A further embodiment according to the present invention in the form of an analysis filter bank may be implemented as a 64-band complex low-delay filter bank. The processing of this complex low-delay filter bank as an analysis filter bank is basically similar to the analysis filter bank as described in connection with FIG. 7a. Due to the similarities and substantially the same process that has been described in relation to Figure 7a, the differences between the complex analysis filter bank described for 32 bands of Figure 7a and the complex analysis filter bank for 64 subbands will be described here.

[0131] Unlike the 32 subbands that the analysis filter bank comprises as represented in figure 7a, the frame vector x (n) comprises, in the case of a 64-band analysis filter bank , 640 elements having the indices from 0 to 639. Therefore, step S110 is modified such that the samples of the matrix x (n) are shifted 64 positions and the oldest 64 samples are discarded. In step S120 instead of 32 new samples, 64 new samples are stored in positions 63 to 0. As shown in FIG. 7c, the incoming time domain audio samples are stored in positions that correspond to an index. decreasing n in the range from 63 to 0. This gives rise to a temporal inversion of the samples stored in the corresponding frame or vector, in such a way that the inversion of the index of the window function has already been taken into account to obtain the analysis window function based on the function of synthesis window (of equal length).

[0132] Since the window c (n), used for the window of the elements of the frame vector x (n), normally comprises 640 elements, the step S130 of linear interpolation of the window coefficients can be omitted to obtain the interpolated windows ci (n).

[0133] Next, during step S140, the samples of the matrix x (n) are multiplied or windows are generated using the sequence of window coefficients c (n), which is again based on the values in the table of the Annex 1. In the case of the window coefficient c (n) are those of the synthesis window function, the generation of windows or the multiplication of the matrix x (n) by the window c (n) are carried out according to the equation z (n) = x (n) c (n), {18) for an = 0, ..., 639. Again, to achieve the low-delay properties of the window function, it is not necessary to implement the window function exactly according to the window coefficients based on the values given in the table of Annex 1. For many applications, an implementation in which the window coefficients satisfy any of the sets of relationships listed in the tables in Annexes 2 to 4 will be sufficient to achieve an acceptable balance between quality and a significant reduction in delay. . However, depending on the details of the implementation, to obtain the window coefficients based on the values given in the tables in Annexes 1 and 3, the changes of additional signs with respect to the corresponding window coefficients must be taken into account. to the indices 128 to 255 and 384 to 511 (multiplication by factor (-1)) according to equations (16a) and (16b).

[0134] Step S150 of the flow chart of figure 7a is replaced by a sum of the samples of the frame vector z (n) according to the equation

Φ) - Σ (n + j ·128) j = o (19) to create the 128-element array u (n).

[0135] Step S160 of figure 7a is then replaced by a step in which 64 new subband samples are calculated according to the matrix operation Mu, in which the elements of the matrix M are given by

where exp () indicates the complex exponential function and i is, as mentioned above, the imaginary unit.

ES 2 631 906 T3

[0136] Fig. 7c is an exemplary flow chart in the form of a real value analysis filter bank for 32 subband channels. The embodiment shown in figure 7c does not differ significantly from the embodiment shown in figure 7a. The main difference between the two embodiments is that the step S160 for calculating the new 32 subband audio values of complex values is replaced in the embodiment shown in FIG. 7c by a step S162 in which 32 are calculated. subband audio samples of real values according to an Mru matrix operation, in which the elements of the Mr are matrix are given by

[0137] As a consequence, each loop of the flow chart produces 32 subband samples of real values where W (k, 1) corresponds to the audio sample of subband 1 of subband k.

[0138] The real value analysis filter bank can be used for example in the framework of a low power mode of an SBR system, as shown in figure 5. The low power mode of the SBR tool differs of the high-quality SBR tool primarily where real value filter banks are used. This reduces the computational complexity and computational effort by a factor of 2, such that the number of operations per unit time is substantially reduced by a factor of 2, since no imaginary part is required to be computed.

[0139] The new filter banks proposed according to the present invention are fully compatible with the low power mode of SBR systems. Thus, with filter banks according to the present invention, SBR systems can operate both in normal mode and in high quality mode with complex filter banks and in low power mode with real value filter banks. For example, the real value filter bank can be derived from the complex filter bank using only the real values (cosine modulated contributions) and omitting the imaginary values (sinusoidal modulated contributions).

[0140] Figure 8a shows a flow chart according to an embodiment of the present invention in the form of a complex value synthesis filter bank for 64 subband channels. As indicated above, the synthesis filter of the SBR processed subband signals is achieved using 64 subband synthesis filter banks according to an embodiment of the present invention. The output of the filter bank is a block of time domain samples of real values as indicated in relation to Figure 1. The process is illustrated by the flow chart of Figure 8a, which also illustrates an embodiment according to the present invention in the form of a method for generating audio samples in the time domain.

[0141] The synthesis filter comprises after a start (step S200), the following steps, in which a matrix v comprises 1280 samples. In step S210 the samples in matrix v are shifted 128 positions and the oldest 128 samples are discarded. In step S220, the 64 new complex value audio subband values are multiplied by a matrix N, where the elements of the matrix N (k, n) are given by

where exp () indicates the complex exponential function and i is the imaginary unit. The actual part of the output of said operation is stored in position 0-127 of matrix v, as illustrated in Figure 8a.

[0142] In step S230, the samples, which are now in the time domain, are extracted from the matrix v according to the equation given in Figure 8a to create a matrix of 640 elements g (n). In step S240, the samples of real values in the time domain of the set g are multiplied by the window coefficient c (n) to produce a matrix w, in which the window coefficients of a preferred embodiment according to the present invention are again the coefficients of the window based on the values indicated in the table in Annex 1.

[0143] However, as indicated above, it is not necessary for the coefficients of the windows to be based exactly on the values given in the table in Annex 1. In different embodiments according to the present invention it is sufficient, if the window coefficients satisfy one of the sets of relationships as provided in the tables in Annexes 2 to 4, to achieve the intended low delay property of the synthesis filter bank. Also, as described in

In relation to the analysis filter bank, other window coefficients can also be used within the framework of the synthesis filter bank. However, depending on the details of the implementation, to obtain the window coefficients based on the values given in the tables in Annexes 1 and 3, the changes of additional signs with respect to the corresponding window coefficients must be taken into account. to the indices 128 to 255 and 384 to 511 (multiplication with factor (-1)).

[0144] In step S250, 64 new output samples are calculated by a sum of samples of the matrix w (n) according to the last stage and the formula given in the flow diagram of figure 8a, before a loop of a flow chart ends in step S260. In the flow chart as represented in FIG. 8a, X [k] [l] (= X (k, l)) corresponds to the value 1 audio subband in the subband having the index k. Each new loop as depicted in Figure 8a produces 64 time domain audio samples and real values as output.

[0145] The implementation as depicted in figure 8a of a complex value analysis filter bank for 64 bands does not require an overlay / add buffer comprising various storage locations as described in relation to the shape embodiment represented in figure 2b. In this case, the overlap and add buffer is hidden in vectors v and g, which is calculated based on the values stored in vector v. The superposition and addition buffer is implemented within the framework of said vectors with said indices greater than 128, in such a way that the values correspond to values of previous or previous blocks.

[0146] FIG. 8b is a flow chart of a real value analysis filter bank for 64 real value subband channels. The real-value synthesis filter bank according to FIG. 8b can also be implemented in the case of a low-power SBR implementation such as a corresponding SBR filter bank.

[0147] The flow chart of figure 8b differs from the flow chart of figure 8a, mainly with respect to step S222, which replaces the S220 of figure 8a. In step S222, the 64 new real value audio subband values are multiplied by a matrix Nr, where the elements of the matrix Nr (k, n) are given by

in which the output of said operation is stored again in position 0-127 of the matrix v.

[0148] Apart from said modifications, the flow chart as represented in figure 8b in the case of a real value synthesis filter bank for the low power SBR mode, does not differ from the flow chart as shown represents in figure 8a the complex value synthesis filter bank for the high quality SBR mode.

[0149] Figure 8c illustrates a flow chart according to an embodiment of the present invention in the form of a complex value synthesis filter bank with reduced sampling and the suitable procedure, which can be implemented, for example, in a high-quality SBR implementation. More accurately, the synthesis filter bank as described in Figure 8c refers to a complex value synthesis filter bank that can process complex value audio subband values for 32 subband channels.

[0150] Synthesis filtering of SBR process subband signals with sample reduction is achieved using a 32-channel synthesis filter bank as depicted in Figure 8c. The output of the filter bank is a block of time domain samples of real values. The process is provided in the flow chart of Figure 8c. The synthesis filter comprises after a start (step S300), the following steps, in which a matrix v comprises 640 samples of the time domain of real values.

[0151] In step S310 the samples of matrix v are shifted 64 positions and the oldest 64 samples are discarded. Next, in step S320, the 32 new real value subband samples or complex value audio values are multiplied by a matrix N, the elements of which are given by

ES 2 631 906 T3

where exp () indicates the complex exponential function and i is the imaginary unit. The real part of the output of said operation is stored in positions 0-63 of the matrix v.

[0152] In step S330, the samples are extracted from the vector v according to the equation given in the flow chart of figure 8c to create a matrix of 320 g elements. In step S340, the window coefficients ci (n) of an interpolated window function are obtained by a linear interpolation of the coefficients c (n) according to equation (15), in which the index n is again at the interval between 0 and 319 (N = 64, T = 10 for equation (15)). As illustrated above, the coefficients of a window function c (n) are based on the values indicated in the table in Annex 1. Furthermore, to achieve the low delay property, as illustrated above, it is not necessary that the window coefficients c (n) are exactly the figures given in the table in Annex 1. It is sufficient that the window coefficients c (n) satisfy at least one set of relationships as indicated in Annexes 2 to Four. However, depending on the details of the implementation, to obtain the window coefficients based on the values given in the tables in Annexes 1 and 3, the changes of additional signs with respect to the corresponding window coefficients must be taken into account. to the indices 128 to 255 and 384 to 511 (multiplication by factor (-1)) according to equations (16a) and (16b). Furthermore, different window functions comprising different window coefficients c (n) can of course also be used in embodiments of the present invention.

[0153] In step S350, the samples of the matrix g are multiplied by the coefficient of the interpolated window ci (n) of the interpolated window function to obtain the sample w (n) of the time domain in windows.

[0154] Next, in step S360, 32 new output samples are calculated by a sum of samples of the matrix w (n) according to the last step S360, before the final step S370 of the flow chart of figure 8c .

[0155] As previously indicated, in the flow diagram of figure 8c, X [k] [l] (= X (k, l)) corresponds to the value 1 in the audio subband channel the index k . Furthermore, each such new loop of a flow chart as depicted in FIG. 8c produces 32 time domain samples and real values as output.

[0156] Fig. 8d shows a flow chart of an example according to the present invention in the form of a real value synthesis filter bank with sample reduction, which can be used for example in the case of a filter bank Low power SBR. The example and flow chart shown in Figure 8d differs from the flow chart shown in Figure 8c of the sample reduction complex value synthesis filter bank only in relation to step S320, which is substituted in the diagram flow shown in FIG. 8d by step S322.

[0157] In step S322, the 32 new real value audio subband values, or subband samples, are multiplied by a matrix Nr, where the elements of the matrix Nr (k, n) are given by

in which the output of said operation is stored in position 0 to 64 of the matrix v.

[0158] Figure 9a shows a further implementation of an embodiment according to the present invention in the form of a method comprising a bank of complex value analysis filters for 64 subbands.

Figure 9a shows an implementation as a MATLAB implementation, which outputs a vector and a state vector. The function as defined in said script shown in figure 9a is called LDFB80 which is provided with a vector x comprising recent audio samples and the state vector as input. The function designation LDFB80 is an abbreviation for low delay filter bank for 8 blocks extending in the past and 0 blocks in the future.

[0159] In the MATLAB programming language, the percent sign (%) indicates observations, which are not

ES 2 631 906 T3 are executed, but merely serve the purpose of commenting and illustrating the source code. In the following description, various segments of the source code will be described with respect to their functions.

[0160] In the S400 code sequence, the buffer represented by the state vector is updated in such a way that the content of the state vector presenting indices 577 to 640 is replaced by the content of vector x that comprises audio input samples recent time domain.

In the S410 code sequence, the window coefficients of the analysis window function stored in the variable LDFB80_win are transferred to the win_ana vector.

[0161] In step S420, which assumes that the last samples are aligned to the right side of the buffer, the actual window generation is executed. In block S420, the content of the state vector is multiplied by elements (. *) With the elements of the win_ana vector that comprise the analysis window function. The output of this multiplication is then stored in the vector x_win_orig.

[0162] In step S430, the content of the vector x_win_orig is reshaped to form a matrix of size 128 5 elements called x_stack. In step S440, the sign change of the stack x_stack is performed with respect to the second and fourth columns of the array x_stack.

[0163] In step S450, the x_stack stack collapses or retracts by adding the elements of x_stack with respect to the second index and simultaneously reversing the order of the elements and transposing the result before storing the result again in the various x_stacks.

[0164] In the code segment S460, the transformation of the time domain into the frequency domain is performed by calculating a complex fast Fourier transformation (FFT) of the content multiplied by elements of the stack x_stack multiplied by the complex exponential function to which provides the argument (-i π n / 128), with indices y in the range 0 to -127 and the imaginary unit i.

[0165] In the S470 code segment, a subsequent rotation is performed defining the variable m = (64 + 1) / 2 and calculating the block that comprises the audio subband values as a vector and according to the equation y (k) = 2 Temp (k) exp (- 2i · Λ · · ((k - 1 + ¡) · ^))

The index k covers the whole number range from 1-64 in the implementation depicted in Figure 9a. Next, the vector y is output as a vector or block comprising the audio subband values 180 of figure 1. The bar above the second factoring equation (26) as well as the conj () function encode the segment S417 of Figure 9a and refer to the complex conjugate of the corresponding complex number argument.

[0166] In a segment of the final code S480, the state vector is shifted 64 elements. The state vector in its shifted form can then be provided to the LDFB80 function as input again in an additional loop of the function.

[0167] Figure 9b shows a MATLAB implementation according to an embodiment of the present invention in the form of a method comprising a bank of complex value analysis filters for 32 subbands. Consequently, the defined function is named LDFB80_32, indicating that the implementation represents a low-delay filter bank for 32 subbands based on an additional overlap of 8 blocks in the past and 0 blocks in the future.

[0168] The implementation of Figure 9b differs from the implementation depicted in Figure 9a, only with respect to some sequences of the code, as will be indicated in the following description. The code sequences S400, S430, S460, S470 and S480 are replaced by the corresponding code sequences, mainly taking into account the fact that the number of subbands or the number of subband values emitted by the LDFB80_32 function is reduced by one factor of 2. Accordingly, step S400 'refers to the state vector that is updated with respect to the last 32 entries corresponding to the indices 289 to 320 with the corresponding 32 audio input samples from the time domain of the recent block 220 as shown. in figure 1.

[0169] However, the main difference between the implementations, as depicted in Figures 9a and 9b, appear in the S410 code sequence of Figure 9a, which is replaced by a S412 code sequence in the implementation depicted in Figure 9b. The code sequence for S412 of FIG. 9b first comprises a copy of the 640 window coefficients comprising windows stored in the LDFB80_win vector for the win_ana local vector. Then,

ES 2 631 906 T3 an interpolation is carried out according to equation (15), in which two consecutive window coefficients represented by the vector elements of the win_ana vector are added and divided by 2 and then stored again in the vector win_ana.

[0170] The following code sequence S420 is identical to the code sequence S420 as represented in figure 9a that performs the real multiplication by elements (. *) Of the generation of windows of values or elements of the state vector with the elements of the win_ana vector that comprise the interpolated window coefficients of the interpolated window function. The output of this operation is stored in the vector x_win_orig. However, the difference between the sequence of the S420 code of figure 9b and the corresponding sequence of the S420 code of figure 9a, is that in the case of figure 9b, not 640 but only 320 multiplications are performed and not 640 in the window generation framework.

[0171] In the code sequence S430 'that replaces the code sequence S430, the x_stack stack is prepared by reshaping the vector x_win_orig. However, since the vector X_win_orig comprises only 320 elements, compared to the corresponding vector in Figure 9a which comprises 640 elements, the matrix x_stack is only a 64 · 5 element matrix.

[0172] The sequences of the sign change code S440 and the sequence of the stack collapse code S450 are identical in both implementations according to Figures 9a and 9b, in addition to the lower number of elements (320 compared to 640).

[0173] An odd complex fast Fourier transform (FFT) of window data is performed on the S460 'code sequence that replaces the S460 code sequence, which is quite similar to the transform of the S460 code sequence of the figure 9a.

However, once again, due to the reduced number of output audio subband values, the temp vector is provided obtaining a fast Fourier transform, the multiplication by elements of the x_stack stack and the complex exponential function of the argument (-i · Π · n / 64), finding the index n in the interval between 0 and 63.

[0174] Subsequently, in the modified code sequence S470 ', the posterior rotation is carried out defining the variable m = (32 + 1) / 2 and generating the output vector and according to equation (26), in which the index k only covers the interval from 1 to 32 and in which the number 128 that appears in the argument of the complex exponential function is replaced by the number 64.

[0175] In the final code sequence S480 ', the buffer state is shifted by 32 elements in the case of the implementation shown in FIG. 9b, where in the corresponding code sequence S480, the buffer is shifted 64 elements.

[0176] Figure 10a shows a MATLAB script illustrating an implementation according to an embodiment of the present invention in the form of a method comprising a complex value analysis synthesis filter bank for 64 subbands. The sequence shown in Figure 10a defining the ILDFB80 function is provided as input parameters to which the vector x representing the block 320 of audio subband values of Figure 2a and a state vector state1. The ILDFB80 designation indicates that the defined function is a low-delay inverse filter, corresponding to 8 blocks of audio data from the past and 0 blocks from the future. The function provides a vector y and a new or redefined state vector as output, where vector y corresponds to block 410 of time-domain audio samples of FIG. 2a.

[0177] In an S500 code sequence, a pre-spin is performed, in which a variable m = (64 + 1) / 2 is defined as well as a temp vector. The elements temp (n) of the vector temp are defined according to the equation temp (n) = j · · exp (2i · π (η - 1 + f) ·,(27) in which the bar above the element of the vector x (n) and the function conj () represent the complex conjugate, exp () represents the complex exponential function, i represents the imaginary unit and n is an index in the range from 1 - to 64.

[0178] In the S510 code sequence, the temp vector is inverted in a matrix comprising in the first column the elements of the temp vector and in the second column, the complex conjugate of the inverted temp vector with respect to the order of the elements such as defined by the vector index. Thus, an odd symmetry of the temp matrix is established in the S510 code sequence based on the temp vector.

ES 2 631 906 T3

[0179] An odd fast Fourier transform (FFT) is performed on a sequence of the S520 code based on the temp matrix. In this code sequence, the real part of the multiplication by elements of the result of the inverse Fourier transform of the matrix temp is performed with the exponential function that presents the argument of (iπ / 128) and is sent to a vector y_knl, where the index n is in the range 0 to 127.

[0180] In the S530 code sequence, an extension of the data and an alternating sign change are formed. To achieve this, the order of the elements of the y_knl vector is reversed and, at the same time, a sign change is performed. Next, a matrix tmp is defined, comprising the first, third and fifth columns of the vector y_knl, in which the second and fourth columns comprise the vector with the changed sign y_knl.

[0181] In the S540 code sequence, the window coefficients stored in the LDFB80_win vector are first copied to the win_ana vector. Next, the synthesis window coefficients are determined based on the analysis window coefficients stored in the win_ana vector by generating a time-reversed version of the analysis window function according to win _ syn (n) = win _ ana (N T - n) where N · T is the total number of window coefficients and n is the index of the window coefficients.

[0182] In a S550 code sequence, the synthesis window is applied to the tmp vector by element multiplication of the vector with the synthesis window function. In an S560 code sequence, the buffer is updated by setting the elements of the "state" vector with indices 577 to 640 to 0 and adding the content of the tmp window vector to the state vector state.

[0183] In a sequence of the S570 code, the output vector y comprising the time domain audio samples is extracted from the state vector obtaining the elements of the state vector extracting the elements of the state vector with indices 1 to 64 .

[0184] In a sequence of the S580 code, the final code sequence of the function as represented in Figure 10a, the state vector state is shifted by 64 elements so that elements with indices 65 to 640 are copied to the first 576 elements of the state vector.

[0185] Figure 10b shows a MATLAB script of an implementation according to an embodiment of the present invention in the form of a filterbank synthesis analysis of complex values for 32 subband values. The naming of the function defined by the script shown in figure 10b illustrates this as the defined function is named ILDFB80_32, which indicates that the defined function is a low-delay inverse filter bank for 32 bands with 8 overlapping blocks from the past and 0 blocks overlap from the future.

[0186] As already mentioned with respect to the comparison of the implementation represented in Figures 9a and 9b, the implementation according to the script of Figure 10b is also closely related to the implementation of the 64 synthesis filter bank -subbands according to figure 10a. As a consequence, the same vectors are provided to the function and output by the function which, however, comprise only half the number of elements compared to the implementation of Figure 10a. The implementation of a 32-band synthesis filter bank for 32 bands differs from the 64 sub-band version depicted in Figure 10a in particular with respect to two aspects. The code sequences S500, S510, S520, S53B, S560, S570, and S580 are replaced by code sequences in which the number of items to be processed and the additional number of item-related parameters are divided by 2. Furthermore, the sequence of the S540 code of the generation of the synthesis window function is replaced by a sequence of the S542 code, in which the synthesis window function is generated as a linearly interpolated synthesis window function according to equation (15 ).

[0187] In the code sequence S500 'that replaces the code sequence S500, the variable m is defined as equal to m = (32 + 1) / 2 and the vector temp is defined according to equation (27), in the that the index n covers only the range from 1 to 32 and in which the factor of 1/128 is replaced by the factor 1/64 in the argument of the exponential function.

[0188] Accordingly, in the S510 'code sequence that replaces the S510 code sequence, the index range covers only the indices of element 32 comprising the vector temp. That is, the index covers only the values from 1 to 32. Therefore, in the sequence of the S520 'code that replaces the sequence of the S520 code, the argument of the exponential function is replaced by (i · π · n / 64) , in which the index n is in the range of 0 to 63. Under the code sequence

ES 2 631 906 T3

S530 ', the index range is also reduced by a factor of 2 compared to the S530 code sequence.

[0189] The S542 code sequence that replaces the S540 code sequence of Figure 10a also copies the window function stored in the LDFB80_win vector to the win_ana vector and generates a time-reversed version win_syn according to equation (28). However, the S542 code sequence of the implementation shown in figure 10b further comprises an interpolation stage according to equation (15), in which for each element of the redefined vector win_syn comprising the window coefficients of the window function synthesis, a linear interpolation of two consecutive window coefficients of the original synthesis window function.

[0190] The S550 code sequence of applying the window to the tmp vector and replacing the tmp elements with the windowing version thereof is identical in terms of code as a direct comparison of the corresponding code sequences in Figures 10a and 10b. However, due to the smaller size of the tmp vector in the implementation of Figure 10b, during one implementation, only half the number of multiplications is performed.

[0191] Also within the framework of the S560 ', S570' and S580 'code sequences that replace the S560, S570 and S580 code sequences, respectively, the indices 640 and 64 are replaced by 320 and 32, respectively. Therefore, said final three code sequences only differ from the implementation code sequences depicted in Figure 10a with respect to the size of the state vectors tmp and y.

[0192] As the embodiments described so far have illustrated, the analysis window generator as well as the synthesis window generator are capable of generating windows with the corresponding samples in the time domain comprised in the corresponding frames by multiplying the same by elements with window coefficients of a window function.

[0193] Before describing a window function, which can be used for example as a synthesis window function and as an analysis window function in its time-reversed version, the advantages of the embodiments will be described in more detail according to the present invention, in particular considering an implementation within the framework of a tool or SBR system as represented in figures 5 and 6.

[0194] Among the advantages, the embodiments according to the present invention and systems comprising more than one embodiment according to the present invention can provide a significant delay reduction according to other filter banks.

However, this low delay property will be described in more detail with reference to Figures 13 and 14. An important aspect in this context is to note that the length of the window function, that is, the number of window coefficients to be applied to a frame or a block of samples in the time domain, it is independent of the delay.

[0195] Furthermore, as will be described in connection with Figures 17 and 18 in more detail, as regards psychoacoustics, embodiments according to the present invention often utilize the temporal masking properties of the human ear better than many other filter banks. Furthermore, as will be described in more detail in connection with Figures 15, 16 and 19, embodiments according to the present invention provide excellent frequency response.

[0196] Also, in many filter banks according to an embodiment of the present invention, a perfect reconstruction can be achieved if an analysis filter bank and the synthesis filter bank are interconnected. That is, the embodiments according to the present invention not only offer an audible output indistinguishable compared to the input of said interconnected set of an analysis filter bank and a synthesis filter bank, but (apart from quantization errors, computational rounding effects and additional effects caused by the necessary discretization) an identical output compared to the input.

[0197] An integration in the SBR module of filter banks according to the present invention can be easily achieved. Although SBR modules typically operate in dual-rate mode, complex value low-delay filter banks according to embodiments of the present invention can provide perfect reconstruction in single-rate mode, while filter banks Original SBR QMFs can only provide a near perfect rebuild. In dual speed mode, the 32-band version of the impulse response is obtained by linear interpolation, also called sample reduction, of two adjacent coefficients or window coefficients of the 64-band impulse response or window function. as described in relation to figure 3.

ES 2 631 906 T3

[0198] In the case of a complex value implementation of a filter bank, a significantly reduced analysis (or synthesis) delay can be achieved for filter banks subjected to critical sampling, where the sampling frequency or processing corresponds to the border frequency according to the Nyquist-Shannon Theory. In the case of a real-valued implementation of a filter bank, an efficient implementation can be achieved using optimized algorithms, as for example illustrated in relation to the MATLAB implementation depicted in Figures 9 and 10. These implementations can be used , for example, in the low power mode of the SBR tool as described in relation to Figures 5 and 6.

[0199] As indicated in connection with Figures 5 and 6, a further reduction with respect to delay can be achieved in the case of an SBR system by using a bank of low-delay filters of complex values according to an embodiment of the present invention. As indicated above, in the SBR 610 decoder, as shown in Figure 5, the QMF 620 analysis filter bank is replaced by a complex low-delay filter bank (CLDFB) according to one embodiment of the present invention. This substitution can be carried out in a calculable way by keeping the number of bands (64), the length of the impulse response (640) and using complex modulation. The delay achieved by this tool is minimized to such an extent that a sufficiently low overall delay is achieved for two-way communication without sacrificing an achievable level of quality.

[0200] Compared, for example, to a system comprising an MDCT and an MDST to form a complex value MDCT-like system, an embodiment according to the present invention provides a much better frequency response. Compared to the QMF filter bank, for example, used in current MPEG-4 SBR, a system comprising one or more filter banks according to embodiments of the present invention provides significantly lower delay.

[0201] Even compared to a low-delay QMF filter bank, the embodiments according to the present invention provide the advantage of perfect reconstruction combined with the lowest delay. The advantages derived from the perfect rebuild property as opposed to the near perfect rebuild of QMF filter banks are as follows. For near-perfect reconstruction, high blocking band attenuation is required to attenuate aliasing to a low enough level. This limits the possibility of reaching a very low delay in the filter design. In contrast, the use of an embodiment according to the present invention currently entails the possibility of independently designing the filter in such a way that high blocking band attenuation is not required to attenuate the generation of aliases to sufficiently low levels. . The attenuation of the blocking band has to be low enough to allow a reduced generation of aliases sufficient for the intended signal processing application. Therefore, a better balance towards a lower delay can be achieved in the filter design.

[0202] FIG. 11 depicts a comparison of a window function 700 as it can be used, for example, in an embodiment according to the present invention in conjunction with the sinusoidal window function 710. The window function 700, which can be used also called CMLDFB synthesis window (CMLDFB = complex modulated low-delay filter bank), it comprises 640 window coefficients based on the values provided in the table in Annex 1. Regarding the magnitude of the windowing functions, it should be noted that general amplification factors or damping factors intended to adjust an amplitude of the windowed signal will not be considered in the following. The window functions can be normalized, for example, with respect to a value corresponding to the lag center, as indicated in relation to Figure 13 or with respect to a value n = N, n = N - 1 or n = N + 1, where N is the length of the block and n is the index of the window coefficients. In comparison, the sinusoidal window function 710 is defined only on 128 samples and is used, for example, in the case of an MDCT or MDST module.

[0203] However, depending on the details of the implementation, to obtain the window coefficients based on the values provided in the tables in Annexes 1 and 3, the additional sign changes with respect to the coefficients must be taken into account window corresponding to indices 128 to 255 and 384 to 511 (multiplication with factor (-1)) according to equation (16a) and (16b).

[0204] Before describing the differences of the two window functions 700, 710, it should be noted that both window functions comprise only real value window coefficients. Furthermore, in both cases, an absolute value of the window coefficient corresponding to an index n = 0 is less than 0.1. In the case of a 700 CMLDFB window, the corresponding value is even less than 0.02.

[0205] Considering the two window functions 700, 710 with respect to their definition sets, several significant differences are apparent. Although the sinusoidal window function 710 is symmetric, the window function 700 exhibits asymmetric behavior. To define this with

ES 2 631 906 T3 for clarity, the sinusoidal window function is symmetric since as a real value it does not exist with respect to all real numbers n, so that the window function 710 is defined for (n0 + n) and (n0 -n), the relation

is satisfied up to a convenient margin (ε> 0; the absolute value of the difference of the terms on the two sides of equation (29) is less than or equal to ε), with w (n) representing the window coefficient corresponding to the index n. In the case of the sinusoidal window, the corresponding index n0 it lies exactly in the center of the top two window coefficients. That is, in the case of the sinusoidal window 710 the index is n0 = 63.5. The sinusoidal window function is defined for the indices n = 0, ..., 127.

[0206] In contrast, the window function 700 is defined on the set of indices n = 0, ..., 639. The window function 700 is clearly asymmetric in the sense that for all numbers of real values there is no at least one real number, such that (n0 + n) and (n0-n) belong to the set of definitions of the window function, for which the inequality p (n0 -. * Κ (ηθ + n) | (3θ) maintains a (almost deliberately) definable margin (ε> 0, the absolute value of the difference of the terms on the two sides of equation (29) is greater than or equal to ε), once again w (n) is the value of the window coefficient corresponding to index n.

[0207] Other differences between the two window functions, which refer both to the block sizes of N = 64 samples, is that the maximum value of the window function 700 is greater than 1 and is acquired for indices in the range of

N <η <2N (31) for the synthesis window. In the case of the window function 700 represented in figure 11, the maximum value acquired is greater than 1.04 acquired at the sample index n = 77. On the other hand, the maximum values of the sinusoidal window 710 are less than or equal to a 1, and are captured at n = 63 and n = 64.

[0208] However, likewise the window function 700 acquires a value of approximately 1 at sample indices of approximately n = N. More accurately, the absolute value or the proper value of the corresponding window coefficient w (N-1) the index n = N-1 is less than 1, while the absolute value or the value of the window coefficient w (N) corresponding to the index n = N is greater than 1. In some embodiments according to the present invention, both window coefficients comply with the relationships

0.99 <w (N - l) <1.0 1.0 <w (N) <1.01 (32) which is the result of the optimization of the audio quality of the filter banks according to the embodiments of the present invention. In many cases it is intended to have a window coefficient w (0) that includes an absolute value as small as possible. In this case, a determinant of the window coefficients p (o) · - l) - w (N - l) · w (w) | «1 (33) must be as close as possible to 1 to achieve a quality of audio that is optimized with respect to possible parameters. However, the sign of the determinant provided by equation (33) is freely selectable. As a consequence of the fact that the window coefficient w (0) is less than or approximately 0, the product of w (N-1) · w (N) or its absolute values must be as close as possible to +/- 1. In this case, the window coefficient w (2N-1) can be chosen almost freely. Equation (33) is the result of using the zero-delay matrix technique as described in New Framework for Modulated Perfect Reconstruction Filter Banks.

ES 2 631 906 T3 perfect modulated) by GDT Schuller and MJT Smith, IEEE Transactions on Signal Processing, Vol. 44, No. 8, August 1996.

[0209] Furthermore, as will be described in more detail in connection with Figure 13, the window coefficients corresponding to the indices N-1 and N lie in the center of the modulation core and therefore correspond to the sample that has a value of approximately 1.0 and matches the filter bank delay defined by the filter function or window function prototype.

[0210] The synthesis window function 700 as represented in figure 11 also exhibits an oscillating behavior with strictly monotonous window coefficients increasing from the window coefficient of the sequence of window coefficients corresponding to the index (n = 0 ) used for the window of the last time-domain audio sample up to the window coefficient that comprises the highest absolute value of all the window coefficients in the synthesis window function 700. Naturally, in the case of the time-reversed analysis window function, the oscillation behavior comprises a strict monotonic decrease of the window coefficients from the window coefficient that comprises the highest absolute value of all the window coefficients of an analysis window function to the window coefficients of the sequence of coefficients of window corresponding to an index (n = 639) used for the window of the last audio sample of the time domain.

[0211] As a consequence of the oscillation behavior, the development of the synthesis window function 700 starts with a window coefficient corresponding to the index n = 0 which has an absolute value less than 0.02 and an absolute value of the coefficient of window corresponding to the index n = 1 less than 0.03, acquiring a value of approximately 1 in an index n = N, acquiring a maximum value greater than 1.04 in an index according to equation (31), acquiring an additional value of approximately 1 in an index n = 90 and 91, a first sign change in the index values of n = 162 and n = 163, acquiring a minimum value less than -0.1 or -0.12755 in an index of approximately n = 3N and another sign change in index values n = 284 and n = 285. However, the synthesis window function 700 may further comprise changes in the signs of other values of index n. When comparing the window coefficients with the values given in the tables in Annexes 1 and 3, the changes of additional signs with respect to the window coefficients corresponding to the indices 128 to 255 and 384 to 511 must be taken into account ( multiplication by factor (-1)) according to equations (16a) and (16b).

[0212] The oscillating behavior of the synthesis window function 700 is similar to that of a highly damped oscillation, which is illustrated by the maximum value of about 1.04 and the minimum value of about -0.12. As a consequence, more than 50% of all window coefficients comprise absolute values less than or equal to 0.1. As indicated in relation to the embodiments described in Figures 1 and 2a, the development of the window function comprises a first group 420 (or 200) and a second group 430 (or 210), the first group comprising 420 a first consecutive part of window coefficients and the second group 430 comprising a second one consecutive part of window coefficients. As already indicated above, the sequence of window coefficients of the window comprises only the first group 420 of window coefficients and the second group of window functions 430, the first group 420 of window coefficients comprising exactly the first sequence consecutive number of window coefficients and the second group 430 comprising exactly the second consecutive part of window coefficients. Therefore, the terms first group 420 and first part of the window coefficients as well as the terms second group 430 and second part of the window coefficients can be used synonymously.

[0213] More than 50% of all the window coefficients that present absolute values less than or equal to 0.1 are included in the second group or second part 430 of window coefficients as a consequence of the highly damped oscillating behavior of the window function 700. Furthermore, more than 50% of all the window coefficients included in the second group or second part 430 of window coefficients also comprise absolute values less than or equal to 0.01.

[0214] The first window coefficient part 420 comprises less than one-third of all the window coefficients in the sequence of window coefficients. Accordingly, the second window coefficient part 430 comprises more than two-thirds of the window coefficients. In the case of a total number of T blocks to be processed in one of the frames 120, 150, 330, 380 of more than four blocks, the first part normally comprises 3/2 N window coefficients, where N is the number of samples of the time domain of a block. Therefore, the second part comprises the rest of the window coefficients or, more accurately, (T-3/2) N window coefficients. In the case of T = 10 blocks per frame as represented in figure 11, the first part comprises 3/2 N coefficients

ES 2 631 906 T3 window, while the second part 210 comprises 8.5 · N window coefficients. In the case of a block size of N = 64 audio samples in the time domain per block, the first part comprises 96 window coefficients, while the second part comprises 544 window coefficients. The synthesis window function 700 as depicted in Fig. Fig. eleven acquires a value of approximately 0.96 at the edge of the first part and the second part at an index of approximately n = 95 or 96.

[0215] Despite the number of window coefficients comprised in the first part 420 and the second part 430, an energy value or a total energy value of the corresponding window coefficients differ significantly from each other.

The energy value is defined by £ = ΣΜηί t ' (34)

Λ. .

where w (n) is a window coefficient and the index n on which the sum of equation (34) is calculated corresponds to the indices of the corresponding parts 420, 430, the whole set of window coefficients or any another set of window coefficients to which the corresponding energy values E correspond. Despite the significant difference in window coefficients, the energy value of the first part 420 is equal to or greater than 2/3 of the total energy value of all the window coefficients. Consequently, the energy value of the second part 430 is less than or equal to 1/3 of the total energy value of all the window coefficients.

[0216] To illustrate this, the energy value of the first part 420 of the window coefficients of the window function 700 is approximately 55.85, while the energy value of the window coefficients of the second part 430 is approximately 22.81. The overall energy value of all the window coefficients of the window function 700 is approximately 78.03, such that the energy value of the first part 420 is approximately 71.6% of the total energy value , while the energy value of the second part 430 is approximately 28.4% of the total energy value of all the coefficients of the windows.

[0217] Of course, equation (34) can be expressed in a normalized version by dividing the energy value E by a normalization factor E0, which in principle can be any energy value. The normalization factor E0 can be, for example, the global energy value of all the window coefficients of the sequence of window coefficients calculated according to equation (34).

[0218] Based on the absolute values of the window coefficients or the energy values of the corresponding window coefficients, a center point or a center of gravity of the sequence of window coefficients can also be determined. The center of gravity or center point of the sequence of window coefficients is a real number and is usually within the range of indices of the first part 420 of the window coefficients. In the case of corresponding frames comprising more than four blocks of audio samples in the time domain (T> 4), the center of gravity nAC based on the absolute values of the window coefficients or the center of gravity nce based on the energy values of the window coefficients is less than 3 / 2N. That is, in the case of T = 10 blocks per frame, the center of gravity is within the index region of the first part 200.

[0219] The center of gravity nAC based on the absolute values of the window coefficients w (n) are defined according to

and the center of gravity nEC considering the energy values of the window coefficients w (n) is defined according to

ES 2 631 906 T3 (36)

ΝΓ-1

Ση ΜηΓ p - “ce Ν-Τ-1

Σ kW η = 0 where N and T are positive integers indicating the number of audio samples in the time domain per block and the number of blocks per frame, respectively. Of course, the center points according to equations (35) and (36) can also be calculated with respect to a limited set of window coefficients by substituting the limits of the above sums accordingly.

[0220] In the case of the window function 700 as represented in FIG. 1, the center of gravity nAC based on the absolute values of the window coefficients w (n) is equal to the value of nAC = 87.75 and the center point or center of gravity nCe with respect to the energy values of the window coefficients w (n) is nEC = 80.04. Since the first window coefficient part 200 of the window function 700 comprises 96 (= 3/2 N; N = 64) window coefficients, both center points lie within the first part 200 of the window coefficients, as indicated above.

[0221] The window coefficients w (n) of the window function 700 are based on the values indicated in the table in Annex 1. However, to achieve, for example, the low-delay property of the filter bank, such As stated above, it is not necessary to implement the window function with as much precision as given by the window coefficients in the table in Annex 1. In many cases, it is more than enough that the window coefficients of a window function that includes 640 window coefficients satisfy any of the relationships or equations that appear in the tables in Annexes 2 to 4. The coefficients of the windows or coefficients Filter values listed in the table in Annex 1 represent preferred values, which could be adapted according to equations (16a) and (16b) in some implementations. However, as indicated, for example, by the additional tables in the Additional Annexes, the preferred values of the second, third, fourth, fifth digits after the decimal point can be varied in such a way that the filters or functions of Resulting windows still have the advantages of the embodiments according to the present invention. However, depending on the details of the implementation, to obtain the window coefficients based on the values given in the tables in Annexes 1 and 3, the changes of additional signs with respect to the corresponding window coefficients must be taken into account. to the indices 128 to 255 and 384 to 511 (multiplication by factor (-1)) according to equations (16a) and (16b).

[0222] Of course, other window functions comprising a different number of window coefficients can also be defined and used within the framework of the embodiments according to the present invention. In this context, it should be taken into account that both the number of audio samples in the time domain per block and the number of blocks per frame, as well as the distribution of the blocks with respect to past samples and future samples, can be varied in a wide range of parameters.

[0223] Figure 12 depicts a comparison of a complex modulated low-delay filter bank window (CMLDFB window) 700 as depicted in Figure 11 and the original SBR QMF prototype filter 720 as used, for example , in the SBR tool according to MPEG standards. As depicted in FIG. 11, CMLDFB window 700 is again the synthesis window according to one embodiment of the present invention.

[0224] Although the window function 700 according to an embodiment of the present invention is clearly asymmetric as defined in relation to equation (30), the original SBR QMF prototype filter 720 is symmetric with respect to the indices n = 319 and 320, when the function window 700 as well as the SBR QMF 720 prototype filter are defined each with respect to each of the 640 indices. That is, with respect to equation (29), the index value no representing the index of the center of symmetry is given by 0 = 319.5 in the case of the SBR QMF 720 prototype filter.

[0225] Furthermore, due to the symmetry of the SBR QMF 720 prototype filter, also the center point nca and nce according to equations (35) and (36), respectively, are identical to the center of symmetry no. The energy value of the SBR QMF 720 prototype filter is 64.00 since the prototype filter is an orthogonal filter. In contrast, the clearly asymmetric window function 700 comprises an energy value of 78.0327 as noted above.

[0226] In the following sections of the present description, the SBR systems as described in connection with Figures 5 and 6, in which the SBR 610 decoder comprises the embodiments according to the present invention in the form of a bank of analysis filters such as filter bank 620 and a

ES 2 631 906 T3 embodiment according to the present invention in the form of a synthesis filter bank for the 640 synthesis filter bank. As will be described in more detail, the overall delay of an analysis filter bank according to the present invention using the window function 700 as shown in Figures 11 and 12 comprises an overall delay of 127 samples, while the tool SBR filter-based original SBR QMF provides an overall delay of 640 samples.

[0227] Replacing the QMF filter banks in the SBR module, for example in the SBR 610 decoder, by a complex value low delay filter bank (CLDFB) provides a delay reduction from 42 ms to 31.3 ms without introducing any degradation in audio quality or any additional computational complexity. With the new filter bank, the standard SBR mode (high quality mode) and the low power mode using only real value filter banks are supported, as shown in the description of the examples regarding the Figures 7 to 10.

[0228] In particular in the field of telecommunications and two-way communication, a low delay is of great importance. Although the enhanced low-delay AAC can already achieve a sufficiently low delay for 42 ms communications applications, its algorithmic delay is still higher than that of the low-delay AAC corecodec, which can achieve delays of up to 20 ms, and that of other audio codecs. telecommunications. In the SBR 610 decoder, the QMF analysis and synthesis steps still cause a 12 ms reconstruction delay. A promising method of reducing such delay comprises using a low-delay filter bank technique according to an embodiment of the present invention and replacing the current QMF filter banks with a corresponding low-delay version according to embodiments herein. invention. That is, additional delay reduction is achieved by simply replacing the ordinary filter banks used in the SBR 610 module with a complex low-delay filter bank in accordance with embodiments of the present invention.

[0229] For use in the SBR 610 module, the new filter banks according to embodiments of the present invention, which are also referred to as CLDFBs, have been designed to be as similar to the QMF filter banks originally used as may be possible. This includes, for example, the use of 64 subbands or bands, an equal length of the impulse responses and a support for dual speed modes equal to that used in SBR systems.

[0230] Fig. 13 depicts the comparison of the window shape CLDFB 700 according to an embodiment of the present invention and the original SBR QMF prototype filter 720. In addition, it illustrates the delay of the modulated filter banks, which can be determined by analyzing the overlap delay introduced by the prototype filter or window function in addition to the frame delay of the modulation core that has a length of N samples in the case of a system based on a DCT-IV. The situation represented in FIG. 13 refers again to the case of a synthesis filter bank. The window function 700 and the prototype filter function 720 also represent the impulse responses of the synthesis prototype filters of the two involved filter banks.

[0231] As regards the delay analysis for both the SBR QMF filter bank and the proposed CLDFB according to an embodiment of the present invention, in the analysis and synthesis only the overlap on the right side and on the left side of the modulation core, respectively, adds delay.

[0232] In both filter banks, the modulation core is based on a DCT-IV that introduces a delay of 64 samples, which is indicated in Figure 13 as delay 750.

In the case of the prototype filter SBR QMF 720 due to symmetry, the modulation core delay 750 is arranged symmetrically with respect to the center of gravity or the center of the corresponding prototype filter function 720, as indicated in figure 13 . The reason for this behavior is that the buffer of the SBR QMF filter bank needs to be filled to a point where the prototype filter function 720 makes the most significant contribution in terms of the corresponding energy values of the values prototype filter will be considered in processing. Of bido the form of function prototype filter 720, this requires that temporary storage is filled at least to the center point or center of gravity of the respective prototype filter function.

[0233] To further illustrate, starting from a fully initialized buffer of the corresponding SBR QMF filter bank, it is necessary to fill the buffer to a point where data processing will result in significant data processing, which requires the corresponding prototype filter or window function to have a meaningful contribution.

In the case of the prototype filter function SBR QMF, the symmetrical shape of the prototype filter 720 produces a delay, which is on the order of the center of gravity or center point of the prototype filter function.

[0234] However, since the delay introduced by the modulation core of the system based on a DCT-IV of N = 64 for samples is always present and the system also comprises a

ES 2 631 906 T3 delay of one block, it can be seen that the synthesis prototype for the SBR QMF introduces an overlap delay of 288 samples.

[0235] As mentioned above, in the case of the synthesis filter banks referred to in figure 13, said overlapping of the additional left part 760 causes the delay, while the overlapping of the right part 770 it refers to our passes and therefore does not introduce an additional delay in the case of a synthesis filter bank.

[0236] On the contrary, if one starts with an initialized memory of the CLDFB, according to an embodiment according to the present invention, both the synthesis filter bank and the analysis filter bank are able to provide in less time some Significant data compared to the SBR QMF filter bank, due to the way the window function presents. In other words, due to the shape presented by the analysis or synthesis window function 700, processing of samples by the window functions indicative of a significant contribution is possible in less time. Consequently, the synthesis prototype or the synthesis window function of the CLDFB introduces only an overlap delay of 32 samples, taking into account the delay introduced by the modulation core 750. The first part 420 or the first group 420 of window coefficients of the window function 700 according to an embodiment according to the present invention comprise, in a preferred embodiment according to the present invention, the 96 window coefficients corresponding to the delay caused by the overlap of the left part 760 in conjunction with the delay of the modulation core 750.

[0237] The analysis prototype function or the analysis filter bank enters the same delay value. The reason is because the analysis filter bank is based on the time-reversed version of the synthesis window function or prototype function. For this reason, the overlap delay is entered on the right side, comprising the same overlap delay value as in the case of the synthesis filter bank. Consequently, in the case of an original QMF prototype filter bank, a delay of 288 samples is also introduced, whereas for an analysis filter bank according to an embodiment according to the present invention, a delay of only 32 samples.

[0238] The table shown in figure 14a provides a summary of the delay values with different modification stages, assuming that the frame length is 480 samples and the sample rate is 48 kHz. In a standard configuration comprising an AAC-LD codec in conjunction with a standard SBR tool, the MDCT and IMDCT filter banks in dual mode cause a delay of 40 ms. Also, the QMF tool itself already introduces a delay of 12 ms. On the other hand, due to an SBR overlap, an additional delay of 8 ms is generated, so that the total delay of said codec is in the range of 60 ms.

[0239] In comparison, an AAC-ELD codec comprising low-delay versions of the MDCT and IMDCT introduces a delay of 30 ms in the dual method. Compared to the original QMF filter bank of an SBR tool, the use of a low delay filter bank and complex values according to an embodiment according to the present invention results in a delay value of only 1 ms compared to the 12 ms of the original QMF tool. By preventing the SBR overlap from occurring, the additional 8 ms overlap from a simple combination of an AAC-LD and an SBR tool can be totally avoided. For this reason, the higher performance, low-delay AAC codec is capable of introducing a total delay of 31 ms into the algorithm, rather than the 60 ms introduced by the simple combination described above. For this reason, it can be appreciated that the combination of the described delay value reduction methods effectively provides a 29 ms total delay value saving.

[0240] The table shown in figure 14b provides an additional overview of the total value of the codec delay caused by the original filter banks and those proposed in a system like the one represented in figures 5 and 6. The data and Values shown in Figure 14b are based on a 48 kHz sample rate and a core encoder frame size of 480 samples. Due to the dual method of an SBR system, as depicted and discussed in Figures 5 and 6, the core encoder effectively runs at a 24 kHz sample rate. Since the core encoder already introduces a 64-sample frame delay into the modulation core, it can be subtracted from the independent delay values of the two filter banks, as described in the context of Figure 13.

[0241] The table in Figure 14b emphasizes the fact that it is possible to reduce the overall delay of the low-delay AAC codec with higher performance comprising the low-delay versions of the MDCT and IMDCT (LD MDCT and LD IMDCT). . While only a total algorithmic delay of 42 ms can be obtained using the low-delay versions of the MDCT and IMDCT, as well as the original QMF filter banks, using low-delay filter banks and complex values according to the embodiments. according to the present invention instead of conventional QMF filter banks, the delay

ES 2 631 906 T3 total in the algorithm can be significantly reduced to only 31.3 ms.

[0242] In order to evaluate the quality of the filter banks according to the embodiments according to the present invention and the systems comprising one or more filter banks, audio tests have been carried out, from which It can be concluded that by means of the filter banks according to the embodiments according to the present invention, the audio quality of AAC-ELD is kept at the same level and the signal is not degraded, not even for the complex SBR mode, nor for real-value, low-power SBR mode. Thus, the delay-optimized filter banks according to the embodiments according to the present invention do not degrade the quality of the audio signal, even though they are capable of reducing the delay by a value greater than more than 10 ms. In relation to transitory periods, it is even observed that slight improvements can be obtained, although not statistically significant. The aforementioned improvements have been observed during audio tests performed with castanets and chimes.

[0243] In order to further verify that in the case of a 32-band filter bank according to an embodiment according to the present invention, the sampling works correctly in the same way for filter banks according to the present invention in comparison With the QMF filter banks, the following evaluation was carried out. First, a logarithmic sine sweep was analyzed with a sampled 32-band filter bank, adding the upper 32 bands initialized to zero. The result was then synthesized by a 64-band filter bank, sampled again, and compared to the original signal. Using a conventional SBR QMF prototype filter, a signal-to-noise ratio (SNR) of 59.5 dB is obtained. However, by means of a filter bank according to the present invention, an SNR value of 78.5 dB is obtained, which is indicative that the filter banks according to the embodiments according to the present invention have at least the same performance in sampled version compared to original QMF filter banks.

[0244] In order to show that such an approach based on non-symmetric and delay-optimized filter banks, which are used in embodiments according to the present invention, provides added value compared to a classic filter bank with a prototype symmetric, then asymmetric prototypes will be compared with symmetric prototypes that have the same delay value.

[0245] Figure 15a shows the frequency response in a far field illustration of a filter bank according to the present invention using a low delay window (graph 800), compared to the frequency response of a filter bank using a sinusoidal window that has a length of 128 coefficients (graph 810). Figure 15b shows an enlarged view of the near-field frequency response of the same filter banks using the same window functions as previously described.

[0246] By making a direct comparison of the two graphs 800 and 810, it can be seen that the frequency response of the low-delay filter bank according to an embodiment according to the present invention is notably better than the corresponding frequency response of a bank of filters that use a sinusoidal window of 128 coefficients and the same delay.

[0247] Likewise, figure 16a shows a comparison of different window functions with a total delay of 127 samples. The 64-band filter bank (CLDFB) features a total delay of 127 samples, including frame delay and overlap delay. A modulated filter bank with a symmetric prototype and the same delay would therefore present a prototype of length 128, as already illustrated in the context of Figures 15a and 15b. The reason is that such 50% overlap filter banks, for example MDCT, sinusoidal windows, or Kaiser-Bessel derived windows are generally a good choice for prototypes. Consequently, Figure 16a shows a view of the frequency response of a filter bank employing a low delay window as a prototype in accordance with an embodiment according to the present invention and compared with the frequency responses of alternative symmetric prototypes that have the same delay value. In figure 16a, in addition to the frequency response of the filter bank according to the present invention (graph 800) and the frequency response of a filter bank that uses a sinusoidal window (graph 810), as already shown in the figures 15a and 15b, two KBD windows are additionally represented that are based on the parameters α = 4 (graph 820) and α = 6 (graph 830). In both figures, in figure 16a and in the enlarged view of said figure, that is to say in figure 16b, it is clearly appreciated that a much better frequency response can be obtained with a filter bank according to an embodiment according to the present An invention presenting a non-symmetric window function or a prototype filtering function with the same delay.

[0248] In order to illustrate said advantage in a more generic way, in figure 17 two filter bank prototypes with delay values different from the filter banks described above are compared. While the filter bank considered in Figures 15 and 16 presents a total delay of 127 samples, which corresponds to an overlap of 8 blocks in the past and 0 blocks in the past.

ES 2 631 906 T3 future (CLDFB 80), figure 17 shows a comparison of the frequency responses of two prototypes of different filter banks with the same delay value of 383 samples. More specifically, figure 17 represents the frequency response of a non-symmetric prototype filter bank (graph 840), which is based on an overlap of 6 sample blocks in the time domain in the past and 2 sample blocks in the past. future time domain (CLDFB 62). On the other hand, figure 17 also represents the frequency response (graph 850) of a corresponding symmetric prototype filtering function that also has a delay of 383 samples. It can be seen that for the same delay value, by means of a non-symmetric prototype or window function an improved frequency response is obtained compared to a filter bank comprising a symmetric prototype filter or window function. In this way, the possibility of obtaining a better balance between delay and quality is demonstrated, as indicated above.

[0249] In Fig. 18, the temporal masking effect of the human ear is depicted. When a sound or tone is generated at the time indicated by a line 860 in figure 18, a masking effect related to the frequency of the tone or sound and the nearby frequencies is generated approximately 20 ms before the sound actually starts. . This effect is called pre-masking and constitutes one of the aspects of the psychoacoustic properties of the human ear.

[0250] In the situation illustrated in figure 18, the sound remains audible for approximately 200 ms until the time indicated by the line 870. During this period, a human ear sound masker is active, which is also called simultaneous masking. . After the end of the sound (indicated by the line 870), the masking in the near frequency of the tone decays slowly over a time interval of approximately 150 ms, as illustrated in figure 18. Such a psychoacoustic effect is also called post-masking. .

[0251] Figure 19 illustrates the comparison of the pre-echo behavior of a conventional HEAAC encoded signal and a HE-AAC encoded signal that is based on a low delay filter bank (CMLDFB) according to an embodiment according to the present invention. Figure 19a illustrates the original temporal signal of castanets, which has been processed with a system comprising a HE-AAC codec (HE-AAC = high efficiency advanced audio codec). The system output based on the conventional HE-AAC codec is illustrated in Figure 19b. If a direct comparison is made between both signals, the original temporal signal and the output signal of the HE-AAC codec, it can be seen that before the start of the castanets sound, in the area indicated by an arrow 880 the output signal of the HE-AAC codec comprises obvious pre-echo effects.

[0252] Figure 19c illustrates the output signal of a system comprising a HE-AAC codec based on filter banks containing CMLDFB windows according to an embodiment according to the present invention. The same original temporal signals indicated in figure 19a and processed using filter banks according to an embodiment according to the present invention reveal a significant reduction in the appearance of pre-echo effects just before the start of the castanets signal. , as indicated by arrow 890 in FIG. 19c. Due to the pre-masking effect, as described in the context of Figure 18, the pre-echo effect indicated by arrow 890 in Figure 19c is much more masked than the pre-echo effect indicated by arrow 880 in the case of a conventional HE-AAC codec. For this reason, the pre-echo behavior of the filter banks according to the present invention, which is also a result of the significant reduction in the delay value compared to conventional filter banks, means that the output adapts much better to the temporal masking properties and the psychoacoustics of the human ear. Consequently, as already indicated when describing the audio tests, the use of filter banks according to an embodiment according to the present invention can even lead to an improvement in relation to quality due to the reduction of the delay value. .

[0253] The embodiments according to the present invention do not increase computational complexity compared to conventional filter banks. Low delay filter banks use the same filter length and modulation mode as, for example, QMF filter banks in the case of SBR systems, so that the computational complexity is not increased. In terms of memory requirements, due to the asymmetric nature of prototype filters, the ROM (read-only memory) requirements for a synthesis filter bank are increased by approximately 320 words in the case of a filter bank based on in N = 64 samples per block and T = 10 blocks per frame. On the other hand, in the case of an SBR-related system, the memory requirements are further increased by another 320 words if the parsing filter is stored separately.

[0254] However, since the current ROM requirements for an AAC-ELD kernel are approximately 2.5 kilo words and for the SBR implementation another 2.5 kilo words, the

ES 2 631 906 T3 ROM memory only suffer a moderate increase of 10%. As a possible trade-off between memory and complexity, if low memory consumption is paramount, a linear interpolation can be used to generate the parsing filter from the synthesis filter, as described in the context of figure 3 and equation (15). Such an interpolation operation increases the number of instructions required by only about 3.6%. For this reason, by replacing the conventional QMF filter banks in the SBR module system with low-delay filter banks according to the embodiments according to the present invention, in some embodiments the delay value can be reduced by more than 10 more without degrading audio quality or noticeably increasing complexity.

[0255] For this reason, the embodiments according to the present invention refer to an analysis or synthesis window or to an apparatus or method for generating windows. On the other hand, an analysis or synthesis filter bank or a method for analyzing or synthesizing a signal using a window is described. Naturally, the computer program implementing one of the above methods is also described.

[0256] To put the embodiments according to the present invention into practice, a hardware implementation, a software implementation, or a combination of both can be carried out. The data, vectors and variables that are generated or received or otherwise stored for further processing, can be stored in different types of memories, for example direct access memories, buffer memories, read-only memories, memories non-volatile memories (for example, EEPROM memories, flash memories) or other types of memories, for example magnetic or optical memories. A storage location could constitute, for example, one or more memory units required to store or save the respective data, for example, variables, parameters, vectors, matrices, window coefficients or other information and data units.

[0257] Software implementations can be run on different types of computers, computer-like systems, processors, ASICs (Application Specific Integrated Circuits), or other integrated circuits (ICs).

[0258] Depending on certain implementation requirements of some embodiments of the methods according to the present invention, the embodiments of the methods according to the present invention can be implemented as hardware, as software or as a combination. The implementation can be carried out using a digital storage medium, in particular a CD disk, a DVD disk or another disk that has a control signal that can be read electronically and cooperates with a computer system, a processor or a programmable integrated circuit, so that an embodiment of the method according to the present invention is implemented. Generally, for this reason, an embodiment according to the present invention is a computer program with a certain programming code stored in a carrier accessible to a machine, executing said code so as to implement an embodiment of the method according to the present invention if the computer program runs on a computer, on a processor or on an integrated circuit. In other words, some embodiments of the methods according to the present invention constitute, for this reason, a computer program presenting a programming code for implementing at least one embodiment of the methods according to the present invention if the computer program it runs on a computer, on a processor, or on an integrated circuit.

[0259] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the first part of the analysis window function comprises a window coefficient having a maximum absolute value greater than 1.

[0260] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the window function of analysis presents an oscillating behavior.

[0261] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that all the window coefficients of the sequence of window coefficients have real value.

[0262] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the frame (120) of audio input samples in the time domain comprises a sequence of T blocks (130) of said type of samples, extending from the first to the last sample of the frame (120), each of the blocks comprising N input samples

ES 2 631 906 T3 of audio in the time domain, with T and N being positive integers and T greater than 4.

[0263] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the window generation comprises an element by element multiplication of the audio input samples in the time domain of the frame (120) with the window coefficients of the sequence of said coefficients.

[0264] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that any input sample of audio in the time domain is multiplied element by element with a window coefficient of the analysis window function, conforming to a sequence of audio input samples in the time domain and the sequence of window coefficients.

[0265] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that for any input sample of audio in the time domain of the frame (120) of the audio input samples in the time domain, exactly one sample is generated in the window.

[0266] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the corresponding window coefficient at an index of the window coefficients n = (T-3) · N has a value lower than -0.1, the index of the sequence of window coefficients being an integer in the range between 0 and N · T - 1, and the window coefficient used to generate the window of the last audio input sample in the time domain being the window coefficient corresponding to the N · T-1 index.

[0267] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the first part of the window coefficients have 3/2 · N coefficients, while the second part of window coefficients has (T-3/2) · N coefficients, from the sequence of window coefficients.

[0268] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the window coefficients c (n) meet the relationships indicated in the table in annex 3.

[0269] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the window coefficients c (n) meet the relationships indicated in the table in annex 2.

[0270] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) is adapted so that the window coefficients c (n) understand the values indicated in the table in annex 1.

[0271] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) or the apparatus (100) is adapted so that the analysis window function (190) comprises a first number of window coefficients obtained from a larger window function comprising a sequence of a larger second number of window coefficients, the window coefficients of the window function being obtained by interpolation of the window coefficients of the larger window function, and the second number being an even number.

[0272] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) or the apparatus (100) is adapted so that the window coefficients of the window function are linearly interpolated.

[0273] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) or the apparatus (100) is adapted so that the window coefficients of the window function of

ES 2 631 906 T3 analyzes are interpolated based on two consecutive window coefficients of the larger window function according to the sequence of window coefficients of the larger window function in order to obtain a window coefficient of the window function.

[0274] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the analysis window generator (110) or the apparatus (100) is matched with the objective of obtaining the window coefficients c (n) of the analysis window function based on the equation where n is an integer representing the index of the window coefficients c (n), and c2 (n) a window coefficient of the larger window function.

[0275] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, The apparatus (100) is adapted so that the current frame (120) of audio input samples in the time domain to be processed is generated by shifting (T-1) late blocks of the directly preceding frame (120) of audio samples. time domain audio input by one block heading towards the early time domain audio input samples and adding a block (220) of recent time domain audio samples as the block comprising the samples audio input late in the time domain of the current frame (120).

[0276] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the apparatus (100) is adapted so that the current frame (120) of audio input samples in the time domain x (n) to be processed is generated based on the offset of the audio input samples in the time domain xprev (n) of the frame (120) directly preceding the audio input samples in the time domain taking into consideration the equation

for a sample time or index n = 32, ..., 319, and the apparatus (100) being further adapted in order to generate the audio input samples in the time domain x (n) of the current frame (120 ) of time domain audio input samples by including 32 incoming time domain input samples below, conforming to an order of the incoming time-time audio input samples in the sense of decreasing the time or n sample rates for the audio input samples in the time domain x (n) of the current frame (120 ), starting at the instant or index of our n = 31.

[0277] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the calculator (170) comprises a time / frequency converter adapted to generate the subband values of audio so that all subband values based on a windowed sample frame (150) form a spectral representation of the windowed samples of the windowed sample frame (150).

[0278] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the time / frequency converter is adapted to generate complex value subband values.

[0279] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the calculator (170) is adapted to calculate an audio subband value for any sample of time domain audio input from a block (130) of time domain audio input samples, the computation of any audio subband value or any audio input sample in the time domain of a block (130) of time domain audio input samples based on the windowed samples of the frame in the window ( 150).

[0280] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the calculator (170) is adapted to calculate the audio subband values based on a multiplication of the window samples (150) by a harmonically oscillating function for any subband value and adding the multiplied window samples, a frequency of the harmonically oscillating function being based on a center frequency of a corresponding subband of the subband values.

[0281] In an apparatus intended to generate audio subband values in audio subband channels

ES 2 631 906 T3 according to an embodiment according to the present invention, the calculator (170) is adapted so that the harmonically oscillating function is a complex exponential function, a sine function or a cosine function.

[0282] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the calculator 170 is adapted to calculate the audio subband values wkl based on the equation for n = 0, ..., 63 and

= Σ z(n + j 64) j = 0 = Σ ORnn = 0

(K + 0.5) (2n - 95) for k = 0, ..., 31, where z (n) is a windowed sample corresponding to the index n, where k is a subband index, where 1 is an index of a block (180) of audio subband values and fosc (x) being an oscillating function that depends on a real value variable x.

[0283] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the calculator (170) is adapted so that the oscillating function fosc (x) is

= cos (x)

where i is the imaginary unit.

[0284] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the apparatus (100) is adapted to process a frame (120) of audio samples audio input in the real value time domain.

[0285] In an apparatus for generating audio subband values in audio subband channels according to an embodiment according to the present invention, the apparatus (100) is adapted to provide a signal indicative of the window function synthesis (370) to be used with the audio subband values or indicative of the analysis window function (190) used to generate the audio subband values.

[0286] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the apparatus (300) is adapted to generate a block (410) of audio samples in the time domain temporal, said block 410 of audio samples in the temporal domain comprising N audio samples in the temporal domain, N being a positive integer.

[0287] In an apparatus for generating time-domain audio samples according to an embodiment according to the present invention, the apparatus (300) is adapted so as to generate the block (410) of audio samples in the domain temporary, based on a block (320) of audio subband values comprising N audio subband values and so that the calculator (310) is adapted to calculate the sequence (330) of intermediate audio samples in the time domain comprising T · N intermediate audio samples in the time domain, where T is a positive integer.

[0288] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the synthesis window function is asymmetric with with respect to the window coefficient of the sequence.

[0289] In an apparatus intended to generate audio samples in the time domain according to a form

ES 2 631 906 T3 of embodiment according to the present invention, the synthesis window generator (360) is adapted so that the first part comprises a maximum value of all the window coefficients of the synthesis window function, presenting a value absolute greater than 1.

[0290] In an apparatus for generating time-domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the synthesis window function (370) present an oscillating behavior.

[0291] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the first part of coefficients comprises 3/2 -N window coefficients and the second part of coefficients comprises (T-3/2) -N window coefficients, where T is an index greater than or equal to 4, indicative of a number of blocks 340 comprised in the frame 330 of intermediate samples in the time domain.

[0292] In an apparatus for generating time-domain audio samples in accordance with an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the generation of the window of the sequence of intermediate samples in the time domain comprise an element-by-element multiplication of the intermediate samples in the time domain by a window coefficient.

[0293] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that any intermediate time domain sample is multiplied element by element by the window coefficient of the synthesis window function 370 according to the sequence of intermediate samples in the time domain and the sequence of window coefficients.

[0294] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the window coefficients of the window function synthesis (370) are real values.

[0295] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the window coefficients c (n) comply the relationships indicated in the table in annex 3.

[0296] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the window coefficients c (n) comply the relationships indicated in the table in annex 2.

[0297] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) is adapted so that the window coefficients c (n) comprise the values indicated in the table in annex 1.

[0298] In an apparatus for generating time-domain audio samples in accordance with an embodiment according to the present invention, the synthesis window generator (360) or apparatus (300) is adapted so that the function of synthesis window comprises a first number of window coefficients obtained from a larger window function comprising a sequence of a larger second number of window coefficients, the window coefficients of the window function being obtained by interpolation of the window coefficients of the larger window function, and the second number being an even number.

[0299] In an apparatus for generating time-domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) or apparatus (300) is adapted so that the coefficients of window of the synthesis window function (370) are linearly interpolated.

[0300] In an apparatus for generating time-domain audio samples in accordance with an embodiment according to the present invention, the synthesis window generator (360) or apparatus (300) is adapted so that the coefficients of window of the synthesis window function (370) are interpolated based on two consecutive window coefficients of the larger window function according to the sequence of window coefficients of the larger window function, in order to obtain a window coefficient of the window function.

[0301] In an apparatus for generating time-domain audio samples according to an embodiment according to the present invention, the synthesis window generator (360) or the apparatus (300) is adapted in order to obtain the window coefficients c (n) of the synthesis window function

ES 2 631 906 T3 based on the equation

where n is an integer that designates the index of the window coefficients c (n), and c2 (n) a window coefficient of the larger window function.

[0302] In an apparatus for generating time-domain audio samples in accordance with an embodiment according to the present invention, the calculator (310) is adapted to calculate the time-domain intermediate samples of the sequence of intermediate samples of the time domain, based on a multiplication of the audio subband values by an oscillating harmonic function and adding the multiplied audio subband values, the frequency of the oscillating harmonic function being based on a center frequency of the corresponding subband.

[0303] In an apparatus for generating time-domain audio samples according to an embodiment according to the present invention, the calculator (310) is adapted so that the oscillating harmonic function is a complex exponential function, a sine function or a cosine function.

[0304] In an apparatus for generating time domain audio samples in accordance with an embodiment according to the present invention, the calculator 310 is adapted to calculate time domain real value intermediate samples based on the complex value subband values.

[0305] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the calculator (310) is adapted to calculate the sequence of intermediate real value samples in the time domain z (i, n) based on the equation Zl, n = ~ Ñ SRe[Xi, kf°scG \ T+ i - y) · (* +!))} for an integer n between 0 and N · T-1, where Re (x) is the real part of the complex number x, where π = 3.14 ... the circular number and fosc (x) an oscillating harmonic function, so that

if the audio subband values supplied to the calculator are complex values, where I is the imaginary unit, and so that

if the audio subband values provided to calculator 310 are real values.

[0306] In an apparatus for generating audio samples in the time domain according to an embodiment according to the present invention, the calculator 310 comprises a time / frequency converter adapted to generate the sequence of intermediate samples in the time domain. , so that the audio subband values provided to the calculator 310 form a spectral representation of the sequence of intermediate samples in the time domain.

[0307] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the frequency / time converter is adapted to generate the sequence of intermediate time domain samples based on values complex value subband.

[0308] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the calculator (310) is adapted to calculate the sequence of intermediate samples in the time domain g (n ) from the audio subband values X (k) based on the equation v (n) = vprev(n - 2N) for an integer n between 20N - 1 and 2N,

ES 2 631 906 T3

for an integer n between 0 and 2N-1 and g (2N j + k) = v (4Nj + k) g (2Nj + N + k) = v (4Wj + 3W + k) for an integer j between 0 and 4 and for an integer k between 0 and N-1, where N is an integer that designates the number of audio subband values and the number of audio samples in the time domain, v being a vector of real value, where vprev is a real vector v of the direct previous generation of audio samples in the time domain, where i is the imaginary unit and π is the circular number.

[0309] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the calculator (310) is adapted to calculate the sequence of intermediate samples in the time domain g (n ) from the audio subband values X (k) based on the equation V(n) = Vprev (n “ 2N) for an integer n between 20N - 1 and 2N,

for an integer n between 0 and 2N-1 and

for an integer j between 0 and 4 and for an integer k between 0 and N-1, where N is an integer that designates the number of audio subband values and the number of audio samples in the time domain , where v is a real value vector, where vprev is a real vector v of the direct previous generation of audio samples in the time domain, and where π is the circular number.

[0310] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the output phase of an overlay adder (400) is adapted in order to process the intermediate samples in window in the time domain in an overlay mode, based on T consecutively provided blocks (320) of audio subband values.

[0311] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the output phase of an overlay adder (400) is adapted to provide the samples in the domain temporal out-i (n), where n is an integer that designates a sample index based on the equation

Tl out, (n) = X ζρ_λ) ιη + λ Ν * = or

ES 2 631 906 T3 where zl, n is an intermediate sample in a window in the time domain corresponding to the sample index n and a frame or sequence index 1 between 0 and T - 1, where 1 = 0 corresponds to the last frame or sequence and values less than 1 to previously generated frames or sequences.

[0312] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the output phase of an overlay adder (400) is adapted to provide the samples in the domain temporal out (k) based on the equation out (k) = w (nn + k), n = 0 where w is a vector that includes the intermediate samples in the window in the time domain and k is an integer that designates an index between 0 and (N-1).

[0313] In an apparatus for generating time domain audio samples according to an embodiment according to the present invention, the apparatus (300) is adapted to receive a signal indicative of the analysis window function (190 ) used to generate the audio subband values, or indicative of the synthesis window function (370), which should be used to generate the audio samples in the time domain.

[0314] According to an embodiment according to the present invention, an encoder (510) comprises an apparatus (560) for generating audio subband values in audio subband channels according to an embodiment according to the present invention.

[0315] According to an embodiment according to the present invention, an encoder (510) further comprises a quantizer and an encoder (570) coupled to the apparatus (560) for generating audio subband values and adapted to quantize and encode the values audio subband values output by the apparatus (560) and output the quantized and encoded audio subband values.

[0316] According to an embodiment according to the present invention, a decoder (580) comprises an apparatus (600) for generating time domain audio samples according to an embodiment according to the present invention.

[0317] According to an embodiment according to the present invention, a decoder (580) further comprises a decoder and a decoder (590) adapted to receive encoded and quantized audio subband values, coupled to the apparatus (600) to generate samples audio in the time domain and adapted to provide the decoded and dequantized audio subband values as the audio subband values to the apparatus (600).

[0318] According to an embodiment according to the present invention, an SBR encoder (520) comprises an apparatus (530) intended to generate audio subband values in audio subband channels, based on a frame of time domain audio input samples provided to the SBR encoder (520) and an SBR parameter extraction module (540) coupled to the apparatus (530) for generating audio subband values and adapted to extract and outputting SBR parameters based on the audio subband values.

[0319] According to an embodiment according to the present invention, a system (610) comprises an apparatus (620) for generating audio subband values from a frame of time domain audio input samples provided to the system (610); and an apparatus (640) for generating time domain audio samples based on the audio subband values generated by said apparatus (640) to generate audio subband values.

[0320] According to an embodiment according to the present invention, a system (610) is an SBR encoder.

[0321] According to an embodiment according to the present invention, a system additionally comprises a high frequency generator (630) interconnected between the apparatus (620) for generating audio subband values and the apparatus (640) for generating audio samples in the time domain and adapted to receive SBR data adapted to modify or add audio subband values. Based on said SBR data and audio subband values from apparatus 620 to generate audio subband values.

[0322] With respect to all the apparatuses and methods according to some embodiments according to the present invention, depending on the details of the implementation, to obtain the window coefficients based on the values given in the tables of Annexes 1 and 3 , additional sign changes can be implemented with respect to the window coefficients corresponding to indices 128 to 255

ES 2 631 906 T3 and 384 to 511 (multiplication by factor (-1)) according to equations (16a) and (16b). In other words, the window coefficients of the window function are based on the window coefficients indicated in the table in Annex 1. In order to obtain the window coefficients of the window function represented in the figures, it is necessary to multiply by (+1) (that is, there is no sign change) the window coefficients in the table corresponding to the indices 0 to 127 , 256 to 383 and 512 to 639, and the window coefficients corresponding to the indices 128 to 255 and 384 to 511 must be multiplied by (-1) (that is, there is a change of sign) in order to obtain the window coefficients of the rendered window function.

Therefore, accordingly, it is necessary to take into account the relationships indicated in the table in annex 3. To express it again in other words, the window coefficients, as indicated in the tables in annexes 1 to 4, are They can modify according to equations (16a) and (16b).

[0323] It should be noted that within the framework of the present application, in an equation based on another equation, the introduction of additional delay values, additional factors, additional coefficients or the introduction of another simple function is understood. Additional constants, simple constants, constant addends, etc. can be omitted. Also included are algebraic transformations, equivalent transformations, and approximations (for example, a Taylor approximation) that do not modify the result of the equation at all, or do so in an insignificant way. In other words, both small modifications and transformations that lead to essentially identical results are included when an equation or expression is based on another equation or expression.

[0324] An apparatus for generating complex value audio subband values in audio subband channels according to an embodiment according to the present invention, comprises an analysis window generator 110 in order to generate a window of a frame 120 of audio input samples in the time domain, arranged in a time sequence extending from an early sample to a late sample, and using an analysis window function 190 comprising a sequence of window coefficients in order to obtain windowed samples. The analysis window function 190 is composed of a first group 200 of window coefficients, which comprise a first part of the sequence of window coefficients, and of a second group 210 of window coefficients, which comprise a second part of the window coefficient sequence. The first part comprises fewer window coefficients than the second part. A global energy value of the window coefficients in the first part is higher than an overall energy value of the window coefficients of the second part, the first group of window coefficients being used for the generation of the window of the late samples in the time domain, and the second group of window coefficients being used for the generation of the window of the early samples in the time domain. The apparatus further comprises a calculator 170 for calculating the audio subband values using the windowed samples, said calculator 170 comprising a time / frequency converter adapted to generate the audio subband values, so that all the subband values based on a frame 150 of windowed samples forms a spectral representation of the windowed samples of the frame 150 of windowed samples, and adapting said time / frequency converter to generate complex value audio subband values.

[0325] In said type of apparatus intended to generate complex value audio subband values 100 according to an embodiment according to the present invention, the analysis window generator 110 can be adapted so that the window coefficient corresponding to the index n = (T-1) · N has an absolute value between 0.9 and 1.1, the index of the window coefficient sequence being an integer between 0 and N · T - 1, and the window coefficient used to generate the window of the last audio input sample in the time domain of frame 120 being the window coefficient corresponding to the N · T-1 index, so that the analysis window generator 110 is adapted so that the time domain audio input sample frame 120 comprises a sequence of T blocks 130 of said sample type, extending from the first to the last audio input sample in the time domain of frame 120, every block comprising N audio input samples in the time domain, and T and N being positive integers and T greater than 4. In an apparatus for generating complex value audio subband values 100 in accordance with an embodiment according to the present invention, the analysis window generator 110 may be further adapted so that the window coefficient corresponding to the coefficient index of window n = N · T - 1 has an absolute value less than 0.02.

[0326] In an apparatus for generating complex value audio subband values 100 according to an embodiment according to the present invention, the analysis window generator 110 can be adapted so that the window generation comprises a multiplication of the audio input samples in the time domain of the 120 x (n) frame to obtain the windowed samples z (n) of the windowed frame, based on the equation

ES 2 631 906 T3

where n is an integer that designates the index of the sequence of window coefficients in the range between 0 and TN-1, where c (n) is the window coefficient of the analysis window function corresponding to the index n, where x (N · T-1) is the last audio input sample in the time domain of a 120 frame of samples of that type, so that the analysis window generator 110 is adapted so that the time domain audio input sample frame 120 comprises a sequence of T 130 time domain audio input samples, extending from the first up to the last audio input sample in the time domain of frame 120, every block comprising N audio input samples in the time domain, and T and N being positive integers and T greater than 4. In an apparatus 100 for generating complex value audio subband values in accordance with an embodiment according to the present invention, the analysis window generator 110 may be further adapted so that the window coefficients c (n) comply with the relationships indicated in the table in annex 4.

[0327] In an apparatus for generating complex value audio subband values 100 according to an embodiment according to the present invention, the apparatus 100 may be adapted for the purpose of using an analysis window function 190 that is a time-reversed or index-reversed version of synthesis window function 370 to be used for audio subband values.

[0328] An apparatus 300 for generating time domain real value audio samples according to an embodiment according to the present invention, comprises a calculator 310 for calculating a sequence 330 of time domain intermediate samples from audio subband values in audio subband channels, said sequence comprising the intermediate early samples in the time domain and the late samples in the time domain, said calculator 310 comprising a frequency / time converter adapted to generate the sequence of intermediate samples in the time domain, so that the audio subband values provided to the calculator 310 form a spectral representation of the sequence of intermediate samples in the time domain . The frequency / time converter is adapted to generate the sequence of intermediate samples in the time domain based on complex value subband values. The apparatus for generating real value audio samples in the time domain 300 according to an embodiment according to the present invention, further comprises a synthesis window generator 360 for generating a window of the sequence 330 of intermediate samples in the domain time using a synthesis window function 370, comprising a sequence of window coefficients in order to obtain intermediate samples in the window time domain, said synthesis window function 370 being composed of a first group 420 of window coefficients comprising a first part of the sequence of window coefficients, and of a second group 430 of window coefficients, which comprise a second part of the sequence of window coefficients, the first part comprising fewer window coefficients than the second part. A global energy value of the window coefficients in the first part is higher than a global energy value of the window coefficients of the second part, the first group of window coefficients being used for the generation of the window of the late samples in the time domain, and using the second group of window coefficients for the generation of the window of the intermediate early samples in the time domain. The apparatus 300 further comprises an output stage from an overlay adder 400 to process the intermediate windowed samples in the time domain to obtain the samples in the time domain.

[0329] In an apparatus for generating real value audio samples in the time domain 300 according to an embodiment according to the present invention, the window coefficient corresponding to the index n = N may have an absolute value between 0, 9 and 1,1, the index n of the sequence of window coefficients being an integer between 0 and TN - 1, and the window coefficient used to generate the window of the last audio input sample in the time domain being the window coefficient corresponding to the index n = 0, where T is an integer greater than 4, which designates the name of blocks included. in frame 330 of intermediate samples in the time domain, the apparatus 300 being able to be adapted in order to generate a block 410 of audio samples in the time domain, said block 410 of time domain audio samples comprising N time domain audio samples, N being a positive integer. In an apparatus for generating real value time domain audio samples 300 in accordance with an embodiment according to the present invention, the synthesis window generator 360 can be further adapted such that the window coefficient corresponding to the index n = 0 has an absolute value less than or equal to 0.02.

[0330] In an apparatus for generating time domain real-value audio samples 300 according to an embodiment according to the present invention, the synthesis window generator 360 can be adapted so that the window coefficient corresponding to the index n = 3N is less than 48

ES 2 631 906 T3

0.1, the apparatus 300 being further adaptable to generate a block 410 of audio samples in the time domain, said block 410 of audio samples in the time domain comprising N audio samples in the time domain, N being an integer positive.

[0331] In an apparatus for generating time domain real-value audio samples 300 according to an embodiment according to the present invention, the synthesis window generator 360 can be adapted so that the generation of the window comprises a multiplication of the intermediate samples in the time domain g (n) of the sequence of intermediate samples in the time domain in order to obtain samples in window z (n) of the plot in window (380), based on the equation z (n) = g (n) · c (T · N - 1 - n) for n = 0, ..., T · N - 1. In a device designed to generate audio samples of value In real time domain 300 according to an embodiment according to the present invention, the synthesis window generator 360 can be further adapted so that the window coefficients c (n) comply with the relationships indicated in the table in annex 4.

window coefficients c (n) meet the relationships indicated in the table in annex 4.

ES 2 631 906 T3

ES 2 631 906 T3 w (72] = 1.035694432087 «6e + 000 w [73] = 1.037683165297586e + 000 w! 74] = E039227995800217e + 000 w (75] = 1.040349586463588e + 000 w | 76) = 1.041086497214721e + 000 w | 77] = 1.041443375950143e + 000 w [78) = 1.041434355650865e + 000 w [79] = 1.041043184216171e + 000 w [80] = 1.040262316588456e + 000 w [81) = 1.039061496136853e + 000 w [82J = 1.021374223001579 e + 000 w [83] = 1.035311720204252e + 000 w [84] = 1.03271295217712le + 000 w (85] = 1.029600494883906e + 000 w (86J = 1.025966756910904e + 000 w [87] = 1.021798805583990e + 000 w | 88) = 1.017100128250049e + 000 w [89J = 1.011867706519706e + 000 w [90] - 1.006109248754940e + 000 w [91] = 9.998235752401580e-001 w [92] = 9.9303798546798 [93) = 9.857387823493258e — 001 w [94] = 9.779405164766706e-001 w [9S] = 9.696426101291272e-001 w [96J = 9.608519516143015e-001 w [97] = 9.515674613550604e-001 w [9.4779 w [98) -001 w [99] = 9.315442093447622e-001 w [100] «9.208194746232827e-001 w [101] = 9.096310803629866e-001 w [102]» 8.979959173503500e-001 w [103] = β.B59232320517536e-001 w | 104] = 8.734366852542127e-001 w (105] = 8.605542791988831e-001 W1106] = 8.472987145504696e — 001 w | 107] = 8.336863467961255e-001 w [108] w (109] = 8.054701312929008e — 001 w (110) = 7.908995350037713e-Q01 w [lll) = 7.760385598209244e-001 w (112) = 7.609051036128973e-001 WÍ113] - 7.455111681431031e-001 WÍ114] w (115] = 7.140087729493950e-001 w (116J - 6.979336851549095e-001 w (117] »6.816667882498023e-001 w [118] = 6.652304141388827e-OO1 w [119] = 6 „486437667370537e-001 w [120J = 6.319284031798550e-001 w [121) = 6.151031151692835e — OO1 w [122] = 5.981877665956570e-0O1 w [12322] = 5.81 £ 124J = 5.641522833259215e-001 w [125] = 5.470652177576862e-001 w [1261 = 5.299509559653194e-001 w [127] = 5.128557121424191e-001 w [128) = —4.9561754214.714453e —001126 w [12966] e-001 w [130] = -4.609828932783459e-001 w [131} - -4.437530233023859e-001 w [132] = -4.265950246465440e-001 w [133) = -4.095160467543179e-001 w [134) = -3.925409172155113e-0Ol w [135J = -3.756821671788237e-001 w (136] = -3.589626517817934e-001 w [137] = -3.423942311297658e-001 w [] = 138 3.259993851O88293e-OOl w (139] = -3.097961805973821e-001 w [140] = -2.937724988593393e-001 w [141J = -2.779637821990255e-001 w [142] = -2.623749159488041e-001 w (143] B299603623e-001 wI144] = —2.318815476758375e — 001 wf145] = -2.169925682529340e-001 w [146] = -2.023548005388463e-001 w [147) = -1.879711746686855e-001

ES 2 631 906 T3 w [148J = -1.738542127021508e — 001 «[149] = —1.600061812296078e-001 w [150 | = —1-464389150679625e-001 «[151] = -1.331544923127771e-001 w [152J = —1 * 201628679722633e — 001 w (153] = -1.074630704470568e-001« [154] = -9.50696695963251le-002 «[155 ] = -8.298103104739203e-002 «{156] = -7.120356992726613e-002 w (157] = -5.973741829536090e-002« 1158] = -4.85900576701681le-002 «[159] = -3.775928110298274e — = -2.726484300166575e-002 «[161] = -1.711323992709580e-002« [162] = -7.298197371320593e-003 «[163] = 2 * 18425692935678 le — 003« [164] = 1.132324047372148e — 002 «[165] - 2.01223699075490Oe-OO2« [166] - 2.857528272530154e-002 w (16728] = 3.66817691e -002 «(168) = 4.439683978044157e-002« [169] = 5.17796476887O7B7e-OO2 «[170] = 5.881296711410786e-002« [171] = 6.550209046893848e-002 «[172] = 7.184073822817207e-7.184073822817207e- ] = 7.783299328224960e-002 «[174] = 8.347150698567406e — 002« [175] = 8.875756217893037e-002 w [176] = 9.368651761350569e-0Q2 «[177] = 9.8 2 62 5112 9 4 6 5 6 2 4 e - 0 02 w (178] = 1.024804711677230e-001 «[179] = 1.063454554357498e-001« [180] = 1.098551252869576e-001 «[181] = 1.130180022553412e-001« [182] = 1.158358935177899e-001 «[183] = 1.183233335449968e — 001« [184] = 1.204854506722672e-001 «[185] = 1.223371395264402e-001« [186] = 1.238868653862843e — 001 «(187) = 1.25147725800 w [188] = 1.261262023246478e — 001 «[189] = 1.268280540744526e-001« [190] = 1.272498700590511e-001 «[191] = 1.273590703506806e-001« [192] = 1.274567595465545e-001 «[193] = 1.275561350483646e — 001« [194] = 1.273648326872248e — 001 «[195] = 1.269415772180714e-001« [196] = 1.262995646340671e-001 «[[194] = 1.273648326872248e — 001« [195] = 1.269415772180714e-001 «[196] = 1.262995646340671e-001« [[194] = 1.273648326872248e «[1971867 [198] = 1.244269583009826e-001 «[199] = 1.232131583108813e-001« [200] = 1.218183974942866e-001 «[201] = 1.202545652840080e-001 w [202] = 1.1852431068Θ9108e-001« [203] = 1.166399102636992e-001 w [2O4J = 1.146042249339280e-001 «[205] = 1.124296184976912e-001 w [206] = 1.101215600923314e-001« [207] = 1.076972053405737e-001 «[208] = 1.051641975499523e-001 w [209] = 1.025397604985405e-001 w [210] = 9.9829579343462S4e-002« [211] = 9.705239536075722e-00216 «1224112 = 9.468299« 1224112 ] = 9.133590931873967e-C02 «(2141 = 8.841813387276727e-002 w (215] = 8.547715661443602e-Q02« 1216] = 8.251962055343706e-002 w (217] = 7.955570759229536e-002] 002 7.955964 —756e-00234e = 7.6764S —756756e-00234E «[219] - 7.360559211914287e-002« [220] = 7.064948295960993e-002 «[221] = 6.771675107480543e-002« [222] = 6.48044Θ450935215e-002 «1223] = 6.192692754258131e-002

ES 2 631 906 T3 «[224] = 5.911363249658311e-002« [225] = 5.637219228757212e-D02 «[226] = 5.368313072045600e-002« [227] = 5.105620793438655e — 002 «[228] = 4.8492Oβ4995895 «[229] = 4.599068181839981e-002« [230] - 4.355568588898841e-002 «[231] = 4.125570251909672e-002« [232] = 3.907137550527191e-002 «[233] = 3.696342556744636e-002 3.493300140502248e-002 «[235] = 3.298151059524886e-002 w [2361 = 3.110861245410919e-002 w [237] = 2.931525594774175e — 002« (238) = 2.760090729801069e-002 w [239] = 2.597956638848436e — 002 «[240] = 2» 443433592149451e-002 w [241] = 2.296470793543091e-002 «[242] = 2.156304510969632e-002« 12431 = 2.023524610221679e-0037e-1.49744e 002 "1245] = 1.778248750467421e-002" E2463 = 1.665187994388476e-002 "[247] = 1.557759513377242e-002 w [248J = 1.4S6206586604537e-002" 1249] = 1.361072086117313e-00264] «[251] = 1.186210743261470e-002« [252] = 1.1O6958962776399e-002 «[253] = 1.033126278863177e-002« [254] = 9.640298325700842e-003 «[255] = 8.996371481700806e-003« [256] = -8 »407748878436545e-003« [257] = -7.876393114319395e-003 «[258] = -7.3éO543918629573e — = 003 -« [ 6.925141135202262e-003 «[260] = -6.500502521462604e-003 w [261] = -6„ 109178606718115e-003 «[262] = -5.741103163221257e-003« 1263] = -5.39456960891996538e —10460 12608919665e = 003.06406508 e-003 w [265] = -4.754191853611012e-003 «[266] = -4.448993249380505e — 003« [267] = —4.133639756278191e — 003 «[268] = -3.311612348723333e-003« [269] = -3 - 5055313189SO422e — 003 «[270] = -3.209092846617964e — 003 w [271) = -2.92715943674 0159e-003« [272] = -2.653818578698405e-003 w [273] = -2.396404 01396145237e-003 «[271.2715963e-003" S89273e-003 «[275] = -1.924844672908215e — 003« [276] = -1.699160580023900e-003 w [277] = -1.480542563288228e-003 «[278] = -1 - 283280633901446e — 003« [279] = - 1.131859661378862e-003 «[280] = -9.730460256556873e-004« [281] = -7.677634115875747e-004 «[282] = -5.599347984905645e-004« [283] = -3.337966579125254e-004 -9.099722643476421e-OO5 w [285] = 1.49823162181604le-004 «[286] = 4» 36644701211681le-004 «[287] = 6.307841647560053e-004« [288] = 6.150316826138937e-004 «B. 990255827053560e-004 «[290] = 1.232134364570107e-003« [291] = 1.471167206249042e-003 «[292] = 1.697652664777771e-003« [293] = 1.98582500255428654e —00372 «[294] = 2.10372« [294] = 2.10372 295] = 1.812176023993582e — 003 «1296] = 1. 3 4 4 6 57 2 62 814 * 7 9 3 e - 00 3 «[297] = 9 - 373975348172919e-004« 1298] = 5 * 621720998949145e-0D4 w [299] = 2.0484985S2413189e — 004

ES 2 631 906 T3 w[300) = —2.004822830002534e-004 w[301) = -6.169854804735951e-004 w[302) = -1.061498982103114e-003 w[3031 = —1.594860949611097e-003 w[304] = —2.124647831574725e—003 w[305] = -2.621537051750861e-003 w(306] = —3.064311083207632e-003 w(307] = -3.460362845825662e-003 w(308] = -3.794425324215804e-003 w(309J = -4.091032597247918e-003 w[310] = -4.369553676668050e-003 w[311] = —4.554811297024067e-003 w[312] = -4.663276675479689e-003 w[313) = -4.722567636185647e-003 w[314] = -4.704321497976561e-003 w(315) = -4.636227793039124e-003 w[316] = -4.517190210387324e-003 w[317] = -4,351667566540186e-003 w[318] = -4.135130493071822e-003 w[319) = -3.870851645947402e-003 w[320] = -3.597475533950260e-003 w[321J = -3.318857985461042e-003 w[322) = -3.000422543655664e-003 w[323J = -2.658042081080524e-003 w[324] = -2.292813563887493e-003 w(325] = —1.914114740669928e-003 w[326J = -1.525818616748839e-003 w(327] = -1.156680209049319e-003 w[328] = -7.804546272743493e-004 w[329) = -4,268574601396473e-004 w[330] = -1.324291707264515e-004 w[331J = 1.218226450050751e-004 w(332] = 3.189336138130849e-004 w[3331 = 4.749931197951235e-004 w[3341 = 5.970696819774243e-004 w(335] = 6.673250213055329e-004 t

w(336] = 6.887783835812338e-004 w(337J = 6.766320515830324e-004 w[338J = 6.944123176012471e-004 w[339] = 7.139919634325070e-004 w{340) = 7.154123487609100e-004 w[341) = 7.376101027486600e—004 w(342J = 6.976561203768226e-004 w[343) = 5.721223454434728e—004 w[344] = 2.934875643581191e-OO4 w[345) = 1.092526149391273e-004 w[346J = 6.415402443848103e—004 w[347] = 1.194730618383423e-003 w[348] = 1.557112059887280e—003 w[349] = 1.891971801393744e-003 w[350] = 2.225524159129023e-003 w(3511 = 2.530906981099261e-003 w[352J = 2.719749515067397e-003 w[353] = 2.729136737522100e-003 w[354] = 2.703019498899013e-003 w[355] = 2.630471852319136e-003 w[356] = 2.470456304276468e-003 w(357] = 2.239142906871446e-003 w[358] = 2.033465291493264e-003 w[359] « 1.948069005335563e-003 w[360] = 1.725029670030533e-003 w[361J = 1.417366709895927e-003 w[362J = 1.127141815310061e-003 w[363] = 8.089811988213151e—004 w[364) = 4.708009521678285e-004 w[365] = 7.882620739833088e-005 w[366] = -2.998739993995956e-004 w[367] = -4.733148292475610e-004 w[368) = -5.791145447913150e-004 w[369] = -6.754935404082003e-004 w(370] = -8,029620210721900e-004 w[371] = -9.726698841994444e-004 w[372) = -1.196637962311630e-003 w[373] = -1.292865844760059e-003 w[374] = -1.146268465739874e-003 w[375) = -1 - 04059805507447le-003

ES 2 631 906 T3

ES 2 631 906 T3 w[452] = -1.728672507238372e-004 w[453] = --1,885286127508226e-004 w[454] = -2.078299015661617e-004 w[455] = -2.123671573189573e-004 w[456] = -2.415166002501312e—Ó04 w[457] = -2.217025456251449e-004 w[458] = -9.907630821710970e-005 w[459] = —8.039231481768845e—005 w[460] = -7.93450941772240Oe—005 w[461J = -5.874199358780108e-005 w[462] = -5.449816072329412e-005 w[463] = -4.489491034408147e-OO5 w[464] = -3.498285982359981e-OO5 w(465] = -1.748284921486958e—005 w[466] = -9.075430772832575e-006 w[467] = —1.052707430241351e—005 w[468] = -6.538878366985722e-006 w[469] = 2.206341308073472e-005 w[470] - 1.769261935287328e—004 w[471] = 6.418658561385O58e—005 w[472] = -8.882305312548962e-005 w[473] = -1.721347222211949e-005 w[474] = -6.093372716385583e—005 w[475] = -7.679955330373515e—005 w[476] = 7.194151087015007e-005 w[477] = 7.245095937243279e-005 w[478] = 7.870354371072524e-005 w[479] = 5.822201682995846e-004 w[480] = 2.666444630171025e—004 w[481] = 7.872592352725688e—005 w[482] = 7.095886893185526e—005 w[483] = 5.643103068471008e-005 w[484] = 6.904415362098980e-005 w[485] = 4 - 694251739991356e—005 w[486] - 3.367998333617662e-005 w[487] = 6.481921021601837e-005 w[488] = 6.582328030188790e-005 w[489] = -4.256442530773449e-005 w[49O] = 4.939392400898679e-005 w[491] = 5.272982009116034e-005 w[492] = 4,005269212731273e-005 w[493] = 2‘. 46187 6679726978e-005 w[494] = 4.469729032194765e-006 w[495] = 3.79851973l621893e-007 w[496] = 1.374896222030490e-006 w[497] = 3.965363805500215e-006 w[498] .= 7.30058886393478Oe—006 w[499] = 1.168894474770061e-005 w(500] = 8.563819899447630e-006 w(501] = 8.975977837330335e-006 w[502] = 2.800455533708622e-005 w[503] = 2.015445311139832e-005 w[504) = 1.125134651175812e-005 w[505] = 5.869707265615299e-006 w[506] = 1.013259758329981e-005 w[507] = 1.088325131492173e-005 w[508] = 7.167101260771279e-006 w[509] = 4.840577540089826e-006 w[510] = -1.469933448634890e-005 w[511] = -8.010079089953001e-006 w[512] = —3.299004046633323e—005 w[513] = -4.373302115187172e-005 w[514] = -3.177468256997963e-005 w[515] = -2.9768240361B2567e-005 w[516] = -2.464228015326852e-005 w[517] = -1.606050838620834e-005 w[518) = -6.261944255489322e-006 w[519] = 4.591009581217994e-007 w[520] = 1.395220723090848e-005 w[521] = 1.622786214398703e-005 w[522] = -2.043464113212971e-006 w[523] = -1.653463907257247e-006 w[524] = -1.551250801467300e-008 w[525] - -1.907927361317977e—006 w(526) = -9.607068622268791e-007 w[527] = -4.636105364510011e-007

ES 2 631 906 T3 w[528] = -2.765649762593200e-007 w[529]· = -1.922074581855119e-006 w[530] = -9.897194091136331e-007 w[531) = —7„873304717454037e—008 w[532] = 2.945239208477290e—008 w[533] = —2.757610624807679e—006 w[534] = -1.402925247695813e-005 w[535] = -9.388962780643742e-006 w(536] = 2.068297421740023e-005 w[537] = 1.496435902895210e—007 w[538] = 6.757014945674924e-009 w[539] = -2.778618354859Θ61e-007 w[540] = -1.569003268449803e—006 w[541] = -1.089500601234349e-006 w[542] = —9.870547653835426e—007 w[543] = 3.867483283567218e—005 w[544] = -1.232693496472088e-005 w[545] = 9.464782951082177e-007 w[546] = 8.254429452O94225e-007 w[547J = 4.883304950437536e-007 w(548] = -2.066961713890010e-007 w[549] = 5.158212471036245e—009 w[550] = 2.267731106642486e-007 w[551] = —4.88084455071395 le—008 w[552] = 3.3616821B3852576e-0Q6 w[553J = 4.677015459111491e—006 w[554] = 2.820292122791583e-008 w[555] = 5„143614846654519e—007 w[556] = 3.818588614859347e-009 w[557] = 1.737276553950212e-007 w[558] = 1.676022048145804e-007 w[559] = -2.986488593070417e-009 w[560] = -1.409927495646886e-008 w[561] = -6.977078748707401e-008 w[562] = -1.280675520205100e-008 w[563] = -2.222072007942510e-009 w[564] = -1.775191290895584e-009 w[565] = -1.686136654621906e-009 w[566] = 5 *818594642226675e-006 w[567] = 2.150883991167946e-006 w[568J = 2,714879009950152e-007 to[569] = -2.567964804401197e-008 w[57Q] = 2.041128570435378e—006 w[571] = 3.262753594084781e—006 w[572] = 3.567581483749161e-006 w[573] = 4.083718802566134e-006 w(574] = 5.364807253588177e-006 w[575] = 4.178050149840223e-006 w[576] = 5.189086332701670e-006 w[577] = 3.357218747491756e-006 w[578] = 6,310207878018869e-006 w[579] = 5.924001540927652e-006 w[580j = 5.161606640346293e-006 w[581] = 3.377814811745950e-006 w[582] = 1.323267689777069e-006 w[583) = -1.074716688428712e-007 w[584] = —3.561585382456484e—006 w[585] = —4.518603099564185e—006 w[586] = 7.301956971603966e-007 w[587] = 5.89190477516102Se-007 w[588] = 2.801882088134371e-Q08 w[589] = 6.322770332405526e-007 w[590] = 2.542598385847351e-007 w[591] = 1.272704908592385e-007 w[592] = 8.226599990523664e-008 w[593] = 5.43371876878914Oe—007 w[594] = 4.211177232106135e-007 w[595] = 3.552991527S551B0e-008 w[596] = -1.3989l3l09540774e-008 w[597) = 1.356727552196146e-006 w(598J = -1.706941020342299e-005 w[599] = 1.013575160981381e-005 w[600] = -2. 285562946018590e-005 w[601] = -8.908041185396514e-008 w[602] = -9.597515277415496e-009 w[603J = -3.225913527455964e-007

ES 2 631 906 T3 w[604] = 1.070242712585309e-006 w[605] = 6.293002327021578e-007 w[606] = 3.575650976036433e—007 w[607] = 2.722295965060517e-005 w[608] = 8.676848186676886e-006 w[609] = 3.428660858940255e-007 w[610] = 4.767793949944890e—007 w[611J = 3,330981930777764e-007 w[612] = 2.399696144635756e—007 w[613] = 7.326611439066549e-009 w[614) = 1.349943693297681e—007 w[615] = -5.393S55749348494e-008 w[616] = 3.6290670655241430-006 w[617] = -5.690530948134642e-006 w[618] = 1.387566465624550e-008 w[619] = 2.443085172403935e-007 w[620] = 1.723217058490933e-009 w[621] = 7.391973323448250e—008 w[622] = 5.303527922331415e-008 w(623] = —8.883499047404846e—010 w[624] = -3.870536804891648e—009 w[625] = —1.846547564287500e—008 w[626j = -4.244090917065736e-009 w[627] = -4.013524925634108e-009 w[628] = -6.325664562585882e-010 w[629] = -6.025110605409611e-010 w[630] = 1.620171502086309e-006 w[631] = 5.4905699.54646963e-007 w(632] = 6.355303179925355e-008 w[633] = -5.426597100684762e—009 w[634] = 4.292861814894369e—007 w[635] = 6.834209542421138e—007 w[636] = 7.099633014995863e-007 w[637] = 8.109951846981774e-007 w[638] - 4.118359768898598e—007 w[639] = 6.571760029213382e-007

ES 2 631 906 T3

<td>Annex 2</td><td></td><td></td><td>1 w[36]</td><td> 1</td><td>= 5.930981800982896e-001</td>

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

I w[226] I w[227] I w[228] I w(229] I w[230) I w[231] I w(232] I w[233) I w[234] I w[235] I w(236] I w[237) I w(238] I w[239] I w[240] i w[241) I w[242] I w[243] I w[244] I w[.245] I w[246] I w(247] I w(248] I w(249] I w(2S0] I w(251) I w[252) I w(253) I w[254) I w[255] I w(256] I w[257] I w(258] I w[259] I w(260) I w[261] I w[262] I w[263]

ES 2 631 906 T3

ES 2 631 906 T3

I w[378] | = 9.862027119530482e-004 I w[379] I = 1.O47654674829846e-OO3 I w[380] | = 1,099000599887377e-003 I w[381] | = 1.151795860160292e-0Q3 I w[382] | = 1.194743370333155e-003 I w[383] I = 1.250742797799558e-003 I w[384] l = 1.287819050086379e-003 I w[385] [ = 1.263569296641556e-003 I w[386] [ = 1.2261131U394085e-003 I w[337] | = 1.177515087338257e-003 I w[388] I = 1.122503050159859e-003 I w[389] | = 1.089428846944533e-003 I w[390] .1 = 1.054963366189962e-003 I w[391] l = 9.019128558297515e—004 I w[392] [ = 7.847839620863715e-004 I w[393] I = 6.205675927856794e-004 I w[394} | = 3.157663628445906e-004 I w[395] I = 2.556449844935384e-004 I w[396] I = 2.520606580606257e-004 I w[397] | = 2.346980949474655e-004 I w[398] | = 2.060394037017961e-004 I w[399] I = 1.635905995590986e-004 I w[400] I = 1.176237128375623e-004 I w[401] 1 = 6.193369904730005e-005 I w[402] i = 3.568554800150508e-005 I w[403J j = 2.443161189273522e—005 I w[404] | = 1.334090914042349e-005 I w[405] | = 2.853437194757816e-006 I w[406] I = 1.039263591111469e-004 I w(407] | = 5.144969377044875e-005 I w[408] I = 9.7116B1816385056e-005 I w[409] I = 2.472023910553232e-005 | w[410] | = 5.397064424090302e-005 I w[411] I = 6.487880719449901e-005 | w(412] | = 5.192444140699947e—005 | w[413] 1 = 9.204876089551197e-005 | w[414] | = l,815837353167847e-004 | w[415] I = 3.595054179561440e-004 ) w[416] | = 5„901617707607606e-007 ( w[417) | = 1.831121301698088e-004 I w[418] | = 9.755685190624611e-005 I w[419] | = 6.606461762989423e-005 ! w[420] | = 3.799971890923797e—005 I w(421] I = 4.150075391929448e—005 I w[422] | = 5.021905476506264e—005 I w[423] | = 5.861800137434713e-005 I w[424] | = 2.126364641291926e—005 I w[425] | = 1.181077582797280e-00.4 I w(426] I = 9.990757789944374e-005 t w[427] | = 1.035782617124906e-004 t w(428] | = 8.870181845310037e-005 [ w[429] | = 5.533953373249822e-005 I w[430] | = 1.580188994455254e-005 1 w[431] | = 1.277184430250593e-006 í w[432] | = 5.009913312943629e-006 I w[433] | = 1.499170392246774e-005 I w[434] | = 2.241545750231630e—005 I w[435] | = 3.628511258723260e-005 I w[436] | = 2.406516798531014e-005 I w[437] | = 2.515118233957011e-005 I w[438] | = 3.759629789955498e-005 I w[439] | = 5,408154543124121e-005 I w[440] | = 4.493916063285122e-005 I w[441] | = 2.806963579578946e-00S | w[442] ,| = 2.364518513682831e-005 1 w[443J I = 1.260639764582286e-005 I w[444] I = 2.599467772603631e-008 I w[445] I = 1.774108392496017e-005 I w[446) I = 5.889276659458115e-006 | w[447] | = 4.663777919108619e—005 I w[448J I = 2.078886359425321e-004 I w[449] | = 2.131405580107761e-004 ¡ w(4S0] I = 1.784192600231068e-004 I w[451] | = 1.744B41754193053e-004 I w[452] | = 1.728672507238372e-004 I w[453] | = 1.885286127508226e-004

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

I w[606] I w[607] I w[608] | w[609] I w[610] I w[611] | w[612] I w[613J I w[614] I w[615] .1 w[616] I w[617) I w[618] I w[619] I w[620] I w[621] I w[622] I w[623] I w[624] I w[625] I WÍ626] I w[627] I w[628] I w[629] I w[630] I w[631] I w[632] I w[633] I w[634] I w[635] 1 w[636]

I w[637] I w[638] I w[639]

3.575650976036433e-007 2.722295965060517e-005 8.676848186676888e-006 3.428660858940255e-007 4.76779394994489Oe-007 3.330981930777764e-007 2.399696144635756e-007 7,326611439066549e-009 1,349943693297681e~007 5.393555749348494e-008 3.629067065524143e-006 5.690530948134642e-006 1.387566465624550e-008 2.443085172403935e-007 1.723217058490933e-009 7.391973323448250e-008 5.303527922331415e-008 8.883499047404846e-010 3.870536804891648e-009 1.846547564287500e-008 4.244090917065736e-009 4.013524925634108e-009 6.325664562585882e-010 6.025110605409611e-010 1.620171502086309e-006 5.490569954646963e-007 6.355303179925355e-008 5.426597100684762e-009 4.292861814894369e-007 6.834209542421138e-007 7.099633014995863e-007 8.109951846981774e—007 4.118359768898598e-007 6.571760029213382e-007

ES 2 631 906 T3

<td>Annex 3</td><td></td><td></td><td></td><td></td><td> 0.592</td><td> <</td><td>w[36]</td><td> <</td><td> 0.594</td>

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

-0.001

-0,001

-0.001

-0.001 < w[606] < W[6O7).

< w[608] < w[609] < w[610] < w[611] < w[612] < w[613] < w[614] < w[615] < w[616] < w[617] < w[618] < w[619] < w[620] < w[621] < w[622] < w[623] < w[624] < w[625] < w(626] < w(627] < w[628] < w[629] < w[630] < w[631] < w[632] < w[633] < w[634] < w[635] < w[636] < w[637] < w[638] < w[639] < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 <. 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001 < 0.001

ES 2 631 906 T3

<td>Annex 4</td><td></td><td></td><td></td><td></td><td></td><td> 0.592</td><td> <</td><td> 1</td><td>w[36]</td><td> 1</td><td> <</td><td> 0.594</td>

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

Contents49

138 members in 26 offices

This recordMember, by publication datePriority claim16 more offices are in the list below

Priority claims3

Priority claims
Document	Office	Kind	Date
862954P	United States of America	–	2006-10-25
86295406	United States of America	P	2006-10-25
2007009199	European Patent Office (EPO)	W	2007-10-23

Members138

Family members, oldest first
Document	Office	Kind
AU2007308415A1	Australia	A1	2008-05-02
AU2007308416A1	Australia	A1	2008-05-02
CA2645618A1	Canada	A1	2008-05-02
CA2667505A1	Canada	A1	2008-05-02
WO2008049589A1	World Intellectual Property Organization (WIPO)	A1	2008-05-02
WO2008049590A1	World Intellectual Property Organization (WIPO)	A1	2008-05-02
TW200836166A	Taiwan Province of China	A	2008-09-01
TW200837719A	Taiwan Province of China	A	2008-09-16
MX2008011898A	Mexico	A	2008-11-06
KR20080102222A	Republic of Korea	A	2008-11-24
EP1994530A1	European Patent Office (EPO)	A1	2008-11-26
AR063394A1	Argentina	A1	2009-01-28
AR063400A1	Argentina	A1	2009-01-28
HK1119824A	Hong Kong, China	A	2009-03-13
HK1119824A1	Hong Kong, China	A1	2009-03-13
CN101405791A	China	A	2009-04-08
MX2009004477A	Mexico	A	2009-05-13
KR20090058029A	Republic of Korea	A	2009-06-08
EP1994530B1	European Patent Office (EPO)	B1	2009-07-01
EP2076901A1	European Patent Office (EPO)	A1	2009-07-08
AT435480T	Austria	T	2009-07-15
ATE435480T1	Austria	T1	2009-07-15
NO20084012L	Norway	L	2009-07-22
NO20091951L	Norway	L	2009-07-22
NO20170452A1	Norway	A1	2009-07-22
DE602007001460D1	Germany	D1	2009-08-13
JP2009530675A	Japan	A	2009-08-27
DK1994530T3	Denmark	T3	2009-08-31
PT1994530E	Portugal	E	2009-10-09
EP2109098A2	European Patent Office (EPO)	A2	2009-10-14
ES2328187T3	Spain	T3	2009-11-10
CN101606194A	China	A	2009-12-16
IL197976A0	Israel	A0	2009-12-24
IL197976D0	Israel	D0	2009-12-24
US2009319283A1	United States of America	A1	2009-12-24
ZA200810308B	South Africa	B	2009-12-30
PL1994530T3	Poland	T3	2009-12-31
US2010023322A1	United States of America	A1	2010-01-28
JP2010507820A	Japan	A	2010-03-11
RU2008137468A	Russian Federation	A	2010-03-27
ZA200902199B	South Africa	B	2010-04-28
KR100957711B1	Republic of Korea	B1	2010-05-12
AU2007308416B2	Australia	B2	2010-07-08
AU2007308415B2	Australia	B2	2010-08-26
RU2009119456A	Russian Federation	A	2010-11-27
MY142520A	Malaysia	A	2010-12-15
RU2411645C2	Russian Federation	C2	2011-02-10
RU2420815C2	Russian Federation	C2	2011-06-10
BRPI0709310A2	Brazil	A2	2011-07-05
KR101056253B1	Republic of Korea	B1	2011-08-11
IL193786A	Israel	A	2011-11-30
TWI355649B	Taiwan Province of China	B	2012-01-01
CN101405791B	China	B	2012-01-11
TWI357065B	Taiwan Province of China	B	2012-01-21
JP4936569B2	Japan	B2	2012-05-23
CN101606194B	China	B	2012-06-27
JP5083779B2	Japan	B2	2012-11-28
CA2645618C	Canada	C	2013-01-22
US8438015B2	United States of America	B2	2013-05-07
US8452605B2	United States of America	B2	2013-05-28
MY148715A	Malaysia	A	2013-05-31
US2013238343A1	United States of America	A1	2013-09-12
IL197976A	Israel	A	2014-03-31
US8775193B2	United States of America	B2	2014-07-08
CA2667505C	Canada	C	2015-07-07
EP2076901B1	European Patent Office (EPO)	B1	2017-05-10
BRPI0716315A2	Brazil	A2	2017-05-30
EP2109098A3	European Patent Office (EPO)	A3	2017-06-21
EP2076901B8	European Patent Office (EPO)	B8	2017-08-16
PT2076901T	Portugal	T	2017-08-23
ES2631906T3This record	Spain	T3	2017-09-06
PL2076901T3	Poland	T3	2017-09-29
NO341567B1	Norway	B1	2017-12-04
NO341610B1	Norway	B1	2017-12-11
EP3288027A1	European Patent Office (EPO)	A1	2018-02-28
NO342691B1	Norway	B1	2018-07-09
HK1251073A	Hong Kong, China	A	2019-01-18
HK1251073A1	Hong Kong, China	A1	2019-01-18
BRPI0709310B1	Brazil	B1	2019-11-05
EP2109098B1	European Patent Office (EPO)	B1	2020-09-09
PT2109098T	Portugal	T	2020-12-18
PL2109098T3	Poland	T3	2021-03-08
EP3288027B1	European Patent Office (EPO)	B1	2021-04-07
ES2834024T3	Spain	T3	2021-06-16
PT3288027T	Portugal	T	2021-07-07
EP3848928A1	European Patent Office (EPO)	A1	2021-07-14
PL3288027T3	Poland	T3	2021-10-18
ES2873254T3	Spain	T3	2021-11-03
EP3848928B1	European Patent Office (EPO)	B1	2023-03-15
FI3848928T3	Finland	T3	2023-06-02
PT3848928T	Portugal	T	2023-06-07
DK3848928T3	Denmark	T3	2023-06-19
EP4207189A1	European Patent Office (EPO)	A1	2023-07-05
PL3848928T3	Poland	T3	2023-07-17
ES2947516T3	Spain	T3	2023-08-10
EP4207189B1	European Patent Office (EPO)	B1	2023-11-22
EP4207189C0	European Patent Office (EPO)	C0	2023-11-22
EP4300824A2	European Patent Office (EPO)	A2	2024-01-03
EP4300825A2	European Patent Office (EPO)	A2	2024-01-03
EP4325723A2	European Patent Office (EPO)	A2	2024-02-21

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Abstract

Term

ES 2 631 906 T3

DESCRIPTION

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

Y

ES 2 631 906 T3 (8)

2N-I

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3 (36)

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

ES 2 631 906 T3

Priority claims3

Members138

Numbers

Titles2

Classification