Electric motor driver and drive control system thereof

ES 2 403 239 T3

DESCRIPTION

Electric motor drive and drive control system

BACKGROUND OF THE INVENTION

FIELD OF THE INVENTION

The present invention relates to an electric motor actuator and a drive control system for controlling a drive of an electric motor, such as a brushless DC motor, regardless of the number of rotations.

DESCRIPTION OF THE PREVIOUS TECHNIQUE

In prior art US5457375 a sensorless switching controller for a polyphase dynamoelectric machine (electric motor) is described, in which a conductive current is supplied to the electric motor. Depending on the driving current, a PWM inverter switches the motor supplying, in a predetermined way, a corresponding current to the motor windings. The operation of the motor is monitored and a commutation angle control is carried out based on an analysis of the waveform of the conductive current having characteristics that are a function of a commutation angle of the motor. Specifically, a control variable that is a function of the switching angle is processed and the control includes additional information on the waveform of the conducting current (fluctuations or transients). The variable is combined with additional sampled data according to a preset formula, and the impulse response of the motor is also taken into account.

In prior art US5920175A, an instantaneous position indicating an apparatus for a sensorless switched reluctance machine system is described, in which an instantaneous position generating circuit develops a signal to control the switching of the switching machine. switched reluctance. The instantaneous position generation circuit includes a digital control counter that increments as a function of the estimated rotor speed of the switched reluctance machine. An output of the digitally controlled counter is fed back to modify the counter increment to ensure that the counter output properly tracks the estimated rotor position. In combination with an accumulator to receive data words, the meter is able to adequately control the switching circuitry even under high speed conditions.

Lately, when constructing a motor driver to drive an electric motor, such as a compressor for use in an air conditioning system, it is more and more necessary to reduce energy consumption in view of environmental protection on a global scale. As an energy-saving technology in a motor driver, an inverter is generally used to drive a high-performance electric motor, such as a brushless DC motor, regardless of the number of rotations.

Also, as a motor drive technology, a sine wave drive technology that uses a sine waveform current is mentioned because it has high performance and low noise, rather than a square wave drive technology that uses a current. rectangular waveform.

When using an electric motor, such as a compressor, in an air conditioning system, since it is difficult to mount a sensor to detect a rotational position of a rotor of the electric motor, use is made of a type of drive technology sine wave, without position sensor, which drives the electric motor estimating a rotor rotation position with a procedure without using a rotation position sensor. Furthermore, as a method of estimating a rotational position of a rotor, a known method of estimating an induced voltage of an electric motor is used. (See, for example, Patent Document 1: Japanese Patent Publication, Laid-Open, No. 2000-350489).

Fig. 9 illustrates a system construction implementing a type of sine wave drive technology, without a position sensor. In Fig. 9, reference numeral 1 indicates a DC power supply, 2 indicates an inverter, 3 indicates a brushless motor, 4 indicates a stator, 5 indicates a rotor, 6 indicates a controller and reference numerals 7v and 7w indicate current sensors. In this construction, the brushless motor 3 comprises the stator 4 and the rotor 5, in which the stator 4 has three phase windings 4u, 4v and 4w which are star connected at a neutral point, as its centering point, and the rotor 5 has a magnet mounted thereon. A phase terminal U 8u, a phase terminal V 8v and a phase terminal W 8w are connected to an unconnected end of the phase winding U 4u, to an unconnected end of the phase V winding 4v and to an unconnected end of the phase winding W 4w, respectively.

Inverter 2 has three series circuits for one phase U, one phase V, and one phase W, which are connected in parallel. Each of the series circuits includes a pair of switching elements connected in series at the

ES 2 403 239 T3 upper and lower flows of a stream. A DC voltage output from DC power supply 1 is applied to all three circuits in series. The series circuit for phase U has a switch element 12u on the up side and a switch element 13u on the down side. The series circuit for phase V has a 12v switching element on the rising side and a 13v switching element on the falling side. The series circuit for phase W has a 12w switching element on the rising side and a 13w switching element on the falling side.

Furthermore, free-flowing diodes 14u, 14v, 14w, 15u, 15v, and 15w are connected in parallel to the switching elements 12u, 12v, 12w, 13u, 13v, and 13w, respectively. The terminals 8u, 8v and 8w of the brushless motor 3 are connected, respectively, to an interconnection point between the switching elements 12u and 13u, to an interconnection point between the switching elements 12v and 13v and to an interconnection point between the 12w and 13w switching elements of the inverter 2.

The currents flowing through the 4v and 4w phase windings of the brushless motor 3 are detected by the current sensors 7v and 7w and the detected current values ​​are supplied to the controller 6. The controller 6 estimates an induced voltage based on the supplied current values ​​and sends a control signal to control the drive of the inverter 2. Therefore, the control by driving the brushless motor 3 is carried out in the above circuit constitution.

As described above, in the conventional type of position sensorless sine wave drive technology implemented estimating induced voltage, at least two current detectors, such as current sensors, must be used to detect currents. phase of an electric motor to be driven, which increases the cost of setting up a motor actuator.

SUMMARY OF THE INVENTION

The present invention is made to solve the above problem and an essential objective of the present invention is to provide an electric motor driver and drive control system that can detect a phase current with great precision in an inexpensive constitution and perfectly drive a electric motor in low-speed rotation zone to high-speed rotation zone having high-speed stability in low-speed zone.

In order to achieve the above-mentioned objective, a first aspect of the invention provides an electric motor driver for driving a three-phase electric motor by controlling a drive as a function of each phase current flowing through each phase winding of the three-phase electric motor, as defined in claim 1.

Other preferred embodiments of the present invention are defined in the dependent claims.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other objects and features of the present invention will be readily understood from the following detailed description when read in conjunction with the preferred embodiments thereof in conjunction with the accompanying drawings, in which like parts are indicated by reference numerals similar and in which:

Fig. 1 is a block diagram showing a system construction of an electric motor actuator according to an embodiment 1 of the present invention;

Fig. 2 illustrates an example of time-related variations of phase current conditions of an electric motor;

Fig. 3 illustrates an example of variations of a PWM signal;

Fig. 4 illustrates current conditions flowing in the electric motor and in an inverter when the PWM signal is varied as shown in Fig. 3;

Fig. 5 illustrates an example of variations of the PWM signal;

Fig. 6 illustrates current conditions flowing in the electric motor and in an inverter when the PWM signal is varied as shown in Fig. 5;

Fig. 7 illustrates an example of PWM signal variations;

Fig. 8 illustrates adjusted values ​​of a speed proportional gain KP by means of calculating PI of

ES 2 403 239 T3 speed and Fig. 9 is a block diagram showing a conventional example.

DETAILED DESCRIPTION OF THE PREFERRED FORM OF EMBODIMENT

A preferred embodiment of the present invention is described below with reference to the accompanying drawings.

(Form of embodiment 1)

Fig. 1 shows a system construction implementing an embodiment 1 of the present invention which is a combination of an electric motor actuator and an electric motor. In Fig. 1, reference number 1 indicates a DC power supply, 2 indicates an inverter, 3 indicates a brushless motor, 4 indicates a stator, 5 indicates a rotor, and reference number 6 indicates a controller. Brushless motor 3 comprises stator 4 and rotor 5, in which stator 4 has three phase windings 4u, 4v and 4w that are star connected at a neutral point, as its centering point, and rotor 5 It includes a magnet mounted thereon and the magnet has magnetic poles N and S. A phase terminal U 8u is connected to an unconnected end of the phase winding U 4u, that is, to the side opposite the neutral point. Similarly, a phase V 8v terminal and a phase W 8w terminal are connected to the unconnected end of the phase V winding 4v and to the unconnected end of the phase winding W 4w, respectively.

Inverter 2 has three series circuits connected in parallel for phase U, phase V and phase W, each of which includes a pair of switching elements connected in series in the upper and lower flow of a current. A DC voltage output from the DC power supply 1 is applied to said series circuits of the inverter 2 through a voltage detector 16 that detects a voltage value applied to the inverter. Series circuits for phase U include a switch element 12u on the up side and a switch element 13u on the down side.

Similarly, the series circuit for phase V includes a 12v switching element on the rising side and a 13v switching element on the falling side. In addition, the series circuit for phase W includes a 12w switching element on the rising side and a 13w switching element on the falling side. Furthermore, free-flowing diodes 14u, 14v, 14w, 15u, 15v, and 15w are connected in parallel to the switching elements 12u, 12v, 12w, 13u, 13v, and 13w, respectively.

The terminals 8u, 8v and 8w of the brushless motor 3 are connected to an interconnection point between the switching elements 12u and 13u, to an interconnection point between the switching elements 12v and 13v and to an interconnection point between the switching elements. 12w and 13w switching of inverter 2, respectively.

The DC voltage applied from the DC power supply 1 to the inverter 2 is converted into a three-phase AC voltage by means of the series circuits constituted by the switching elements of the inverter 2 to thereby drive the motor without brushes 3. In order to implement a target speed applied from the outside, a correction value is calculated based on a difference between the target speed and a current speed and in order for the inverter to send a calculated voltage to be applied to the brushless motor. , a PWM signal is generated in the controller 6 to control the actuation of the switching elements of the inverter 2.

The controller 6 includes a PWM signal generating part 9 for generating a PWM signal (PWM) for driving-controlling the switching elements of the inverter 2 and a duty cycle correction part 19 for correcting the PWM signal that it is converted to a drive signal (DRV) by means of a base driver 10 to electrically drive the switching elements. Next, the drive signal is applied from the base drive 10 to the inverter 2 to drive the switching elements 12u, 12v, 12w, 13u, 13v, and 13w.

Furthermore, the controller includes a current detector 11, which is provided in an inverter bus bar, to detect an inverter bus bar current to determine a phase current (iu, iv, iw represented by "is") of the brushless motor 3 and an induced voltage estimating part 17 to estimate an induced voltage (eu, ev, ew represented by "es") of the brushless motor 3.

Furthermore, the controller 6 includes a rotor speed / position estimating part 18 for estimating a one-pole angle rotation position 0m and a rotational speed wm of the rotor of the brushless motor 3, an angle PI calculating part 20 for calculating a correction value (CRA) by performing a proportional and integral calculation based on information of the rotor angle polar position and a velocity PI calculating part 21 for calculating a value correction (CRS) by carrying out a proportional and integral calculation based on the estimated rotor speed.

ES 2 403 239 T3

Next, an operation of the controller 6 is briefly described. An induced voltage es = (eu, ev, ew) of the brushless motor 3 is estimated by means of the induced voltage estimation part 17 as a function of the information of the phase current "is" = (iu, iv, iw ) of the brushless motor 3 detected by the current detector 11 on the inverter busbar, an output voltage Vs * calculated by means of the PWM signal generation part 9 and a voltage applied to the inverter detected by the detector voltage 16.

Furthermore, a pole position at angle 0m and a rotational speed wm of the rotor of the brushless motor 3 are estimated by means of the rotor speed / position estimation part 18. The PWM signal to control the drive of the brushless motor 3 is generated by means of the PWM signal generation part 9 as a function of correction values ​​(CRA, CRS). The first correction value CRA is obtained by a proportional and integral calculation carried out by means of the calculation part PI of angle 20 as a function of the information 0m of the angle polar position of the rotor. The second correction value CRS is obtained by the proportional and integral calculation carried out by means of the speed calculating part PI 21 as a function of the estimated rotational speed wm in relation to a target speed ω *. Therefore, the speed of the rotor is controlled to be the target speed with information such as deviation of the estimated speed wm of the rotor 5 from the target speed ω * applied from the outside.

Next, an operation of the induced voltage estimation part 17 is described. The current detector 11 determines the phase currents iu, iv and iw (represented by "is"), which flow through the phase windings, detecting the inverter bus bar current and the detected phase current values ​​are supplied to the induced voltage estimating part 17. Furthermore, the output voltage Vs * calculated by means of the PWM signal generation part 9 and a voltage applied to the inverter, which was detected by means of the voltage detector 16, are supplied to the induced voltage estimation part 17 Therefore, the phase voltages (vu, vv and vw) applied to the phase windings are calculated by means of the induced voltage estimation part 17 based on the information provided above.

In theory, the induced voltage values ​​eu, ev and ew (represented by es) induced in the phase windings are calculated from the above values ​​of the phase currents (iu, iv, iw) and the phase voltages (vu, vv, vw) with the following equations (1), (2) and (3). In the following equations, R is resistance and L is inductance and d (iu) / dt, d (iv) / dt and d (iw) / dt are time differentiations of phase currents iu, iv, and iw, respectively.

eu = vu - R · iu - L · d (iu) / dt ··· (1) ev = vv - R · iv - L · d (iv) / dt ··· (2) ew = vw - R · iw - L · d (iw) / dt ··· (3)

When equations (1), (2) and (3) are developed in detail, the following equations (4), (5) and (6) are obtained.

eu = vu

- R iu

- (la + La) d (iu) / dt

- Las cos (20m) d (iu) / dt

- Las · iu · d [cos (20m)] / dt + 0.5 · La · d (iv) / dt

- Las cos (20m - 120 °) d (iv) / dt

- Las · iv · d [cos (20m - 120 °)] / dt + 0.5 · La · d (iw) / dt

- Las cos (20m - 120 °) d (iw) / dt

- Las · iw · d [cos (20m + 120 °)] / dt ··· (4) ev = vv

- R iv

- (la + la) d (iv) / dt

- Las cos (20m + 120 °) d (iv) / dt

- Las · iv · d [cos (20m + 120 °)] / dt + 0.5 · La · d (iw) / dt

- Las cos (20m) d (iw) / dt

- Las · iw · d [cos (20m)] / dt + 0.5 · La · d (iu) / dt

- Las cos (20m - 120 °) d (iu) / dt

- Las · iu · d [cos (20m - 120 °)] / dt ··· (5) ew = vw

ES 2 403 239 T3

- R iw

- (la + La) d (iw) / dt

- Las cos (20m - 120 °) d (iw) / dt

- Las · iw · d [cos (20m - 120 °)] / dt + 0.5 · La · d (iu) / dt

- The cos (2Om + 120 °) d (iu) / dt

- Las · iu · d [cos (20m + 120 °)] / dt + 0.5 · La · d (iv) / dt

- Las cos (20m) d (iv) / dt

- Las · iv · d [cos (20m)] / dt ··· (6)

In this case, d / dt indicates a time differentiation and an electrical angular velocity, converted from an estimated velocity wm is used as dO / dt that appears in the differential calculus referring to a trigonometric function. Furthermore, d (iu) / dt, d (iv) / dt and d (iw) / dt are obtained by means of a first order Euler approximation. Furthermore, the phase current value w iw, the phase current value u iu and the phase current value v iv have a relationship which is represented in an equation (14) which will be mentioned later. In this case, R indicates a resistance for one phase of the winding, La indicates a leakage inductance for one phase of the winding, La indicates a mean value of an effective inductance for one phase of the winding and Las indicates an amplitude of the effective inductance for a phase of the winding.

In the operation of the induced voltage estimation part 17 the following equations (7), (8) and (9) are used which simplify equations (4), (5) and (6). According to simplification, the phase current values ​​iu, iv and iw are assumed to be sine waves and the phase currents iu, iv and iw are obtained from an amplitude of orders of current ia and from a phase of orders of current βΤ and are simplified.

eu = vu + R ia sin (Om + βϊ) + 1,5 (la + La) cos (Om + βϊ)

- 1,5 · Las · cos (Om - βϊ) ··· (7) ev = vv + Ria · sin (Om + βϊ - 120 °) + 1,5 · (la + La) · cos (Om + βϊ - 120 °)

- 1,5 · Las · cos (Om - βϊ - 120 °) ··· (8) ew = vw + R ia · sin (Om + βϊ - 240 °) + 1,5 · (la + La) · cos (Om + βϊ - 240 °)

- 1,5 · Las · cos (Om - βϊ - 240 °) ··· (9)

Next, an operation of the position / speed estimation part of the rotor 18 is described. The angle position Om and the rotational speed wm of the rotor 5 are estimated using the estimated induced voltage values ​​es = (eu, ev and ew) estimated by means of the induced voltage estimation part 17. In particular, an angle Om estimated by means of the position / speed estimation part of the rotor 18 is corrected using an error of the induced voltage to approximate a real value, whereby the estimated speed wm is obtained.

First, the induced voltage reference values ​​(eum, evm and ewm) of the phases are obtained with the following equations.

eum = em sin (Om + βϊ) evm = em sin (Om + βϊ - 120 °) ewm = em sin (Om + βϊ - 240 °) ··· (10)

In this case, an induced voltage amplitude value em is obtained corresponding to the amplitude values ​​of the estimated induced voltages eu, ev and ew. Next, a deviation ε is calculated between the obtained induced stress reference values ​​(esm) and the estimated induced stress values ​​(es).

As shown in the following equation (11), the deviation ε is obtained by subtracting the reference value of induced voltage esm from the estimated value of induced voltage es.

ε ~ es - esm ··· (11)

ES 2 403 239 T3 (in this case, s represents phases u, v and w)

The estimated angle 0m is the actual value when the deviation is zero. Therefore, the estimated angle 0m is subject to the proportional and integral calculation carried out by means of the PI calculation part of angle 20, such that the deviation ε is zero and the resulting correction value (CRA) is transmitted from the angle PI calculation part 20 to the PWM signal generation part 9. Furthermore, the estimated speed wm is obtained by calculating a variation value of the estimated angle 0m in the position / speed estimation part of the rotor 18 and the estimated speed wm is transmitted to the speed calculation part PI 21.

In the PI speed calculation part 21, in order to implement a target speed ω *, using a difference Δω between the target speed ω * and the estimated speed ωιτι, the correction value (CRS) is obtained with Equation (12).

ΚΡΔω + ΚΙΔω ··· (12) (KP: proportional gain, KI: integral gain)

The obtained correction value (CRS) is transmitted from the speed PI calculation part 21 to the PWM signal generation part 9. In the PWM signal generation part 9, a voltage V * is calculated that goes to be sent based on the correction values ​​(CRA, CRS) of the PI calculation part of angle 20 and of the PI calculation part of speed 21. Next, a voltage Vs * is obtained to be applied to each phase (in this case, representing the phases u, v and w) from the voltage value V * according to the following equation (13).

Vu * = V * sin (0m + βΤ)

Vv * = V * sin (0m + βΤ - 120 °)

Vw * = V * · sin (0m + βΤ - 240 °) ··· (13)

Furthermore, for inverter 2 to send the voltage obtained Vs * (s: phases u, v and w) to be applied to each of the phases, the PWM signal is corrected by means of the duty cycle correction part 19 and is sent as the drive signal (DRV), through the base driver 10, to be applied to each of the switching elements 12u, 12v, 12w, 13u, 13v and 13w. Specifically, the duty cycle correction part 19 corrects a duty cycle of the PWM signal generated by the PWM signal generating part during the detection of the inverter busbar current by means of the detector. current 11. Next, each of the switching elements 12u, 12v, 12w, 13u, 13v and 13w are electrically actuated by means of the drive signal (DRV) to generate a sinusoidal alternating current to be applied to each of the phases thereby driving the brushless motor.

According to the present embodiment, the estimated angle 0m is provided using the deviation ε between the estimated induced voltage value es and the induced voltage reference value esm in the rotor position / speed estimation part 18 and hence Thus, the sinusoidal drive of the brushless motor 3 is implemented by the flow of the sinusoidal phase currents.

In this case, referring to Figs. 2 to 6, the phase current of the brushless motor 3 appearing in the current flowing in the inverter bus bar is described. Fig. 2 illustrates a condition of the phase current flowing in each phase winding of the brushless motor 3 each electrical angle section. In the section on electrical angles 0 to 60 °, it is shown that the phase currents flow from the unconnected ends to the neutral point of the phase winding U 4u and from the phase winding W 4w and from the neutral point to the unconnected end of the phase V 4v winding.

In the section of electrical angles 60 to 120 °, it is shown that currents flow from the unconnected end to the neutral point of the phase winding U 4u and from the neutral point to the unconnected ends of the phase winding V 4v and the winding of phase W 4w. Therefore, it is shown that the conditions of the phase currents flowing through the phase windings are varied every 60 ° electrical angle.

For example, when the PWM signals generated by the PWM signal generating part 9 at the electrical angle 30 ° of Fig. 2 vary as shown in Fig. 3 (U, V, W, X, Y and Z are signals to drive the switching elements 12u, 12v, 12w, 13u, 13v and 13w, respectively, which are described by means of an active Hi), no current flow appears (zero) at one time (1), a current appears flowing through phase winding W 4w at one time (2) and a current appears flowing through phase winding V 4v at one time (3) at the inverter bus bar, as shown in Fig. 4.

Furthermore, when the PWM signals generated by the PWM signal generating part 9 are varied by the electrical angle 30 ° in Fig. 2, as shown in Fig. 5, no current flow appears (zero ) at time (1), a current appears flowing through phase winding U 4u at time (2) and a current appears flowing through phase winding V 4v at time (3) in the bus inverter collector, as

ES 2 403 239 T3 shown in Fig. 6.

Therefore, it can be seen that the phase currents appear in the inverter busbar according to the conditions of the switching elements 12u, 12v, 12w, 13u, 13v and 13w of inverter 2.

When the currents corresponding to the two phases are provided at the adjacent times, as described above, it is obvious that the currents of the three respective phases are obtained with a relation of the following equation (14):

iu + iv + iw = 0 ··· (14)

However, when the PWM signals generated by the PWM signal generating part 9 are varied by the electrical angle 30 ° in Fig. 2, as shown in Fig. 7, no current flow appears ( zero) at time (1) and only a current flows through phase V winding 4v at time (3) appears on the inverter bus bar. When the PWM signals of such variations are repeated, the currents of the respective three phases are not provided and the induced voltage estimation part 17 cannot estimate the induced voltage and the brushless motor 3 cannot be driven.

In order to avoid the above problem, in the duty cycle correction part 19, when it is necessary to detect a phase current flowing in each of the phase windings of the brushless motor 3, the generated PWM signals are checked. by means of the PWM signal generation part 9. If the signal variation is as shown in Fig. 7, the generated PWM signals are corrected, for example, to those shown in Fig. 3.

Also, when the inverter bus bar current is shown as an example in Fig. 3, a current oscillation is generated at the time of switching. Furthermore, a noise or the like is superimposed on the current detector 11. In order to eliminate such influences, the PWM signal generating part 9 has a low-pass filter or the like (not shown).

While a speed proportional gain (KP) of the speed PI calculation part 21 can be varied linearly according to a speed in a low speed zone of the electric motor, the stability and responsiveness deteriorate somewhat when a pulse width is reduced (at the moment of a low speed) due to the previous correction and the low pass filter. Therefore, in a preferred embodiment, the speed calculation part PI 21 can be adapted to adjust the values ​​of the speed proportional gain KP in various stages according to a rotational speed ω, as shown in Fig. 8. Therefore, a suitable speed proportional gain KP can be adjusted according to a load to thereby prevent the oscillation or overcurrent from being generated.

When the speed proportional gain KP is increased above the line (1) of Fig. 8, the control is too sensitive and the overcurrent flows in each of the switching elements of the inverter 2. When the speed proportional gain KP is decreased below line (2), responsiveness is reduced due to lack of gain, and pendulous may be generated. However, such problems can be avoided with the multi-stage adjustment detailed above.

Furthermore, in a preferred embodiment, the rate PI calculation part 21 may include a memory 21a, such as EEPROM or the like, in which values ​​of the rate proportional gain KP can be stored, so that they can be read and record on the outside of the controller 6. Therefore, the values ​​according to a difference in loads, produced by a difference in capacity or the like of a compressor or the like that uses a three-phase electric motor, are stored in memory 21a so that they correspond to each load, with which it is possible to share an electronic control unit without changing a ROM of a microcomputer.

In addition, duty cycle information of the PWM signals sent from duty cycle correction part 19 is applied to current detector 11. While current detector 11 detects the current from the inverter bus, the current detector 11 detects the current from the inverter busbar. Current 11 determines which phase current of the three phases of brushless motor 3 appears on the inverter bus bar.

Therefore, the current detector 11 determines the respective currents flowing through the three phases as a function of the inverter bus current in relation to the duty cycle information. The resulting current values ​​of the three phases are used in the induced voltage estimate calculation of the induced voltage estimation part 17 below.

As described above, by providing the current detector 11 which detects the inverter busbar current and the duty cycle correction part 19, the phase current flowing in each of the phase windings can be detected. of the brushless motor 3. In addition, the values ​​of the speed proportional gain KP are adjusted in several stages according to a rotational speed by means of the PI calculation part of

ES 2 403 239 T3 speed 21, stability in a low speed area can be improved.

As described above, according to the electric motor driver of the present invention, the effect can be obtained that a sine wave drive in a range from low to high speed can be stably implemented in a construction of economical system, without providing two or more correine detectors in a wiring harness between the inverter and the electric motor.

In addition, since the speed proportional gain for the PI speed control is stored in a memory that can be read and written, various loads can be matched (differences from the three-phase electric motor 10 or a product to be used) varying only memory data, so that a microcomputer and electrical controller can be shared.