Apparatus and method accounting for effects of discontinuities at the output of automatic gain control in a multi carrier system

Apparatus and procedure to take into account the effects of discontinuities on the output of automatic gain control in a multiple carrier system

Reference to patent related applications

The present patent application is related to the following US patent applications pending with this:

"TIME CORRECTIONS IN A MULTIPLE CARRIER SYSTEM AND PROPAGATION TO A CHANNEL ESTIMATE TIME FILTER" by Bojan Vrcelj et al., Which has the US patent application number 11 / 373,764, filed on March 9, 2006, assigned to the assignee of this application.

"TIME ADJUSTMENTS FOR CHANNEL ESTIMATION IN A MULTIPLE CARRIER SYSTEM" by Matthias Brehler et al., Which have a US patent application number 11/777, 251, filed July 12, 2007, assigned to the assignee of the present request.

Background

Countryside

This disclosure refers to devices and procedures to take into account automatic gain control (AGC) in a wireless system of multiple carriers, and more particularly, to adjust the combination coefficients to take into account the AGC, which are used to combine interlaced pilot tones in an interlaced filter to determine the channel estimate.

Background

Orthogonal Frequency Division Multiplexing (OFDM) is a digital modulation procedure whereby a signal is divided into several narrowband channels into different orthogonal carrier frequencies with respect to each other. These channels are sometimes called subbands or subcarriers. In some aspects, OFDM is similar to Conventional Frequency Division Multiplexing (FDM), except for the way in which the signals are modulated and demodulated. One of the advantages of OFDM technology is that it reduces the amount of interference or crosstalk between channels and symbols in sineral transmissions . However, time-varying and frequency-selective fading channels present problems in many OFDM systems.

In order to take into account the fading channels that are variable in time and selective in frequency, channel estimation is used. In coherent detection systems, the reference values ​​or "pilot symbols" (also known simply as "pilots") incorporated into the data of each OFDM symbol can be used for channel estimation. Time and frequency tracking can be achieved using the pilots in the channel estimate. For example, if each OFDM symbol consists of a number of subcarriers and a number P of pilots, a number N of subcarriers can be used for data transmission and the number P of them can be assigned to the pilot tones. The number P of pilots sometimes extends evenly over the N subcarriers, so that every two tones of pilots are separated by N / P1 data subcarriers (or, in other words, each pilot is produced in each N / P is the carrier) . Such uniform subsets of subcarriers in an OFDM symbol and over a number of symbols that occur over time are called interlaced.

In an application area, OFDM is used for digital broadcasting services, such as the Forward Link Only (FLO) standards, Digital Video Broadcasting (DVBT / H (ground / manual)), and Integrated Services Digital Broadcasting (ISDB T). In such wireless communication systems, the characteristics of the channel, in terms of the number of derivations of the channel (ie, the number of samples or "length" of a Finite Pulse Response (FIR) filter that is used to represent the channel from a received signal) with significant energy, the path gains and the path delays are expected to vary quite significantly over a period of time. In an OFDM system, a receiver responds to changes in the channel profile by appropriately selecting the OFDM symbol limit (i.e., window time correction) to maximize the energy captured in a Fast Fourier transform window ( FFT)

In OFDM receivers it is common for a channel estimation block in a receiver to be stored in a buffer and then pilot observations of multiple OFDM symbols are processed, resulting in a channel estimate that has a better average of noise and resolves longer channel delay enlargements. This is achieved by combining the observations of the length P channel of the OFDM symbols consecutively timed in a longer channel estimate in a unit called a time filtering unit. Longer channel estimates, in general, can lead to more robust time synchronization algorithms. However, automatic gain control (AGC) can limit the performance of the interlacing combination. In particular, the AGC introduces discontinuities in a channel, negatively affecting the combination of interlacing with increasing severity the more interwoven are combined, such as in the DVB and ISDB systems in particular. As a consequence, the adverse effects of AGC in the interlaced combination degrades the channel estimate.

Attention is drawn to an article by LINDE LP, entitled "An AGC strategy for digital adaptation modems in frequency hopping applications" COMMUNICATIONS AND SIGNAL PROCESSING, 1989. COMSIG 1989.. PROCEDURES OF THE CONFERENCE OF SOUTH AFRICA IN STELLENBOSCH, SOUTH AFRICA, JUNE 23, 1989, NEW YORK, NY, USA, IEEE, US, January 1, 1989 (19890101), pages 19 24, XP010042974 ISBN: 978 - 0 87942 - 713 9

Summary

In accordance with the present invention, there is provided a method for adjusting the effect of AGC, as set forth in claim 1, and an apparatus for use in a wireless transceiver, as set forth in claim 9, Embodiments of the invention are claimed in the dependent claims.

Brief description of the drawings

Figure 1 illustrates a block diagram of an exemplary transceiver in accordance with the present disclosure.

Figure 2 is a diagram of an exemplary pilot tone scaling scheme, used in particular OFDM standards.

Figure 3 is a diagram of a visualization of the pilot tone combination of the exemplary pilot tone step scheme of Figure 2.

Figure 4 illustrates a graph of a channel gain over time in a system without automatic gain control.

Figure 5 illustrates a graph of a channel gain over time in a system that uses automatic gain control.

Figure 6 a procedure for determining the combination coefficients adjusted to take into account the automatic gain control time in a wireless device.

Figure 7 illustrates an apparatus for determining the adjusted combination coefficients that estimate the automatic gain control time in a wireless device.

Figure 8 illustrates an exemplary graph of a simulation that shows improved performance characteristics of a system that takes into account automatic gain control with respect to a system that does not take into account automatic gain control.

Detailed description

The present disclosure explains an apparatus and procedures for adjusting the effects of automatic gain control when the interlaced pilots are combined in an interlaced filter of a communication system, such as an OFDM system. The procedures and devices disclosed achieve the reversal of the effects of discontinuities introduced by automatic gain control (AGC) by combining interlaced pilots. Consequently, the channel estimate, and therefore the transceiver performance, are improved.

Figure 1 illustrates a block diagram of an exemplary OFDM transceiver option of a transceiver according to the present disclosure. The system of Figure 1, in particular, can employ the disclosed techniques to make time adjustments using pilot tones, which are used for channel estimation. The system 100, which can be a transceiver or one or more processors, hardware, firmware, or a co-molding thereof, receives a transmitted RF signal, as shown. A front process block 102 receives the RF signal and performs various process functions, including analog to digital conversion, reductive conversion, and the AGC (Automatic Gain Control) unit 103. The AGC unit 103 may also include a low noise amplifier control (LNA), a digital variable gain amplifier (DVGA), or a combination of both.

After frontal processing 102 and AGC103, the resulting signals are sent to a sample server 104, which performs the real-time window (for example, the FFT time window) for sampling the sub-carriers within the signal. The output of the sample server 104, which is a synchronized digital signal, is then introduced to an optional frequency rotator, which works in conjunction with, and under the control of, a frequency tracking block to cause rotation or displacement. of the seral phase in frequency in order to make fine adjustments or corrections in the frequency.

The signals, either the sample server 104 or the frequency rotator 106, if used, are sent to the Fast Fourier Transform (FFT) 110, which performs a discrete Fourier transform of the signal. More particularly, the FFT 110 extracts the data carriers from the pilot carriers. The data is sent to a demodulator 112 for data demodulation, and to a subsequent decoder 114 for decoding the data according to any suitable coding scheme used. The decoder output is a bit stream for use by other processors, software or firmware within a transceiver device.

Pilot tones extracted by the FFT 110 are sent to a buffer memory of pilots 116, which stores a number of intertwined pilots of the one or more OFDM symbols. According to an example disclosed in the present specification, buffer 116 may be configured to accommodate multiple interlaced in the buffer for use in the combination of interlaced. The interlaced pilots stored in the buffer memory are delivered by the buffer 116 to a channel estimation unit or block 118, which estimates the channels that use the interlaced pilot tones inserted by the transmitter (not shown) in the signal symbols digital. As will be explained further, the channel estimate provides a channel impulse response (CIR) Ik, n to be used in time tracking and a channel frequency response Hk, n to be used in the demodulation of the channel data by the demodulator 112. The channel impulse response (CIR) Ik, n, in particular, is delivered to a time tracking block 120, which performs an algorithm or a time tracking procedure to determine a time decision for the FFT window that It is used by the sample server 104. The system 100 also includes a processor 121, such as, for example, a digital signal processor (DSP), in communication with the channel estimation unit 118 and can be used to implement various processing operations, such as those to be discussed. later in relation to the procedure of figure 6.

As mentioned above, in a transceiver used in an OFDM system, a channel estimation unit or block (for example, 118) is used to obtain an estimate Hk, n of the channel's channel transfer function in each carrier k and the moment n of the OFDM symbol for the demodulation of the data symbols and an estimation Ik, n of the corresponding channel impulse response (CIR) for use in time tracking. In both DVBT / H and ISDBT systems, in particular, the pilot tones are transmitted in accordance with a predetermined interleaving stepping scheme 200 as illustrated in Figure 2, which illustrates the scheme of the few first carriers k and symbol moments n . As can be seen in Figure 2, at a given symbol time n, the pilot tones p are inserted into each 12th carrier up to a total of up to NK / 12 pilot tones per symbol n of ODFD (for example, at the time of symbol 0 in Figure 3 there may be a number NK / 12 of pilot tones in which the carrier 0 is used for a pilot tone, but NK / 12 1 for symbols that have stepped pilots, such as a symbol moment of OFDM 1, 2 and 3 in figure 2), in which NK is the total number of carriers. For the following symbols, the insertion of pilot tones is compensated with 3 x (n mod 4) tones, based on the time 0 (n = 0). As a consequence, in symbol 1, the first pilot tone is inserted into carrier 3, in symbol 2 the first tone is inserted into carrier 6, and so on. As further illustrated, the pilot tones pl.m are inserted every Ith bearer for a respective interlaced m, in which l is equal to 12 in this example, and m = mod4 (ie 0: m: 3), in The one that mod means a module operation. Thus, after four OFDM symbols (for example, OFDM symbol times 0 3), the pattern is repeated. For example, Figure 2 illustrates the first pilot (i.e., l = 0), the interlaced pattern is staggered for m = 0 to 3, as can be seen by the four pilots p0.0, p0.1, p0.2 , and p0.3 inserted in symbols 0, 1, 2 and 3, respectively.

As an example, known channel estimation algorithms in systems that use the interlacing illustrated in Figure 2, typically combine interlaced pilots from seven (7) consecutive OFDM symbols, which are stored in a buffer memory of interlaced pilots ( not shown), in a paired way to find a channel estimate for a moment n. In particular, each pair of pilot tones corresponds to the same pilot (that is, the pilot I esimo) in the different instances of OFDM symbol times and are combined to estimate the channel corresponding to the data time. As an example of this type of combination, Figure 3 illustrates a diagram 300 of the exemplary interlacing of the pilot symbols p shown in Figure 2 with an additional visual representation of the combination of pilot tones. As illustrated, a first pilot pl.m for l = 0, for example, is combined in time for each of the carriers (ie, interpolated in time). As can be seen in Figure 3, a pair 302, 304 of pilots (p0.1) in carrier 3 (that is, a displacement of three carriers (3 × n mod4), therefore it is part of the interlaced m + 1) and times n +1 and n 3, respectively, are combined at the time of the symbol n (where n = 0 in this example), as indicated by vertical arrows. In addition, an interpolated pilot tone 306 can then be interpolated in frequency with other interpolated pilot tones 308 or a pilot tone existing at the time n of the OFDM symbol 310, as illustrated by the horizontal arrows in Figure 3.

The combination of pilot tones can be performed using any known techniques including interpolation techniques. It is also noted that interlaced can be combined in the frequency or time domain, as will be explained in detail below. From a theoretical point of view, both combination strategies (frequency or time domain) produce exactly the same performance. It is noted, however, that combining in time may present less tension in an IFFT channel in a fixed point implementation (since it is shorter).

When using the pilot dispersion scheme illustrated in Figures 2 and 3, all available positions of scattered pilot tones are used for the combination of pilot tones. As a result, the channel impulse response (CIR) covers 1/3 of the useful OFDM symbol moment (4/3 of the maximum guard).

A first strategy to combine interlaced pilot tones is the combination in the frequency domain, as mentioned above, using a filter. The combination of pilot pilots in the frequency domain can be expressed mathematically, as shown in equation (1) below, which pro

Portion the estimate Hk, n of the pilot tone

In equation (1) above, NP is the length of the final estimate of the channel in the time domain, ml. [nk1 4 are the filter filter coefficients, and Nc and Nnc are the causal and non-causal filter lengths, respectively. It is noted that the notation [14 is an abbreviated notation, in which sub-index 4 is a reminder of the xmod4 module operation. For simplicity only the filtering of pilot tones corresponding to the same interlacing as the filter output is allowed. In other words, the filter works vertically, as indicated in Figure 3 for the example that is currently disclosed in which Nc = Nnc = 3. According to this example, the filter coefficients ml. [nk] 4 are chosen to perform a linear interpolation between two pilot tones, and are shown in Table 1 below. As can be seen in the table, the filter coefficients effectively weigh the weight of the effect of these tones closer to the carrier 0 (for example, k = 1), which in this example, give more weight than the tones (for example, k = 3) farther apart in frequency.

Table 1 Filter coefficients m for linear interpolation

<dl><dt>k </dt><dd> 0 1 2 3 </dd></dl>

<dl><dt>m0.k = </dt><dd> 1 0,75 0,5 0,25 </dd></dl>

<dl><dt>m1.k = </dt><dd> 0 0,25 0,5 0,75 </dd></dl>

It is noted that a more general filter could incorporate pilot tones of interlaced others (that is, it also works diagonally), with a corresponding increase in complexity. After filtering is done

the IFFT of Hk, n leads below a certain threshold are set to zero, and after filling with zeros with 2NP zeros (to interpolate in frequency) an FFT is taken to reach the estimate Hk, n of the final channel in which NP is the length of the final estimate of the channel in the time domain.

Although the combination of the interlaced in the frequency domain, as explained above, is very simple, another strategy is to combine the interlaced in the time domain, as contemplated in US patent application number 11 / 373,764, for a direct link only system (FLO). In a current example, the same time domain combination can be done for DVBT / Hy OFDM systems of ISDBT, for example. However, due to the four (4) interwoven in DVBT / H and ISDBT systems (see, for example, Figures 1 and 2), the mechanics are slightly different than in an FLO system, in which only two ( 2) interlaced are used to obtain the "real" and "excess" channel leads. In the present example, 4 different interlaced, such as those used in DVBT / H and ISDBT systems, are used to obtain four segments of the full channel impulse response (CIR).

First, an IFFT of the pilot tones of each interlaced is taken. More specifically, the

NkNk

zero fill of the (or +1 for interlacing 0) pilot tones Pl.ma NIL, in which NK represents

12 12 sets the number of carriers, and NIL represents the length of the interlaced in frequency after filling with zeros (i.e., extending a signal (or spectrum) with zeros to extend the time limits (or frequency band)). In DVBH systems, for example, the number of NK carriers is 1705, 3409, or 6817, depending on the mode of operation. In ISDBT systems, as an additional example, they typically have 108, 216, or 432 NK carriers depending on the mode of operation. In DVBH systems, for example, the length of NIL interlaced is 256 or 512 or 1024, depending on the mode of operation. ISDBT systems, as another example, would have an interlacing length of 16 or 32 or 64 depending on the mode of operation. After the fill of

Nk

zeros of the tones, an IFFT is taken to obtain a time domain estimate of hk, n of the channel by

interlaced, which is governed by the following equation (2):

In preparation to combine the interlaced channel estimates in the time domain that has a length NIL with a channel estimate with a length NP (in which NP = 4 NIL), the phases of hk, m have to be adjusted. As a consequence, the channel estimate is adjusted according to the following equation (3):

in which bk.m are called interlaced buffers. Because each interlaced channel estimate must be used four (4) times for the calculation of the channel estimates in OFDM symbol times, consecutive, bk, m are stored, which requires at least 7 spaces of NIL complex storage for the example currently disclosed.

Interlaced buffers can be combined to form a channel estimate.

min of time hk, n having a length of NP = 4 NIL. The channel estimate hk, n can then be divided into four segments, as illustrated in Figure 4. Each of the four segments u has a length of NIL, in which each of the segments u can be obtained from the buffers as demonstrated in the following relationship:

For the same ml.k filter coefficients the derivations of the time domain channel obtained here are simply the IFFT of the combined pilot tones of equation (1) above. The combination in the time domain can simply be seen as a way to implement a fast algorithm for the discrete Fourier transform (DFT) of the pilot tones combined in frequency. More particularly, the equivalence is derived as follows for the case in which exactly four consecutive interlaced are used and all four (4) ml.k coefficients of the filter are one (a more general case with the filtering will be considered later) . So every

Time interleaving hk, m can be seen as obtained from a domain channel of the frequency Hk, n by reducing and forward sampling (in frequency). Since the reduction in frequency sampling corresponds to a distortion in time and a shift in frequency to a phase shift in time, one skilled in the art will appreciate that governs the following relationship in equation (5) given below.

With the purpose of the current derivation of the interlaced combination in the time domain, it is assumed that the channel is constant. In this way, to obtain the hk + uNILn of the interlaced hk, n the coefficients kmu, can be found according to equation (6) as follows:

What can be achieved if: 5

which ensures that in the linear combination of equation (6) the coefficients in front of hk + uNIL, add the unit and for all other distortions, the sum of the coefficients is zero.

As an expert in the art will recognize, the solution for akmu is thus

Recognizing also that the relationship

, the distortion and the combination coefficients of the interlacing buffer can be extracted from this solution.

The additional filtering introduced with the coefficients ml, k can be seen to work only with a given interlacing, so that it is equivalent in the domains of time and frequency (i.e., linear operations are interchangeable). If the filtered interlaces are then combined in the frequency or time domain, they are the same according to the methodologies currently described. Consequently, the equation

(4) above can be rewritten as the following equation (9):

in which the inner sum corresponds to the interleaving filtering and the outer sum corresponds to the phase unevenness and interlacing that are combined in the time domain.

As explained above, the combination coefficients (ml, k in this presentation) to combine the interlaced pilots are constant, as can be seen in Table 1 above, in which the coefficients are linearly interpolated over time. However, the coefficients ml, k can be chosen according to different criteria / methodologies. For example, the coefficients can be chosen to minimize the minimum mean square error (MMSE) between the current channel and the channel estimate. It is noted that the design of the combination coefficients of the interleaving filter according to the MMSE criteria exploits the time correlations of the fading process (which are the same in the frequency and time domains).

An exemplary derivation for an MMSE interlaced estimator is as follows. The pilot tones observed Zk.n are supposed to be:

in which Hk, n is the complex channel coefficient of the carrier k at the moment n and fk.n is the complex additive of Gaussian white noise (AWGN). To simplify, it is noted that the widening of the pseudo-random binary sequence (PRBS) is ignored in this explanation. The observations are then combined to form the following estimate:

It is noted that this can be easily extended to more pilot tones and other temporary compensation. For the purposes of this example, however, the perfect knowledge of the second order statistics of the Hk process, n. Due,

in which rHH (l) is the normalized autocorrelation of the fading process in the temporary compensation of l, E indicates the expected value, and C / N0 is the carrier-to-noise ratio.

Applying the principle of orthogonality, as illustrated in equation (13) as follows:

This provides the following equation (14) to find the coefficients m.

in which I is the 2 x 2 identity matrix.

5 When interlaced are combined, either in the frequency or time domains, some time adjustments are necessary due to the phase shift between the pilot tones in a current OFDM symbol n and the previous interlaced. The fine tracking algorithms of known times, for example, delay or advance the position of the FFT window on a sample server (which will be explained later). These time settings correspond to phase shifts in the frequency domain and therefore affect the estimation of the

10 channel: Pilot tones at time n have a phase shift compared to the previous interlaced and, therefore, the channel estimate must be set to correct this phase shift to combine with the interlaced buffer memories. The advance or delay of the FFT window can also be referred to as an advance or delay in the derivation of samples of the OFDM symbol.

Regardless of which methodology used to determine the coefficients of the combination is chosen, in

fifteen OFDM systems the AGC (Automatic Gain Control) can limit the performance of the interlaced combination. As a visual example, Figure 4 shows a graph of the gain of the channel without automatic gain control (AGC). Without AGC, the graph of the channel gain changes smoothly. When AGC is used in a receiver, such as AGC 103, the receiver gain is adjusted so that the samples within a symbol (or more precisely within the FFT window) have a more or less constant power. . This setting

twenty The gain, which may include analog stages (such as a Low Noise Amplifier, LNA) and / or digital stages (such as a Digital Variable Gain Amplifier, DVGA) allows the receiver to operate with fewer bits in the blocks afterwards of adjustment, since the dynamic range of the signal is reduced

As can be seen in Figure 5, the smoothly changing channel of Figure 4 is "chopped" into pieces with discontinuities by the AGC. In addition, this effect of AGC on channel estimation is more pronounced the more

25 interlaced are combined: It is recognized, however, that the performance of the receiver when the interlaced are combined improves if the discontinuities introduced by the AGC are "reversed" or denied. This can be done in the most efficient way by changing the combination coefficients ml, k to reverse the effects of AGC. In mathematical terms, pilot observations on any receiver can be represented by the following equation:

in g (n) it is the gain of AGC (for example, the combined LNA / DVGA) at a time and Zkn represents a theoretical pilot observation, without AGC. The value Zk, n can be further defined as follows:

where Hk, n is the current complex channel coefficient of a carrier k at a time n, and fk.n is the additive

35 Gaussian white noise complex (AWGN). Therefore, an interlaced combination filter in the channel estimation block operates on the adjusted AGC observations according to equation (17) that follows, in order to normalize the AGC gain.

As can be seen in this equation, this normalization is done by multiplying the pilot tone of one between

40 linked by the ratio of the gain of AGC, g (n), of a time symbol n of a gain of AGC g (m) of an interlaced m. For the purposes of the present description, the ratio of g (n) ag (m) is referred to as normalization gain, which serves to normalize the AGC gain at a predetermined time n. It is noted that for the previous relationship (17), in one example, the value of m can be limited according to the condition (n - 3): m: (n + 3) in the case of a combination scheme of 7 interlaced for DVB systems or

Four. Five of ISDB. This may be less for FLO systems or other systems that have interlaced combination schemes of less than 7 interlaced.

It is noted that the AGC adjustment can be performed in the time or frequency domains with the advantage of exactly equal yields. The adjustment can thus be incorporated into the interleaving filter by the definition of an adjusted combination coefficient ml, k according to the following relationship (18).

In equation (18), the combination coefficient ml, k is multiplied by the normalized AGC gain, which can be derived from equation (17). It is noted that for equation (18), a system using 4 interlaced is assumed, such as the system that has been illustrated in Figure 2. Therefore, the value of m can be represented by (n (kl 4)) in a four interlaced scheme. One skilled in the art will appreciate that Equation (18) can be modified to take into account other systems, such as the 2 interlaced system used in FLO systems. This adjusted coefficient can then be substituted in equation (1), for example, to

determine an estimate of channel Hk, n. However, AGC gain is typically not stored linearly, but in the logarithmic domain with b-bit precision, that is, l (n) = rnd (2b log2 (g (n))). Therefore equation (18) becomes:

fifteen The integer part of (l (n) l (n (kl 4))) / 2b in equation (19) corresponds to a simple displacement. Therefore, the power of 2 of the non-integer part can be approximated with a polynomial of degree 2. One skilled in the art will appreciate that Equation (19) can be efficiently implemented in a digital signal processor (DSP). Since the result could exceed the bit width of the FFT machine, the result must be saturated to the bit width of the FFT machine.

Figure 6 illustrates a flow chart of a procedure for determining the combination coefficients in a system of multiple OFDM carriers, in which the coefficients are standardized to take into account the effects of the AGC. As shown, the procedure 600 begins at a start block 602. The flow then proceeds to block 604, in which a normalization gain of an automatic gain control applied is determined. The normalization gain normalized to a predefined moment, such as a moment of

25 symbol n. The procedure of block 604 effects the relationship g (n) / g (m) that has been explained above in relation to equations (17), (18) and (19). After determining the normalization gain in block 604, the flow proceeds to block 606 in which two or more combination coefficients of an interlacing filter are determined. The coefficients can be determined by any of a number of known predetermined criteria, such as, for example, by means of linear interpolation or MMSE as explained above. It is noted that block 606, although shown sequentially after block 604, the operation of block 606 can occur, alternatively, before the operation of block 604 or concurrently with the operation of block 604. In addition, it is noted that a processor 121, such as, for example, a digital signal processor (DSP), channel estimation block 118, a combination thereof, or any other suitable means, may affect the operation of blocks 604 and 606, for example.

35 After the operations of blocks 604 and 606 have been completed, the flow proceeds to block 608, in which the combination coefficients (for example, ml, k) are modified as a function of the determined normalization gain. This operation has been described above in relation to equations (18) and (19), in which a modified, adjusted or adjusted coefficient ml, k is calculated. It is noted that a digital signal processor (DSP), such as DSP 121, channel estimation block 118, a combination of these, or any other suitable means, may affect the functionality of block 608. After that adjusted or modified combination coefficients have been determined, process 600 ends in block 610. The combination coefficients are used by the interlacing filter (for example, 118) to determine a channel estimate, as explained above and also in the related application entitled "TIME SETTINGS FOR CHANNEL ESTIMATION IN A SYSTEM MULTIPLE CARRIER "which has the Agent File number 061615U1,

Four. Five presented simultaneously with this. It is noted that process 600 is repeated continuously during reception and signal processing (for example, channel estimation) in a transceiver.

Although, for purposes of simplicity of explanation, the methodology is shown and described as a series or number of acts, it should be understood that the processes described in this specification are not limited by the order of the acts, since some acts they may occur with different orders and / or in conjunction with other acts than those shown and described herein. For example, those skilled in the art will appreciate that a methodology could alternatively represent a series of interrelated states or events such as, for example, in a state diagram. On the other hand, not all illustrated acts may be required to implement a methodology in accordance with the object methodologies disclosed in this specification.

Figure 7 illustrates an apparatus 700 for determining the combination coefficients for the estimation of the channel in a wireless device. Elaparate 700 receives a gain information from the automatic gain control (AGC) at an input 702, which delivers the signal to a module 704 to determine a normalization gain of a standardized automatic applied gain control at a predefined time. As an example, input 702 can receive AGC gain information from AGC, such as AGC 103 through a communication link 122, as illustrated in Figure 1. In addition, module 704 can be implemented by channel estimation. and the interlaced filter 118, DSP 121, a combination of these or any other suitable process means

The apparatus 700 also includes a module 706 for determining two or more combination coefficients of an interlaced filter based on a predetermined criterion. Module 706 may be implemented by channel estimation block 118 in Figure 1, a DSP (121), a combination of both, as examples, or any other suitable processing means.

The determined normalization gain is emitted by means 704 and two or more combination coefficients are emitted by module 706. Both of these outputs are introduced in module 708 for the modification of the combination coefficients on the basis of the gain of normalization determined. As explained above, module 708 can modify or adjust the coefficients by multiplying the normalization gain by the combination coefficients to achieve the adjusted combination coefficients. It is noted that module 708 can be used to perform one of the equations (17) (19) above. In addition, module 708 can be implemented, for example, by channel estimation block 118, DSP 121, or any combination thereof.

The adjusted decommissioning coefficients are issued by module 708 for use by another process in a transceiver to determine a channel estimate of the OFDM signal received. In a particular example in relation to the determination of the channel estimate, Figure 7 illustrates a module 710 within an apparatus 700 for combining two or more interlaced symbol pilots received in a transceiver using the adjusted combination coefficients. Module 710 can be implemented by the channel estimation unit and interlace filter 118, as shown in Figure 1, as an example. It also notes that the apparatus 700 may be implemented within a transceiver, such as an OFDM transceiver, and may consist of hardware, software, firmware, or any combination thereof.

Figure 8 provides a graphic example of simulation results evidencing a performance improvement obtained by the AGC adjustments that have been explained in the present specification. This figure illustrates the carrier to noise ratio (C / N), which is specified in dB, which is required to achieve a bit error rate after Viterbi decoding (VBER) of 2 x 104 in a typical urban channel with 6 paths (TU6) and maximum variable Doppler frequency, as an example. As can be seen in the figure, the performance of the transceiver has been improved for high speeds. In particular, a transceiver is operated in a maximum Doppler of approximately 100 Hz when AGC settings are used (see, for example, the graph bounded by squares), while without the AGC settings, the transceiver is limited to 70 Hz (see , for example, the graph delimited by rhombuses).

In view of what has been set forth above, one skilled in the art will appreciate that the apparatus and the described procedures perform an improved channel estimation performance in the receiving portion of a transceiver. This is achieved, in particular, by investing the discontinuities introduced by the AGC by determining a normalization gain, which is normalized at a particular symbol moment. This normalization gain, in turn, is used to adjust the combination coefficients used in an interlaced filter to determine the channel estimate.

It should be understood that the specific order or hierarchy of stages in the processes that have been revealed is an example of exemplary approaches. Based on the design preferences, it is understood that the specific order or hierarchy of stages in the processes can be reorganized keeping within the scope of the present description. The accompanying procedural claims present elements of the different stages in a sample order, and are not intended to be limited to the specific order or hierarchy presented.

Those skilled in the art will appreciate that information and signals can be represented using any of a variety of different technologies and techniques. For example, the data, instructions, orders, information, signals, bits, symbols, and chips that may have been referenced throughout the above description may be represented by voltages, currents, electromagnetic waves, fields or particles magnetic, optical fields or particles, or any combination thereof.

Experts will further appreciate that the different illustrative logic blocks, modules, circuits, and algorithm steps described in relation to the embodiments disclosed in this document can be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, several illustrative components, blocks, modules, circuits, and stages have been described above in general in terms of their functionality. How this functionality is implemented as hardware or software depends on the application and design limitations imposed by the general system. Experts can implement the described functionality in various ways for each particular application, but those implementation decisions should not be interpreted as causing a separation with respect to the scope of this description.

The various illustrative logic blocks, modules and circuits described in relation to the embodiments disclosed herein can be applied or performed with a general purpose processor, a digital signal processor (DSP), an application specific integrated circuit (ASIC) , a field of programmable door array (FPGA) or any other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination of these designed to perform the functions described herein. A general purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A processor can also be implemented as a combination of computing devices, for example, a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.

The steps of a procedure or algorithm described in relation to the embodiments disclosed herein may be included directly in the hardware, in a software module executed by a processor, or in a combination of the two. A software module can reside in RAM, in flash memory, in ROM, in EPROM, in EEPROM, in registers, on hard disk, on a removable disk, on a CDROM, or on any Another form of storage medium known in the art. An exemplary storage medium (for example, memory 124 in Figure 1) is coupled to the processor so that the processor can read the information, and write the information, to the storage medium. In the alternative, the storage medium can be integral to the processor. The processor and storage medium may reside in an ASIC. The ASIC can reside in a user terminal. In the alternative, the processor and the storage medium may reside as discrete components in a user terminal.

The examples described above are only exemplary and those skilled in the art can now make numerous use and modifications of the examples disclosed above, without departing from the concepts of the invention disclosed in the present specification. Various modifications to these examples may be apparent to those skilled in the art and the generic principles defined herein may be applied to other examples, for example, in an instant messaging service or any other applications of wireless data communication in general. , without departing from the scope of the appended claims. Therefore, the scope of the disclosure is not intended to be limited to the examples shown herein, but rather the broadest scope consistent with the novel principles and characteristics that have been described herein is agreed. The word "copies" is used exclusively here in the sense of "serving as an example, case, or illustration". Any example described herein as "exemplary" should not necessarily be construed as preferred or advantageous with respect to other examples. As a consequence, the novel aspects described herein should be defined only by the scope of the following claims.