Frequency hopping spread spectrum system with high sensitivity tracking and synchronization for frequency unstable signals

ES 2 349 698 T3 i

DESCRIPTION

FIELD OF THE INVENTION

The present invention relates to frequency hopping spread spectrum radio systems, and more particularly to a spread spectrum radio system using a receiver that follows or tracks and synchronizes with unstable frequency hopping spread spectrum signals to starting from a multitude of low-cost end-point transmitters such as those used, for example, in automatic wireless meter reading systems.

BACKGROUND OF THE INVENTION

Wireless automatic meter reading systems are well known. Each utility meter is typically provided with a battery-powered encoder that collects meter readings and periodically transmits them over a wireless network to a central station. Power limitations imposed by the need for the encoder to be battery powered and by regulations controlling radio transmissions effectively prevent direct radio transmissions to the central station. Instead, wireless meter reading systems typically use a layered network of overlapping intermediate receiving stations that receive transmissions from a group of meter encoders and send these messages to the next higher layer in the network, as described. , for example, in US Patent No.^s5,056,107. These types of layered wireless transmission networks consider the use of unauthorized low-power wireless transmitters in the thousands of endpoint encoder transmitters that have to be deployed as part of a metropolitan area utility meter reading system. wide.

In 1985, in an attempt to stimulate the production and use of wireless network products, the FCC amended Part 15 of the radio spectrum regulation, which controls devices without authorization. The amendment authorized wireless network products to operate in the industrial, scientific and medical (ISM) bands that use spread spectrum modulation. ISM frequencies that can be used include 902 to 928 MHz, 2.4 to 2.4835 GHz, and 5.725 to 5.850 GHz. The FCC allows users to operate spread spectrum wireless products, such as utility metering systems, without obtaining authorization from the FCC if the products meet certain requirements. This deregulation of the frequency spectrum eliminates the need for user organizations to plan

ES 2 349 698 T3 costly in time and money to coordinate radio installations that avoid interference with existing radio systems.

Spread spectrum modulators use one of two methods to spread the signal over a wider area. The first method is Direct Sequence Spread Spectrum, or DSSS, while the second is Frequency Hopping Spread Spectrum, or FHSS. DSSS combines a data signal at the broadcast station with a higher data rate bit sequence, which many refer to as a chip code (also known as a processing gain). A high processing gain increases the resistance of signals to interference. FHSS, on the other hand, relies on distributing a randomly skipped data signal across several defined frequency channels to avoid interference.

The FHSS operates by taking the data signal and modulating it with a carrier signal that jumps from frequency to frequency as a function of time over a wide band of frequencies. With FHSS the carrier frequency changes periodically. The frequency hopping technique reduces interference because an interfering signal from a narrowband system will only affect the spread spectrum signal if both are transmitting on the same frequency and at the same time. Thus, the added interference will be very low, resulting in little or no error.

A hop code determines the frequencies that the FHSS transmitter will transmit and in what order. To properly receive the signal the FHSS receiver is conventionally set to the same hopping code and listens to the incoming signal at the same time and corrects the frequency. However, in order for this approach to be effective, both the FHSS transmitter and the FHSS receiver have to be synchronized with each other in the same hopping code pattern and they have to be keeping track of the same frequency.

Synchronization has to be performed by synchronizing the FHSS transmitter and receiver and at the time as described, for example, in US Patent No.^s5,386,435, but this requires extremely precise clocks in both the FHSS transmitter and receiver or the use of some external channel used to synchronize the clocks. More conventionally, an encoded preamble is used at the beginning of each transmission to synchronize the FHSS transmitter and receiver. US Patent No.^s 6,052,406 describes an FHSS system that uses a correlator to synchronize a sampled incoming data stream with a

ES 2 349 698 T3 known synchronization pattern at the same time as a phase adjustment splits the sampled data stream into first and second sampled sequences. US Patent No.^s 6,052,407 describes the FHSS system for a wireless phone system that constitutes a table of energy data of the spectrum of transmissions in time and uses this data table to further correlate the incoming signals to determine the synchronization with the hopping pattern. of frequency. US Patent No.^s 6,178,193 describes an arrangement that uses a fade period correlated power calculation to adjust the transmit power level of an FHSS transmitter to achieve better synchronization.

The tracking of FHSS transmissions has conventionally depended on the stability of the transmitted frequency. Generally, a transmitter will drift or drift in frequency over time due to aging or changes in temperature or voltage. Traditionally, frequency stabilization circuits have been incorporated into the FHSS transmitter level in order to control and adjust any frequency drift. Synthesizers, such as a phase locked loop (PLL), are used to control or stabilize the transmitter output frequency as described, for example, in US Patent No.^s 5,940,428. Each modulated signal passes through these circuits before transmission. Unfortunately, such PLL circuits produce unwanted power consumption and add significant costs to the FHSS transmitter. For example, in a wireless meter reading system, where cost and battery power are major issues, these unintended consequences of stabilization circuits can pose a significant obstacle to system design and fabrication.

It is possible to eliminate synthesizer circuitry at the level of the FHSS transmitter. Conventional technology adjusts this frequency offset of the transmitter signal by increasing the intermediate frequency (IF) bandwidth of the FHSS receiver to accommodate frequency drift. However, this solution decreases the sensitivity of the FHSS receiver as the IF bandwidth of the FHSS receiver increases. In low power transmissions of an FHSS system the sensitivity of the receiver is essential in order to be able to extract the weak FHSS signals from the background noise.

US Patent No.^s 6,188,715 describes an FHSS system for multi-sensor transmitters that intermittently transmit very status messages.

ES 2 349 698 T3 short. The FHSS receiver uses Fast Fourier Transform (FFT) to detect the transmitted carrier power at several different frequencies in order to improve signal acquisition and timing. Once the FFT determines which frequency contains a signal of interest from the wideband FHSS signal, the FHSS receiver tunes one or more narrowband frequency receivers using a digitally programmable Finite Impulse Response (FIR) filter in response to the output of the FFT and the status of the time and frequency registers representing the frequency hopping code. While this arrangement can improve synchronization, the use of separate narrowband frequency receivers further requires that the FHSS transmitters use frequency stabilization circuitry in order to provide sufficient frequency tracking.

As a consequence of these unwanted obstacles associated with the use of frequency stabilization circuits in conventional meter reading systems there is a need for a low-power, low-cost receiver that is capable of identifying, locating and tracking or tracking signals FHSS received from a transmitter that does not use frequency stabilization circuitry. Additionally, the receiver may be able to adapt to these potentially unstable signals while maintaining high receiver sensitivity.

WO95 / 14064 discloses a low power frequency hopping spread spectrum confirmation location system to allow confirmation location over a long distance.

US587426 discloses a cluster of spread spectrum confirmation locators to minimize transmission collisions.

Document WO88 / 01816 discloses a method for performing the initial synchronization of the transmission and reception of a communication signal with frequency hopping, comprising the steps of transmitting a preamble defined by a sequence of synchronization pulses on a single channel and the transmitting random dummy pulses on the remaining channels to hide the preamble.

SUMMARY OF THE INVENTION

The invention is defined in the appended claims.

The FHSS system of the present invention considers the deployment of end-point transmitters that have free-running local oscillators that are allowed to drift over time, effectively creating a per-hop system.

ES 2 349 698 T3 of unchannelized frequency. Not only can the frequency vary between messages sent by the endpoint transmitters, the present invention also enables a robust receiver design that can tolerate drift even during the transmission of a single message. The end result is a robust receiver that improves overall system performance while allowing a significant reduction in the cost of thousands or hundreds of thousands of endpoint transmitters that are deployed as part of the system.

BRIEF DESCRIPTION OF THE DRAWINGS

Figure 1 is a general schematic diagram of a frequency hopping spread spectrum (FHSS) system in accordance with the present invention.

Figure 2 is a diagram of an embodiment of an encoded FHSS packet that is received in accordance with the present invention.

Figure 3 is a block diagram of a portable meter reading radio transceiver in accordance with one embodiment of the present invention.

Figure 4 is a circuit diagram of the transceiver shown in Figure 4.

Figure 5 is a schematic diagram showing a preferred embodiment of the operation of the FFT algorithm in accordance with the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now to Figure 1, a general schematic diagram of a preferred embodiment of a frequency hopping spread spectrum (FHSS) wireless system 10 in accordance with the present invention will be described. The FHSS system 10 includes a multitude of endpoint transmitters 12 and at least one radio 14,16 having at least one receiver in a preferred embodiment, the endpoint transmitters 12 being battery operated meter reading encoder transmitters operatively connected to a utility accountant. In this embodiment it is expected that any one of hundreds to hundreds of thousands of endpoint transmitters 12 will be deployed as part of an FHSS system 10 installed in a metropolitan area, for example. Alternatively, the endpoint transmitters 12 may be low power sensors, detectors, or other data encoders that can transmit encoded data using an FHSS signal 20. Preferably, endpoint transmitters 12 are deployed at a multitude of fixed locations throughout a total coverage area. Alternatively, the

ES 2 349 698 T3 end point transmitters 12 could be mobile transmitters operating within one or more coverage areas, such as handheld locators or transponders.

In one embodiment the radio comprises a mobile radio 14 that receives encoded data in the form of FHSS signals 20 from a multitude of endpoint transmitters 12 and stores the data for later download or retransmission when a user traverses an area associated with the multitude of end point transmitters 12. Preferably, mobile radio 14 is a portable radio carried by an operator traveling a meter reading path. Alternatively, mobile radio 14 may include one or more radios mounted on a vehicle traveling on streets to collect meter read data. In another embodiment the receivers comprise a plurality of fixed intermediate radios 16 arranged in a hierarchical network of overlapping coverage areas that receive coded data from the endpoint transmitters 12 and relay the data to a central station 18. Reference is made to the previously identified pending application entitled "Spread Spectrum Meter Reading System Utilizing Low Speed ​​/ High Power Frequency Hopping" for a more detailed description of a fixed intermediate radio network 16 and a central station 20 , the discussion of which is incorporated herein by reference. The radios 14,16 can be designed as half duplex radios (they transmit or receive but not both at the same time), however this arrangement has been found to have some limitations. Preferably, the radios 14, 16 are made as a full duplex design (transmit and receive simultaneously). Alternatively, the radios 14, 16 may include only one receiver.

In one embodiment radios 14, 16 are capable of wireless relaying of data 22 to central station 18. Alternatively, the radios 14, 16 can store data until they are manually or automatically downloaded to the central station 18, or the radios can be equipped with other communication channels 24, such as telephone lines, power lines, satellite, mobile phone or similar to transmit immediately or in a store mode and forward the data received from the endpoint transmitters 12, either individually or combined into larger blocks or summarized over time in order to create a measured function associated with one or more endpoint transmitters 12.

It will be understood that endpoint transmitters 12 may be of the bubble variety in which encoded data is automatically transmitted periodically by transmitter 12 (either according to a time pattern

ES 2 349 698 T3 predefined or pseudo-random), or the transmitters 12 can be polled or interrogated to respond to a warning tone, for example, transmitted by the mobile radio 14 or the fixed radio 16 and then transmit the FHSS signals 20 with the data encoded in response to the polling or interrogation signal.

In a preferred embodiment shown in Figure 2 the FHSS signals 20 are preferably sent as encoded data packets 30 transmitted as a frequency hopping spread spectrum signal transmitted in the band between 910-920 MHz as unauthorized transmitters of spread spectrum by hopping operating in accordance with FCC Part 15.249 (transmitter power less than 500 mW) or Part 15.247 (transmitter power less than 5 W). For the purposes of the present invention, transmitters 12 operating within any of these regulations are considered low power transmitters. Preferably, the scrambled packets 30 are sent according to a predefined protocol. Such a protocol is the ERT for meter encoder transmitters manufactured by Itron, Inc., the assignee of the present invention described, for example, in [reference to FCC license]. Another such protocol is the PET defined in the previously identified co-pending application entitled "Spread Spectrum Meter Reading System Using Low Speed ​​/ High Power Frequency Hopping." In a preferred embodiment the encoded data of packet 30 is key modulated on-off (OOK). Other amplitude modulation (AM) techniques can also be used. It is also possible for the encoded data to be modulated using other modulation techniques such as frequency modulation (FM) or frequency drift key modulation (fsk), although additional circuitry may be required to implement these techniques, such as a person of ordinary skill in the art will appreciate.

Contrary to existing approaches to decoding ERT packets, for example, the present invention takes a radically different approach to decoding ERT packets. The normal consumer message packet 30 that is sent by an ERT module 12 is made up of three main components. A preamble 32 begins the message packet 30, and is a series of bits that are always the same. The preamble 32 is used to allow the radios 14, 16 to synchronize with the arriving packet. The body 34 of message 30 contains the consumption, ID, stamp, and type of information for that particular endpoint transmitter 12 at that particular point in time. Final code 36 of message 30

ES 2 349 698 T3 is preferably a CRC (cyclic redundancy check) code containing information that is used to verify that packet 30 was decoded exactly.

In a preferred embodiment the endpoint modules 12, even those that normally operate in the form of a bubble, respond to an alert tone that is an appropriate frequency carrier modulated into the programmed alert tone. The warning tone causes the endpoint transmitter 12 to generate a burst of pulses of some (programmable) numbers of copies of the same message 30. The response from the endpoint transmitter 12 is asynchronous, that is, the message packet 30 can be sent at any time after approximately 3/4 of a second of a valid alert tone. A typical endpoint transmitter 12 will respond to a valid alert tone with eight packets 30, which will be sent at slightly different frequencies (according to a frequency hopping table). In this embodiment the endpoint transmitters 12 provide the ability to select a wide range of warning frequencies (952-956 MHz) and a field of warning tones (28-62 Hz).

As shown in Figure 3, the radio 14 is preferably provided with a transmitter board 40 that can emit the desired warning tones. A receiver board 42 is used to receive the FHSS signals. Preferably, both transmissions and receptions are routed through a common antenna 44 through a diplexer 48. Power and user interface data is provided to the radio 14 through an interface 46. The primary parts of the transceiver 14 include a microcontroller 50, such as a microcontroller 8051, a signal processor 52, preferably a digital signal processor (DSP) 52, a PIC controller 54, the receiver RF circuit section 56, the receiver section 58 of transmitter RF circuits, and diplexer 48. 8051 controller 50 acts as the traffic cop for radio 14. Controller 50 directs data and commands from DSP 52 to processor 54 and sends back status and data from these chips to interface 46. DSP 52 is preferably a Texas Instruments DSP chip and is the mathematical calculator for the correlator, decoder, and from the FFT machine as will be described. The DSP 52 also sets the frequency of the receiving center and switches the attenuation of the front end amplifier on and off. The RF section 56 of the receiver applies an amplifier, mixer, and various sensitive filters to bring incoming ERT packets to the decoder. The PIC controller 54 on the transmitter board 40 programs the transmit frequency, enables the power amplifier, and modulates the RF carrier with the appropriate warning tone. Section 58 RF Transmitter

ES 2 349 698 T3 contains a voltage controlled oscillator (VCO) that creates the outgoing RF carrier, and a power amplifier section 60 to create the high power warning signal. Diplexer 48 is a series of filters designed to allow simultaneous operation of transmitter 40 and receiver 42. Almost all full-duplex radios show some loss of receiver sensitivity with the transmitter enabled, unless the transmitter frequency is very different from the receiver frequency. Radio 14 shows much less receiver desensitization than conventional FHSS receivers except receivers that use a very large and very expensive diplexer for isolation.

In beneficiary Itron Inc.'s earlier FHSS read systems, the receiver was designed to sample bit-segmented data from the detector's output to search for ERT 30 packets. These earlier systems used the approach, so to speak, that if a packet 30 looks like an ERT packet, and it looks like an ERT packet, so it has to be an ERT packet. That is, the receiver samples the aspect to look for the preamble 32 of an ERT packet 30, and when the receiver has recognized the particular sequence of bits of the preamble 32 the receiver synchronizes with the timing of the bits of the preamble 32 (and sometimes the first body bit pair 34), and then uses that timer to decode the remaining bits of packet 30.

The receiver 42 of the radios 14, 16 according to the present invention goes one step further. Receiver 42 samples the detector output without segmenting the data bits. The preamble 32 of the packet 30 is always the same, that is, the receiver 42 knows how to search to see the beginning of a packet 30. This allows the receiver 42 to use some mathematical concepts known as correlation preferably applied by a correlator. According to the present invention a correlator is formed by circuits or a processor or controller programmed to compare the incoming bit stream with the known values ​​designed in the message. In the embodiment shown in Figure 3 the correlator is applied in the DSP 52. The correlator gives low correlation values ​​until an ongoing match is observed (preferably in about twenty bits). In this case, the output of the correlator becomes very high. Thus, with the present invention, there is no doubt as to whether the packet 30 is a valid ERT packet, the correlator allows the radio 14, 16 to know if the packet 30 is a valid ERT packet (within the statistical probability of a false match ). Not only that, but the receiver 42 has an exact timing of the packet bit stream, which allows

ES 2 349 698 T3 radio 14, 16 decode the remaining parts of packet 30 in the center of each bit, which increases the number of clearly decoded packets.

In earlier Itron receivers, when the ERT packet 30 was weak (near the low noise level at the receiver) it was impossible to distinguish the ERT packet from noise. With a correlator applied in accordance with the present invention the receiver 42 can effectively detect the presence of a valid packet below the base noise level (which is commonly referred to as searching in the noise). The preferred embodiment of the correlator can produce a sensitivity increase of at least 12 dB over existing Itron receivers. As will be described, for weak signals, the use of the FFT engine by DSP 52 can also provide an additional 9 dB of improvement in sensitivity when analyzing the end portion of the preamble to determine the narrow band frequency contained in packet 30. .

In a preferred embodiment endpoint transmitters 12 send packets 30 in the 910-920 MHz band. The vast majority of existing endpoint transmitters 12 send some or all of their packets 30 in the 913-918 MHz band. Previous receiver designs of the receiver needed to limit the width of the aspect that can be searched at any one time to about 1.4 MHz in order to achieve the desired receiver sensitivity. The advanced digital signal processing techniques used in the receiver 42 of the present invention achieve the same sensitivity as previous radios, but the receiver 42 can see a 7 MHz window across the broadband frequency. In previous designs it was possible that the receiver window would not be where the endpoint transmitter 12 was transmitting at the time that the transmitter 12 was transmitting the packets 30. Previous receivers would scan the ERT transmission band (actually older receivers scanned less than the full 10 MHz band, typically around

1.4 MHz in one go) looking for 30 packets, but could sometimes lose a packet if it was transmitted in a part of the band that was not being scanned at the time. In the receiver 42 of the present invention, at least about sixty-six percent (66%) and preferably all of the useful bandwidth is monitored at all times. If an ERT packet 30 is being transmitted, and the FHSS signal 20 for such a packet is reasonably strong at receiver 42, it will recognize that packet.

ES 2 349 698 T3

The receiver 42 of the preferred embodiment preferably scans the entire useful portion of the broadband for simultaneously looking for a correlation output signal indicative that there is a packet in the environment. In this embodiment, most ERT transmitters 12 transmit with frequencies between 913 MHz and 918 MHz, so that receiver 44 examines a 7 MHz band for correlation. The advantages of correlation are perceived because the correlator knows that the preamble of an ERT 30 packet is similar, and looks for a match. The receiver 42 has no way of using a correlation technique to detect or decode the remainder of the ERT packet 30 because after the preamble 32 the bit pattern in the body 34 and the final code 36 will vary randomly according to the content of encoded data, such as ID, consumption, CRC, or other variable data information. The present invention executes the DSP 52 as a correlator on a significant portion of the FHSS broadband signal 20 to detect the existence of the preamble 32. Once the packet 30 has been detected because of the preamble 32, then the receiver 42 has a signal processor 52 that performs a Fast Fourier Transform (FFT). From the FFT the broadband frequency in which the packet 30 is being transmitted by the transmitter 12 is effectively determined.

Contrary to existing techniques that use a broadband signal processing algorithm, the present invention uses the broadband signal processing algorithm to also decode from the data obtained from the FFT that is contained in the remaining part. of package 30. Preferably, as the data samples are collected by FFT decoding the wideband signal is also sampled and decoded by a wideband circuit in the form of a received signal strength indicator (RSSI) amplifier 62 coupled to a 12-bit analog-to-digital converter 64 shown in Figure 4. If the data can be decoded because of good signal strength in wideband mode by the wideband circuit, the FFT decoding is aborted. This saves computational time and allows receiver 46 to start searching for another transmission.

Preferably, the DSP 52 of receiver 42 uses a technique known as Fast Fourier Transform (FFT) or other comparable advanced digital signal processing to process the broadband signal. The FFT engine coded and executed by the DSP basically fragments the 7 MHz band into thirty-two identical channels of approximately 250 MHz. The FFT shows the presence of

ES 2 349 698 T3 ERT transmit power on one of these channels. It would be very difficult to tell just by looking at an FFT data stream when an ERT packet is circling amid random noise. However, remember that the correlator is able to tell the FFT machine almost exactly when the ERT data is flowing through the tube. The FFT machine is then able to search all thirty-two channels and determine where the ERT data is coming from from the pattern of changing data. Preferably this is done by mapping into 34 of 40 bits in preamble 32. The last six bits are used to search for the target channel container. Since the expected state of the bits is known as they are part of preamble 32, the FFT channel containers are scanned for the container that contains the strongest representation of the expected bit pattern of preamble 32. The FFT engine is much more sensitive than that of the previous version of the receivers due to the amount of noise with which it competes is much less in the narrow channels.

A further feature of the preferred embodiment of receiver 42 is its ability to track or trace an ERT packet 30 if the FHSS signal 20 drifts in frequency. This is done by decoding the target channel, ie the channel with the strongest signal in the ERT packet 30. If the FHSS signal 20 drifts in frequency and is no longer present on the target channel, the decoding will fail in the CRC check. Receiver 42 saves all FFT information during the decoding process and will again execute the decoding operation on the averaged target channel along with the next contiguous channel. This is done on each contiguous channel giving receiver 42 the ability to track or track an ERT that is drifting at a frequency higher or lower than the target channel frequency. Therefore, in the receiver 42 of the present invention the FFT produces better sensitivity than any of the existing FHSS receiver for encoders 12 of the low power transmitter.

Because the FFT works best on weak signals, and because the math takes extra time to execute an FFT, the receiver 42 preferably has two different stages of operation. Receiver 42 will actually sample a signal called RSSI (Received Signal Strength Indicator) for strong FHSS 20 signals, and will switch in the FFT mode for weaker FHSS 20 signals. This dual-mode capability is automatic (the receiver makes a power determination on the fly and adjusts accordingly) and enables excellent reception of FHSS 20 signals at all levels.

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Referring now to Figure 4, a detailed circuit diagram of the radio 14 will be described. The main parts of the receiving board 42 of Figure 4 are the circuitry 62 of the linear detector, a frequency generator 72, a mixer 74 , an IF amplifier 76 and a filter 78, the analog-to-digital (A / D) converters 64, 66, and the DSP 52. The RF signal received from the antenna 44 is received in the range 910-920 MHz. The IF frequency generator 72 generates a frequency in the range 840-850 MHz. To create the IF signal 80 the signal from the IF frequency generator 72 is injected on the low side to the mixer 74. Following the use of the mixer 74, an amplifier IF 76 and an IF filter 78 are used to create IF MHZ signal 70. The signal is then fed into DSP 52 after passing through high speed A / D converter 64 and low speed A / D converter 66. The high speed 64 A / D converter samples the 70MHZ signal. This is in signal sampling, which is the digital equivalent of a mixer. In the sampling, the 70 MHZ IF is transferred to the base band. The samples are then fed to DESP 52, which executes a 64-point FFT, as will be described. The 64-point FFT creates thirty-two single frequency bins, each 256KHZ wide.

The components of the receiver board 42 of the Figure 4 receiver of a preferred embodiment of the present invention also include an RF amplifier 70 which preferably is composed of two linear amplifiers (LNA) and a surface acoustic wave filter (SAW) to produce a 30 dB gain and a 1.2 dB NF NF. One of the LNAs has a NF of 9 dB, a P1 dB of -22 dBm, and an IP3 of 10 dBm. The other LNA has an NF of 1.6 dB, a gain of 17 dB, a P1 dB of +12 dBm, and an IP3 of 10 dBm. Mixer 74 preferably has a CG of +9 dB, a P1 dB of -7 dBm, and an IP3 of -8 dBm. The IF amplifier 76 preferably has a P1 dB of 9.5 dBm and an NF of 5.5 dB. The IF filter 78 preferably has a bandwidth of 7 MHZ and an IL of 9 dB. The high-speed A / D converter 64 is preferably a 12-bit converter operating at a speed of 16,384 MSPS, while the low-speed A / D converter 66 is preferably a 12-bit converter operating at a speed of 262,144. KSPS.

When the correlator of the present invention compares a known data pattern with sampled data the preferred sampling rate is 8 times the data rate, in this case 262.144 kHz. This sampling is performed by a low-speed 12-bit A / D converter 66 connected to the IF's RSSI 62. When a correlation occurs, the output of the correlator is synchronized within 1/8

ES 2 349 698 T3 one bit. This begins the timing to decode the remainder of the packet 30 from this point on. Preferably, packet 30 decoding uses the center 3/4 of each bit and the first 1/8 and last 1/8 of each bit are discarded due to uncertainty. A particular advantage of this embodiment is that it enables the present invention to perform effective data sampling at a lower rate than the sampling rate prescribed by Nyquist's theorem (i.e., sampling has to occur at more than twice the rate effective data transfer). In one sense, the correlator effectively allows the decoder to operate as a synchronous power detector for the broadband signal.

Preferably, preamble 32 of packet 30 consists of 20 bits plus a sync bit. This data is Manchester encoded so that we have 42 possible "chips" (ie transition states) to correlate. The first 34 chips are used to correlate and the last 6 to determine the best container for data decoding. There is a 2-chip gap between mapping and decoding to allow for processor installation. The last 6 chips are preferably in a 1 0 1 0 pattern, so that once a correlation has been determined, the high-speed A / D converter 64 can sample the 70 MHz IF. Preferably, the 70 IF signal MHz is actually undersampled, however the frequency is shifted.

In this embodiment the IF 80 signal sampling rate is 16.384 MHz. This data is fed to DSP 52 by high speed A / D converter 64, which allows DSP 52 to perform FFT operation at twice the speed of the data rate, or 65.536 kHz. A 64-point FFT is performed that produces thirty-two frequency bins 82 as shown schematically in Figure 5. After 12 sets of samples have been converted by the FFT machine each frequency container 82 is evaluated for energy in matching the pattern of the last six chips. For each container 82 all samples expected to be of a value 1 (shown schematically at 84) are added and the sum of all values ​​expected to be zero (shown schematically at 86) is subtracted from the sum 84 to give a value of the energy 88 of each container 82. If a container 82 contains random noise the total value 88 of the energy of that container will be very low since the sum of six noise values ​​minus six noise values ​​is a low noise value. If there is data present and of the correct timing to match the exactly correlated preamble, the ones will add up to 6 times the present energy

ES 2 349 698 T3 average, and then the noise will be subtracted six times. Actually the sum of 1 84 values ​​will be the signal plus the noise, and the sum of the zero values ​​86 will be subtracted until the noise just leaves the sum of the signal values. At the extreme the container 82 that has the highest sum of signals for the energy value 88 is considered as the container that contains the signal. Once the best container is known, that container 82 is marked with a signal and the sampling and FFT conversion process continues on the remainder of the data stream.

DSP 52 preferably has sufficient computing power to get all samples in 2 times oversampling of the data; however the preferred DSP can only perform one FFT per bit time. In this embodiment data is entered into the buffer and a second set of FFTs is performed in a post-processing operation. This supposes a slight delay in the ability of the radio 14, 16 to re-decode new packets so that a sample is also taken from the low speed A / D converter tied to the RSSI line. It operates at 8 times the oversampling speed of the correlator. The values ​​are used to test and decode the data. If the signal is strong enough the data can be decoded from RSSI alone. If the decoding is useful then the DSP dumps the FFT data and starts the correlator again looking for another data packet. If RSSI decoding is unsuccessful then DSP 52 performs the second set of FFTs and tries to decode in the best container. If this is not successful then the DSP 52 averages the best bin to the next higher bin and performs another decoding. If this is not successful try again to average the best container with a lower container. As soon as one of the attempts is successful then the data is dumped and the correlator starts over. If all attempts fail then the packet is marked as bad data. By averaging the adjoining containers with the best container the radio can track or trace a signal drifting from one container to another.

As previously described, the RSSI voltage from IF stage filter 78 is sampled with an A / D converter 66 and input to DSP 52. Preferably, the sample is placed in a DSP 52 internal sample stack with the oldest sample shifting out of the stack when the newest sample is shifted in, preferably the stack is thirty-four samples long, with each sample representing one of the bits of the preamble 42 to be mapped. It should be noted that the length of the stack is not

ES 2 349 698 T3 critical to the operation of the present invention. Good correlations have been shown using a subset of this number of samples. The number of samples (thirty-four) is actually a subset of the entire ERT 32 preamble that is transmitted. After the new sample has been taken and saved, each sample in the stack is compared to a value that represents the known and expected preamble. In a preferred application if the known value of a bit in preamble 32 is expected to be a binary one it is assigned a value of one, whereas if the known value is expected to be a binary zero the bit is assigned a value of a negative one. The known value is compared by multiplying the unknown value of position one by the known value of position one. The unknown value of position two is then multiplied by the unknown value of position two and so on until the thirty-four unknown values ​​have been multiplied by their corresponding known values. Then the results of all the multiplications are added. This gives a correlation value to the current set of samples. Another sample is taken at the appropriate time and the entire process is run again. The appropriate time is determined by the data rate of the ERT message. As previously described in a preferred application the signal is oversampled 8 times since the list is 8 times longer than the list of known bits. The sampling rate is also 8 times faster than the data rate. This allows the preferred embodiment to more accurately synchronize with the data.

The correlation operation is preferably a digital convolution application, in which a known function of the preamble 32 is compared to an unknown function, the data samples. Correlation on 34 data chips effectively increases the sensitivity of the radio by averaging the data sample in all 34 bits. In practice this allows detection of a preamble 32 with 6 to 12 dB of better sensitivity than it can be decoded. The output of the correlator is compared to a threshold value. The threshold is determined by taking an average of up to 256 raw input samples. This average is preferably a measure of the intensity of the input signal and the correlator is a measure of the probability of a preamble. If there is a strong signal the correlator value and the correlator threshold increase. Similarly, both values ​​decrease at low signal levels. What is being examined is the relationship between the two values. The actual threshold is preferably obtained empirically and is a percentage of the average of the signals. If he

ES 2 349 698 T3 correlator exceeds this value the present invention assumes that a preamble 32 is present. The probability of detection can be adjusted by setting the threshold level.

The lower the level, the greater the probability of finding a preamble and the greater the probability of a false detection.

Once a preamble is detected there are preferably 8 chips left in the preamble 32 that still have to be read. At this point the high speed A / D converter 64 is activated. Although the A / D converter 65 could have been running uninterrupted, preferably the high speed A / D converter 66 is turned off to save power. The high-speed A / D converter 66 samples the IF 70 signal. The samples are then fed to DSP 52 which executes a 64 point FFT to create thirty-two single frequency bins 82, each 256 kHz wide as previously described. Preferably, each frequency container 82 is represented by a complex number that is converted to power by squaring the real and imaginary parts and adding them together. The start of this process takes a short time, preferably the two data samples of the incoming preamble are discarded. This leaves six bits of the incoming preamble of which the expected state is known. Since the DSP knows what it is looking for, the DSP 52 examines the frequency bins 82 of the next six bit samples of the known pattern. When there is a container 82 that contains a pattern 10101 (representing 12 known chips) as well as the expected preamble, that container is rated as the best container described in connection with the description in Figure 5. This is essentially how it is done. the initial “frequency setting”. It is not really a frequency setting but rather it determines the frequency on which a transmission is occurring. Once the best container has been determined the DSP uses the FFT algorithm to try to decode the body 34 of the ERT 30 message on that channel. Since the ERT message 30 contains a CRC error detection byte 36 it is possible to determine a successful decoding if the decoding passes the CRC test.

While decoding is taking place, preferably all FTT processing data is saved. If the decoding fails to find a good CRC test the receiver may have missed the ERT 30 message because the FHSS 20 signal had a drift in frequency, or was exactly on the border between two containers 82. In this case the decoding process FFT is repeated on the data representing the best container averaged with the next

ES 2 349 698 T3 next higher container. If the message drifted higher in frequency then it would have some of its energy shifted to the next higher container 82. If the message drifted to a lower frequency then it would have some of its energy shifted to the next lower container 82. Although the preferred embodiment uses a single contiguous container average it will be apparent that other combinatorial schemes could also be used to accomplish this process. The use of container data contiguous at the frequency associated with the best container allows the preferred embodiment of the present invention to track or track the ERT 30 message in frequency. If the CRC fails averaging in the highest contiguous container then decoding is performed by averaging the best container with the lowest contiguous container following or tracking the lowest frequency message. This decoding process did not adjust the oscillators on the radio. Tracking is a result of the output of the FFT frequency containers and the signal is tracked or tracked in the containers.

Another difference between the preferred embodiment of receiver 42 according to the present invention and previous versions of FHSS receivers is that legacy radios have always used an ERT signal detector with a logarithmic response. This means that the signal detector would see well both weak signals and strong ERT signals, also known as having a wide “dynamic range”. The problem with logarithmic detectors is that they are susceptible to the presence of an interfering signal, which basically hangs the detector to one side and will not allow the detector to effectively see ERT signals. The effect of this problem would be complicated by the much wider window for detection inherent in the receiver 42. To overcome this problem the receiver 42 uses a linear detector 62 that can see small ERT signals with and without the presence of other energy interfering in the band. The problem with linear detectors is that they usually show a small dynamic range. The receiver 42 of the preferred embodiment of the present invention compensates for this by using a front end amplifier 71 which is very sensitive to weak signals and which is attenuated to muffled voiced ERT signals. This essentially gives the best of both worlds, a wide dynamic range, and a low sensitivity to in-band interference factors.

It will be seen that because the weak signal processing by the FFT machine is a digital signal process that can be repeated in the data

ES 2 349 698 T3 stored, the present invention allows the ability to simultaneously process multiple message packets by multiplexing the digital sample stream on multiple FFT machines, which can be performed as segmented threads on a single DSP or by multiple DSPs, depending on the sampling rate 5 and the processing power of the DSPs.

Although the present invention has been described with respect to the preferred embodiment, it will be understood that numerous changes and variations can be made in the aspects of the invention and that the scope of the present invention is intended to be consistent with the claims that follow.