Systems and methods for loop length and bridged tap length determination of a transmission line

ES 2 265 411 T3

DESCRIPTION

Systems and methods to determine the length of the loop and the length of bridged taps of a transmission line.

This invention relates to the determination of the characteristics of a transmission line. In particular, this invention relates to a system for estimating loop length and bridged tap length, to a method for determining a loop length and at least one bridged tap length for a transmission line, and to means of storage of information that comprise the respective information.

The collection and exchange of diagnostic and test information between transceivers in a telecommunications environment is an important part of a telecommunications deployment, such as an ADSL (Asymmetric Digital Subscriber Line). In the case where the transceiver connection is not performed as expected, for example when the data rate is low, when there are many binary errors or the like, it is necessary to collect diagnostic and test information from the remote transceiver. This is done by sending a technician to the remote location, for example a visit from a repair van, which is time consuming and expensive.

In DSL technology, communications over a subscriber local loop between a central office and a subscriber establishment are performed by modulating the data to be transmitted on a multiplicity of discrete frequency carriers that are added together and then transmitted over the subscriber loop. Individually, non-overlapping discrete communication subchannel carriers are bandwidth limited. Collectively, the carriers form what is effectively a broadband communications channel. At the receiving end, carriers are demodulated and data is recovered.

DSL systems experience disturbances from other data services and adjacent telephone lines, such as, for example, ADSL, HDSL (High bit rate Digital Subscriber Line), ISDN (Integrated Services Digital Network - Integrated services digital network), T1 (leased or dedicated line) or similar. These disturbances may begin after the target ADSL service has already started and, since DSL for Internet access is considered a permanently operating service, the effect of these disturbances must be mitigated by the target ADSL transceiver.

The identification, measurement and characterization of the condition of a transmission line is a key element of an ADSL deployment. In the case where the transceiver connection does not go as expected, for example because the data rate is low, there are many binary errors, a data link is not possible or the like, it is important to be able to identify the length of the loop and the existence, location and length of any bridged outlet without having to dispatch a technician to the site of a remote modem for diagnostic testing.

The article "The Modeling Aspect of Transmission Line Networks" by Patrick Boets et. al., Proceedings of the Instrumentation and Measurement Technology Conference, US, New York, IDEE, May 12, 1992, pages 137 to 141, ISBN: 0-7803-0640-6, describes a fault recognition system for networks of lines of transmission. A synthetic reflectogram is formed in the frequency domain using models that describe the attenuation and dispersion of an impulse propagating in a uniform transmission line. This synthetic reflectogram is compared to a digitally recorded reflectogram in the time domain, so that the first reflection caused by a fault can be recognized. The delay between the previously located fault and the reflectometer can be measured. This delay is converted into a distance with the help of the speed of propagation to provide an operator with information about the section of the network where the fault may be.

US 5,128,619 A describes a method and system for automatically determining the length, attenuation, impedance and existence of bridging taps in installed communication cables. A reflectometer is also used. A waveform that includes incident and reflected pulses is analyzed to determine the presence of significant pulses. The presence of more than one significant positive impulse or one negative impulse indicates the presence of at least one fault in the cable.

US 4,630,228 A describes a transmission line analyzer for automatically detecting at least the location of discontinuities in a line, comprising frequency domain reflectometer means for providing a complex, composite, time-varying signal containing location information. A fast Fourier transform means automatically processes the complex signal to determine spectral peaks which are then converted to the location of the discontinuity.

US 5,994,905 A describes a frequency domain reflectometer and a method of locating faults in a transmission line used by the reflectometer. The reflectometer suppresses harmonics that can be interpreted as faults that do not actually exist. In general, the reflectometer applies a scan signal to the transmission line in order to obtain a reflected scan response signal. The reflectometer then obtains a sweep response spectrum from the respective sweep response signal. The reflected sweep response signal includes a plurality of spectral peaks representing the frequency components of the reflected sweep response signal. The reflectometer then mathematically determines the spectral peaks of the reflected swept response spectrum that were generated due to harmonics in the response signal of

ES 2 265 411 T3 reflected scan. After determining that a spectral peak is a harmonic of a fundamental spectral peak, the reflectometer subtracts from the harmonic spectral peak a percentage of the fundamental spectral peak, thereby suppressing a harmonic from the reflected sweep response signal. The reflectometer then obtains from the adjusted scan response spectrum a location of an impedance mismatch in the transmission line.

US 5,068,614 A describes an improved swept frequency domain reflectometry instrument having an input / output terminal coupled to one end of a cable. A radio frequency at the input / output terminal is converted to an intermediate frequency using a local swept frequency. The radio frequency is derived from the local swept frequency and a variable offset frequency. A controller controls the variable offset frequency, monitoring the amplitude of the intermediate frequency. A periodic counter counts the intermediate frequency at a first maximum response of the intermediate frequency and further counts the offset frequency at a second maximum response of the intermediate frequency. The location of a discontinuity in the cable is calculated from the counted frequencies.

It is an object of the present invention to provide an information storage system, method and means for determining a loop length and at least one bridged tap length for a transmission line, where a simple and effective determination is possible.

The above object is achieved with a loop length and bridged tap length estimation system according to claim 1, a method according to claim 5 or information storage means according to claim 9. Preferred embodiments are the subject of the claims subordinate.

This invention describes a system and method for estimating the loop length, the number of bridged taps and the length of the bridged taps on a transmission line from readily available modem data. The loop length, the number of bridged taps, and the length of the bridged taps can be estimated by comparing a measured frequency domain channel impulse response of the transmission line with a model of a transmission line that is composed of multiple sections and multiple you take a bridge. Diagnostic and test information describing the condition of the line can then be exchanged, for example, by two transceivers during a diagnostic link mode.

These and other features and advantages of this invention are specified in or are apparent from the following detailed description of the embodiments.

Brief description of the drawings

Embodiments of the invention will be described in detail with reference to the following figures, in which:

Figure 1 illustrates an exemplary multi-section loop with multiple bridged taps;

Figure 2 illustrates a graph of the measured received reverb signal and the theoretical model for downstream data;

Figure 3 illustrates a graph of the measured received reverb signal and the theoretical model for upstream data;

Figure 4 is a functional block diagram illustrating an exemplary loop length and bridged tap length estimation system in accordance with this invention;

Figure 5 is a flow chart summarizing an exemplary general method for determining loop length and bridged tap lengths in accordance with this invention;

Figure 6 is a flow chart summarizing an exemplary method for estimating loop length and bridged tap length in the upstream direction in accordance with this invention; Y

Figure 7 is a flow chart summarizing an exemplary method for estimating loop length and bridged tap length in the downstream direction in accordance with this invention.

Detailed description of the invention

Exemplary embodiments of this invention will be described in connection with the application thereof to an ADSL transceiver environment. However, it should be appreciated that, in general, the systems and methods of this invention will work equally well for any multi-section loop with one or more taps bridged.

For example, during ADSL modem initialization, the subscriber loop frequency domain channel impulse response is measured at a set of discrete frequency values. The measured frequency values ​​are designated as H_m(f¡) and fj = i Af, for i = 0, 1, ..., k-1, where Af is the frequency spacing between adjacent samples.

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Figure 1 illustrates an exemplary model of a loop with N sections and M bridged taps. The frequency domain model for the channel impulse response for the loop in Figure 1 can be written as H (x, f), where f is the frequency and the vector x contains the lengths (d_i) of the N loop sections and the lengths (b,) of the M bridged taps:

x = [di, d2 ..., d_N, bi, b₂ bM]

Assuming that the number of sections of the multi-section subscriber loop, N, and the number of bridged taps, M are known, an estimate of the optimal parametric vector x that best approximates the response H can be determined_m(fj) of measured channel impulse, given the model H (x, f). The set x * of optimal parametric vectors can be estimated by minimizing the norm of the difference between the measured frequency response and that of the model, to the discrete frequency values ​​f_i = i Af, for i = 0, 1, ..., k-1. This minimization can be done using the expression:

<img file="ES2265411T3_D0001.tif" />

If the number of bridged taps in the loop is not known, assuming a large number of bridged taps in the model's frequency response and assuming that the minimization will converge to a solution with the correct number of bridged taps with non-zero length, the taps The remaining bridged will have zero length.

The frequency domain model H (x, f) can also incorporate the effect of, for example, an imperfectly matched transmission line, including the effects of load and source impedances.

More particularly, loop characterization algorithms employ a model-based approach to estimate the length of the loop and the lengths of up to two bridged taps. A channel characterization algorithm compares the measured channel impulse response with the channel impulse response of the loop model, which consists of a single gauge wire and contains up to two bridged taps. However, it should be appreciated that the basic model can be extended to include multiple gauge wires and multiple bridged taps. The loop length and the lengths of the bridged taps are the parameters of the theoretical channel impulse response. The system modifies the theoretical molding parameters and evaluates the difference between the measured channel impulse response and the theoretical channel impulse response. The loop length / bridged tap lengths that minimize the error function are then declared as estimated values. The presence of a bridged tap is declared if its length is greater than a predetermined length, such as one hundred feet (about thirty meters). This threshold for the detection of bridged taps was established experimentally. It was determined that for most loops there is a chance that a phantom bridged tap with a small length will be detected due to modeling inaccuracies and noise in the measurement system. Since the lengths of these bridged ghost taps were almost always below 100 feet (about 30 m), the exemplary threshold was set at 100 feet (about 30 m). However, in general, the threshold can be altered depending on the particular operating environment and the complexity of the model.

There are two independent algorithms that perform loop characterization for downstream data (DS) and upstream data (US). During modem initialization, the data collection software collects the reverb signal by averaging K consecutive frames, where K&gt; 64. However, it should be appreciated that the more averaging performed, the less noisy the measurement will be. However, since there is a prescribed number of frames in the training of a standard modem, in which the reverb signal is transmitted, the exemplary number of averages was set to 64. The received reverb signal obtained in this way is a estimation of the impulse response of the entire channel, including the front-end responses of the transmitting and receiving modems. The reverb signal received in the frequency domain is obtained according to:

κ

Rx (f) = ^ Σ FFTN (rx (n))

K k = i (1) where f is a dummy variable denoting frequency and rx (n), for n = 1,. . . , N, are the samples of the reverberation signal received in the time domain within the frame, where N is the number of samples contained in a single frame. Equation 1 may contain a slight abuse of notation because, in reality, the frequency variable f is not continuous, but discrete, and for this reason the channel impulse response is available at a set of discrete frequencies called tones that are multiples of Af = 4312.5 Hz:

F_i = iAf, i = 1 ..., N / 2 (2)

ES 2 265 411 T3

The reverb signal is transmitted over a portion of the entire ADSL spectrum. For example, the reverb signal is available in 224 (96 in G. Lite) tones of f₃₂ = 32Af to f₂₅₅ = 255Af in the downstream channel and in 26 tones of f₆ = 6Af af₃₁ = 31Af in the upstream channel. The downstream reverb signal is collected at the customer site equipment (CPE) and the upstream reverb signal is collected at the central office (CO). Although there is no difference in the data collection process for the upstream or downstream reverb signal, the characteristics of these two data sets are quite different. Specifically, the needle reverb data below contains significantly more information. In addition, there are more samples of the frequency domain reverb signal available in the downstream direction and these samples cover a large range in the frequency domain where the effects of bridged taps on the impulse response can be easily detected. However, there is a crucial difference between upstream and downstream data sets that is complicated by using the same interpretation algorithm for both. In the downstream channel, the matching of the leading end impedance to the loop impedance tends to be better than in the upstream channel. This makes it possible to use a simplified channel model for the downstream channel. Unfortunately, the impedance matching in the upstream channel is generally not as good as in the downstream channel and a more complicated channel impulse response should be used.

Due to these complications in channel modeling and the lack of sufficient data samples, the basic upstream channel characterization algorithm is limited in terms of estimation precision and the number of bridged taps that can be detected. However, by extending the channel model to include multiple sections of varying gauges and / or more than two bridged taps, the presence of more than two bridged taps can be detected and more accurate results can be achieved for individual loop section lengths if there is a change in yarn gauge along loop. The only concession is that as the number of model parameters increases, the computational effort required to estimate the parameters will also increase.

The following describes the theoretical details leading to the derivation of the frequency domain channel impulse response from the model and explains in detail the characterization of the channel for the upstream and downstream data. The downstream and upstream interpretation algorithms both employ the same least squares minimization concept, where the square of the error norm between the actual and theoretical channel impulse responses is minimized, but differ in the theoretical impulse response of channel used.

For the characterization of the downstream data loop, an exemplary two-wire loop is distinguished by its characteristic impedance:

Ζο (ω) =

R + ¡ω !, G + jmC

And its constant of propagation:

Y (f) = ^ / (R + jmL) (G + jmC) where ω = 2 nf is the angular frequency and R (resistance), L (inductance), G (admittance) and C (capacitance) are the dependent constants loop frequency and vary with wire gauge. For a perfectly finished loop or a very long loop, with length d, and two bridged taps of lengths b1 and b2, the transfer function of the loop H (d, b₁, b₂, f) is given by:

g-dy (f)

H⁽d, h, h_;,^F) [2 + tgh (bi γ] · [2 + tgh ^ y)] (3)

In logarithmic scale:

log | H (d, b1, b2, f | = log (2) - dy (f) - log [2 + tgh (b ©] - log [2 + tgh (b2γ)] (4)

Note the dependence of loop loss on cable length. The actual transfer function of the loop can be measured during modem initialization. The measured transfer function of the loop is then matched to that of a loop of length d with two bridged taps, as given in Equation 3. In other words, determining d, b1, and b2 minimizes the following least squares error criterion:

<img file="ES2265411T3_D0002.tif" />

where Rx (f_i) is the received reverb signal sampled at f_I = if ei_i are the first and last tones Rx (f_i).

ES 2 265 411 T3

An example of algorithm operation for an exemplary loop is illustrated in Figure 2. The measured received reverb signal Rx (f) and the theoretical model H (d, bi, b) are represented.₂, f) which were obtained by finding the parameters d, b₁, b₂ of the model that best matches the data. Specifically, the observed received reverb signal (dotted line) Rx (f) is plotted against the theoretical channel model (solid line) H (d, b₁, b₂, f) as a function of frequency for an exemplary 6000 ft (about 1830 m) loop with a single exemplary 1300 ft (about 397 m) bridged tap. The exemplary loop consisted of a 6,000-foot (about 1830-foot) 26-awg wire. (American Wire Gauge) with a bridged tap of 1300 feet (about 397 m) and 26 awg. next to the CPE. The model parameters that best matched the observed data were found to be d = 6000 ft (about 1830 m), b₁ = 1300 feet (about 397 m) and b₂ = 0 feet (0 m).

It follows from equation 5 that the interpretation algorithm basically performs a search on the variables d, b1 and b2 and finds those that minimize the cost function given below:

or

E (d, bi, b₂) = Σ | H (d, bi, b2, fi) - Rx (fi) | 2 _ i = if ⁽⁶⁾

Since the cost function E (d, b1, b2) is a nonlinear function of d, b1, and b2, the function contains many local minima. Therefore, many well-known optimization algorithms should not be used, such as the Gauss-Newton one, since these algorithms are unable to handle multiple local minima and converge on a local minimum of the cost function. In this exemplary embodiment the global minimum of E (d, b1, b2) is desired. For this reason, a global brute force minimization algorithm is used where the cost function is sampled at the points (d^p, bi, b2), d^p = pAD, bq = qAb₁ and b2 = rAb₂ with p = 1, ..., P, q = 1, ..., Q and r = 1, ..., R. Then the parameters (d^p, bq, b2) that result in the minimum cost among the sampled values. This requires evaluating the cost function at P x Q x R locations.

In order to be able to determine the theoretical transfer function of the loop, H (d, b₁, b₂, f), it is necessary to store the frequency-dependent propagation constant -y (f) for a number of wires of different gauges. In an exemplary embodiment, 24 awg wires are used. and 26 awg. requiring 4 x N locations to store the real and imaginary parts of γ (Γ) for N ADSL tones. Additionally, it is necessary to store the analog front end (AFE) compensation curves, which occupy N locations in memory. Depending on where the algorithm is implemented, the loop transfer function can be determined directly from equation 4, for example, if the algorithm was implemented on a personal computer or workstation, or it may be necessary to store the terms log [ 2 + Ightb, γ) | at regular intervals as required by the sampling procedure for (d^p, bq, b2). For example, it is possible to precompute and store the log [2 + tgh (bry)], i = 1, 2, from b₁ = 100 feet (about 30 m) ab₁ = 2000 ft (about 610 m) at 100 ft (about 30 m) intervals. Assuming low processor power, the terms log [2 + Ightb, γ) | they can be predetermined and stored, which occupies about 20 x N locations for the real part only. Thus, in this exemplary embodiment, the total memory is approximately (20 + 4 + 1 + 3) x N = 28 x N, where 2 x 256 locations are needed to store intermediate variables determined during algorithm execution.

Although it will not be shown here, it is possible to simplify the computation of the cost function E (d, b₁, b₂) so that only twelve multiplications and fifteen additions are needed. This means that the total computational complexity of the algorithm is approximately P xQ xR x (11 multiplications +15 sums) plus some additional boot computations that are negligible compared to the previous figure.

Unlike the case of downstream interpretation, for upstream interpretation it is more accurate to assume that the line is not perfectly finished. Specifically, the impedance mismatch in the transmitter-line connection in the CPE modem and the impedance mismatch in the receiver-line connection in the CO modem become important factors to be considered. Although the basic idea behind the channel characterization algorithm for the upstream data remains the same and involves matching a theoretical channel transfer function with the actual measured transfer function, the calculation of the theoretical channel transfer function arrives to be more much more complicated. As with the downstream interpretation case, the channel transfer function is again measured by averaging K frames of the received reverb signal as given by equation 1.

The theoretical model for the channel transfer function in the upstream case can be described in two stages. The first stage consists of writing the equations for the current and voltage at the source (CPE), I_s, V_s, in terms of current and voltage at the load (CO), I_L, V_L, by applying the ABCD matrices:

<img file="ES2265411T3_D0003.tif" />

where toⁱ, B, F^s and F^L are 2x2 matrices whose elements are groupings of N elements. Here, Aⁱ is a matrix representing the frequency domain response of the i-th section of the loop, B is the matrix representing the

ES 2 265 411 T3 bridged tap response and F^s and F^L are the matrices representing the analog front end (AFE) hardware frequency domain response of the modem circuitry for the TX (source) and RX (load) paths. The channel transfer function can be derived from equation 7 and is given by:

V_L

H (d1, d2, b, F) = -t, ^VS (8) where d1 is the length of the section before a bridged tap and d2 is the length of the section after the bridged tap. Note that the CO interpretation algorithm uses a two-section single bridged tap model. This is due to the limited number of direct frequency elements, fi = iAf, from pitch i = 6 to i = 32, in which the transfer function is available.

The inputs of the above matrices are given as follows:

A11 = A22 = cosh (ydi)

AZ senh (yd.), Ai_!1 = A^j12Z2

Matrix B inputs:

B11B22 = ¹

<td></td><td>^B12</td><td> = 0,</td><td>^B21</td><td> = ^Z-¹(^b)</td><td></td><td></td>

<td>Where Zj¹ = tgh (by) / Z₀ and finally:</td><td></td><td></td><td></td><td></td><td></td><td></td>

<td></td><td>_FS eleven</td><td>= F^S = ⁼ 22 ⁼</td><td> 1,</td><td>F1^S2 = 0,</td><td>F^S = ₂₁⁼</td><td>ZS</td>

<td></td><td>_FL eleven</td><td>= F^L = ⁼ 22 ⁼</td><td> 1,</td><td>F1^L2 = 0,</td><td>F^L = ₂₁⁼</td><td> /</td>

The estimation algorithm minimizes the difference between the measured and real transfer functions:

rmn \\ H ©, d₂, b, f) -Rx (ff₂ (9) d_td, b ^¿

An example of the operation of the upstream estimation algorithm for loop length and bridged tap length is illustrated in Figure 3. Here, the measured received reverb signal Rx (f) and the theoretical model H (d, b1, b2, f) that was obtained by finding the model parameters d, b1, b2 that best match the data are displayed. The exemplary loop consisted of a 7,700 ft (about 2,348 m) 26 awg wire. with a 600 ft (about 183 m) and 26 awg bridged tap at a distance of 5900 ft (about 1799 m) from the CO. The model parameters that best matched the observed data were found to be d1 = 7,900 feet (about 2409 m), d2 = 0 feet (0 m), and b = 500 feet (about 152 m). Note that although the parameters d1 and d2 found by the algorithm are different from their real values, the real values ​​are d₁ = 5900 feet (about 1799 m) and d₂ = 1800 feet (about 549 m), and the sum of d₁ + d₂it is within 200 feet (about 61 m) of the actual loop length. This example illustrates that although the loop length is fairly accurate, the location of the bridged tap is difficult to estimate reliably.

From the expressions leading to the theoretical channel transfer function, H (d₁, d₂, b, f), it is obvious that for the exemplary computation of the theoretical channel response Z need to be stored_S, Z_L, Z₀ and γ, for 24 awg. and 26 awg., and that Z¡ (b₁), which characterizes the bridged tap, depends on the length of the bridged tap. Assuming an exemplary resolution of 100 feet (about 30 m) at the bridged tap length and an exemplary maximum detectable bridged tap length of 2000 feet (about 610 m), there are 20 different clusters Z_j(b₁). Finally, the elements of sinh (.) And cosh (.) Of the matrices A are stored₁ already₂. Next, assuming a resolution of 500 feet (about 152 m) in the loop length and a maximum measurable loop length of 20,000 feet (about 6100 m), there will be 80 x 46 locations to store A inputs_i. In total, to store these variables, there will be 108 x 46 memory locations, including the storage of Rx (f) and H (d₁, b₁, b₂, f), and another 10 x 46 locations are needed to store intermediate variables during algorithm execution, giving a total of approximately 118 x 46 memory locations for this exemplary embodiment.

Figure 3 illustrates the observed received reverb signal (dashed line) Rx (f) plotted against the theoretical channel model (solid line) H (d1, d2, f) as a function of frequency for an exemplary 7700 ft loop ( about 2348 m) with a single 600 ft (about 183 m) bridged intake.

During the search process, P values ​​are selected for d₁, Q values ​​for b and R values ​​for d₂ and the cost function is determined for each combination of d₁, d₂, b. Thus, to determine the channel impulse response, there are 4 x (8 x 23 complex multiplications + 4 x 26 complex sums). Thus, the total computing cost in this exemplary embodiment is P x Q x R x (32 x 26 complex multiplications + 4 x 26 complex additions).

ES 2 265 411 T3

Figure 4 illustrates an exemplary loop length and bridged tap length estimation system in accordance with one embodiment of this invention for downstream data. In particular, the loop length and bridged tap length estimation system 100 comprises a downstream loop length and bridged tap length determination device 200, a loop length and bridge length determination device 300. Upstream bridged tap, a central office modem 20 and a user facility modem 30, connected by a link 10, such as a twisted pair. The downstream loop length and bridged tap length determining device 200 comprises a controller 210, an I / O interface 220, a storage device 230, a reverberation signal determining device 240, a signal output device loop length 250 and a bridged tap output device 260, connected by link 5. The loop length and upstream bridged tap length determining device 300 comprises a controller 310, an I / O interface 320, a storage device 330, a reverberation signal determining device 340, a impedance 350, a modem identification device 360, a loop-length output device 370, and a bridged tap output device 380, connected by link 5.

Although the exemplary embodiment illustrated in FIG. 4 shows the components of the loop length and bridged tap length estimation system and associated components positioned, it should be appreciated that the various components of the loop length estimation system 100 and of the bridged tap length can be located in distant portions of a distributed network, such as a local area network, a wide area network, an Intranet and / or the Internet, or within a dedicated loop length and bridged tap length estimation system. Thus, it should be appreciated that the components of the loop length and bridged tap length estimation system 100 may be combined into one device or placed at a particular node in a distributed network. As will be appreciated from the following description, and for computational efficiency reasons, the components of the loop length and bridged tap length estimation system 100 may be arranged at any location, such as on a general purpose computer or within of a distributed network, without affecting the operation of the system.

Furthermore, the links 5 may be wired or wireless links or any other known or subsequently developed element or elements that are capable of supplying electronic data to and from the connected elements.

In operation, for the determination of the loop length and the bridged tap length in the downstream direction, the controller 210, in cooperation with the I / O interface 220, causes the initialization of the modem 20. The determination device 240 From the reverb signal, in cooperation with modem 20, controller 210, and I / O interface 220, determines a transfer function by averaging K consecutive frames of a reverb signal. The loop length, a first bridged tap length, and a second bridged tap length are input from an input device (not shown), such as a computer, laptop, terminal, transmission line test device. or similar, or are retrieved from storage device 230.

Controller 210, in cooperation with storage device 230, then determines the frequency domain propagation function for a specified wire gauge and frequency domain loop model. The calibrated and compensated reverb signals in the frequency domain are stored in the storage device 230 and the reference wire gauge is entered into or retrieved from the storage device 230.

Controller 210, in cooperation with storage device 230, determines the number of elements in the Rx function and the difference between the actual and measured transfer functions. The loop length output device, in cooperation with the I / O interface, then outputs the estimated loop length to, for example, a computer, laptop, terminal, transmission line testing device or similar. Additionally, the bridged tap output device outputs the estimated length of the bridged tap to, for example, a computer, laptop, terminal, transmission line testing device or the like.

In operation, to determine the loop length and the length of the bridged tap in the upstream direction, the controller 310, in cooperation with the I / O interface 320, triggers the initialization of the modem 30. The modem determining device 340 Reverb signal, in cooperation with modem 30, controller 310, and I / O interface 320, determines a transfer function by averaging K consecutive frames of a reverb signal.

Next, controller 310, in cooperation with storage device 230, determines the frequency domain propagation function for a specified wire gauge, where the specified wire gauge is input or retrieved from storage device 330.

Controller 310, in cooperation with storage device 330 and impedance determining device 350, determines the frequency domain impedance of the specified wire gauge. Next, controller 310, in cooperation with storage device 330 and impedance determining device 350, determines the transmit impedance of the CPE modem and the receive impedance of the CO modem.

ES 2 265 411 T3

Controller 310, in cooperation with storage device 330, determines the matrix representing the frequency domain responses of the i-th section of the loop, the matrix representing the bridged tap response, and the matrix F^S which represents the AFE circuitry for the source (TX) and load (RX) paths and stores them in the storage device 330, and estimates the transfer function H. The calibrated and compensated reverb signals in the domain The frequency and reference gauge of the wire are entered into or retrieved from the storage device 330.

The modem identification determining device 360 ​​then determines the identification of the modem of the CO that picks up the upstream reverb signal, and the identification of the modem of the CPE that transmits the upstream reverb signal. Knowing the number of elements in the Rx function, the controller 310 minimizes the difference between the actual and measured transfer functions and outputs, with the cooperation of the loop length output device 370 and the bridged tap output device 380, the estimated loop length and the estimated length of the bridged tap, respectively.

Figure 5 illustrates an exemplary method of determining a loop length and bridged tap lengths. In particular, control begins at step S100 and continues to step S110. In step S110, the channel impulse response is estimated based on a measured reverb signal. Next, in step S120, the theoretical channel impulse response of a loop model is determined using a loop length and the lengths of the bridged tap. Next, in step S130, the loop length and the bridged tap lengths of the model are modified. Control then continues to step S140.

In step S140, the difference between the measured channel impulse response and the theoretical channel impulse is monitored. Next, in step S510, the estimated values ​​of the loop length and the bridged tap length are declared based on the loop lengths and the bridged tap lengths that minimize the error function between the measured channel impulse response and the theoretical channel impulse response. Control then continues to step S160, where the control sequence ends.

Figure 6 illustrates an exemplary method of determining loop length and bridged tap length for downstream data. In particular, control begins at step S200 and continues to step S210. In step S210, a modem is initialized. Next, in step S220, a transfer function is determined by averaging K consecutive frames of the reverb signal. Then, in step S230, the loop length is entered. The control then continues in step S240.

In step S240, a first length of the bridged tap is input. Then, in step S250, a second length of the bridged tap is input. Next, in step S260, the frequency domain propagation function is determined for a specified yarn gauge. Control then continues in step S270.

In step S270, the frequency domain loop pattern is determined. Next, in step S280, the calibrated and compensated reverb signals are input in the frequency domain. Then, in step S290, the reference wire gauge is input. Control then continues in step S300.

In step S300, the number of elements is entered in the Rx function. Next, in step S310, the difference between the actual and the measured transfer function is determined. Then, in step S320, the estimated length of the loop is determined. Control then continues to step S330.

In step S330, the estimated length of the bridged tap is determined. Control then continues to step S340, where the control sequence ends.

Figure 7 illustrates an exemplary method of determining loop length and bridged tap length for upstream data. In particular, control begins at step S500 and continues at step S510. In step S510, the modem is initialized. Then, in step S520, the transfer function is determined by averaging K consecutive frames of the reverb signal. Then, in step S530, the frequency domain propagation function is determined for the yarn gauge in use. Control then continues to step S540.

In step S540, the wire gauge frequency domain impedance is determined. Next, in step S550, the transmission impedance of the CPE modem is determined. Then, in step S560, the receive impedance of the CO modem is determined. Control then continues at step S570.

In step S570, the matrix representing the frequency domain responses of the ith section of the loop is determined. Next, in step S580, the matrix representing the bridged tap response is determined. Then, in step S590, the matrix F is determined^S which represents the AFE circuitry for the source (TX) and load (RX) paths. Control then continues in step S600.

In step S600, the transfer function H is estimated. Next, in step S610, the calibrated reverb signal and the compensated one are input in the frequency domain. Then, in step S620, the yarn reference gauge is entered. Control then continues to step S630.

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In step S630, the ID of the modem of the CO collecting the upstream reverb signal is entered. Then, in step S640, the modem identification of the CPE transmitting the upstream reverb is entered. Next, in step S650, the number of elements in the Rx function is input. Control then continues to step S660.

In step S660, the difference between the actual and measured transfer functions is minimized. Then, in step S670, the estimated loop length is determined. Next, in step S680, the estimated length of the bridged tap is determined. Control then continues to step S690, where the control sequence ends.

As illustrated in Figure 4, the loop length and bridged tap length estimation system can be implemented in a general purpose computer with a single program, or in a general purpose computer with separate programs. However, the loop length and bridged tap length estimation system can also be implemented in a special purpose computer, a programmed microprocessor or microcontroller or a peripheral integrated circuit element, an ASIC (Application-Specific Integrated Circuit - application-specific integrated circuit) or other integrated circuit, a digital signal processor, a hard-wired logic or electronic circuit, such as a discrete element circuit, a programmable logic device, such as a PLD (Programmable Logic Device), PLA (Programmable Logic Array Array), FPGA (Field-Programmable Gate Array - field gate array programmable), PAL (Phase Alternating Line), a modem or similar. In general, any device capable of implementing a finite state machine that in turn is capable of implementing the flow diagrams illustrated in Figures 5-7 can be used to implement the loop length and length estimation system. bridged tap according to this invention.

Furthermore, the described method can be easily implemented in software using object software or object-oriented software development environments that provide portable source code that can be used on a variety of computer or workstation hardware platforms. Alternatively, the loop length and bridged tap length estimation system can be partially or completely implemented in hardware using standard logic circuits or VLSI (Very Large Scale Integration) design. Whether software or hardware is used to implement the systems in accordance with this invention depends on the speed and / or efficiency requirements of the system, the particular function, and the particular software or hardware systems or microprocessor or microcomputer systems being used. using. However, the systems and methods for estimating the loop length and the bridged tap length illustrated here can be easily implemented in hardware and / or software, using any known or subsequently developed systems or structures, devices and / or software, by those of ordinary skill in the applicable art based on the functional description provided herein and a general basic knowledge of computer techniques.

Furthermore, the described methods can be easily implemented as run in software on a general-purpose programmed computer, a special-purpose computer, a microprocessor, or the like. In these cases, the methods and systems of this invention can be implemented as a program installed on a personal computer, such as a Java language.^® o CGI (Common Gateway Interface), as a resource that resides on a graphics server or workstation, as a routine installed on a dedicated loop length and bridged tap length estimation system, a modem, a dedicated loop length and / or bridged tap length estimation system, or the like. Also, the loop length and bridged tap length estimation system can be implemented by physically incorporating the system and the method into a software and / or hardware system, such as the hardware and software systems of a system or modem. dedicated to estimate the loop length and the bridged tap length.

Thus, it is clear that, in accordance with the present invention, systems and methods have been provided for estimating loop length and bridged tap length. Although this invention has been described in conjunction with a number of embodiments thereof, it is apparent that many alternatives, modifications, and variations would be or are apparent to those of ordinary skill in the applicable arts. Accordingly, it is intended to encompass all alternatives, modifications, equivalents, and variations that are within the scope of the appended claims.