Fractal antennas, resonators and loading elements

ES 2 236 745 T3

DESCRIPTION

Antenna, resonators and fractal charge elements.

Field of the invention

The present invention relates to antennas and resonators and, specifically, to the design and tuning of ground radials of non-Euclidean antennas, counterweights or ground planes, top loading elements, and antennas employing such elements.

Background of the invention

Antennas are used to radiate and / or receive, typically, electromagnetic signals, preferably with antenna gain, steedability, and efficiency. Practical antenna design traditionally involves trade-offs between various parameters, including antenna gain, size, efficiency, and bandwidth.

Antenna design has historically been dominated by Euclidean geometry. In such designs, the closed area of ​​the antenna is directly proportional to the perimeter of the antenna. For example, if the length of a Euclidean square (or “quad”) antenna is doubled, the bounded area of ​​the antenna is quadrupled. Classic antenna design has dealt with planes, circles, triangles, squares, ellipses, rectangles, hemispheres, parabolas, and the like (as well as lines). Similarly, resonators, typically capacitors ("C") coupled in series and / or in parallel with inductors ("L"), are traditionally implemented with Euclidean inductors.

With respect to antennas, the prior art design philosophy has been to choose a Euclidean geometric construction, e.g. eg, a “quad”, and explore its irradiation characteristics, with special emphasis on frequency resonance and energy patterns. The unfortunate result is that antenna design has for too long focused on the ease of antenna construction, rather than the underlying electromagnetism.

Many antennas of previous technology are based on forms of closed loop or of island. Experience has long shown that small size antennas, including loops, do not perform well, one reason being that radiation resistance ("R") decreases rapidly when the size of the antenna is shortened. A small loop, or even a short dipole, will exhibit a radiation pattern of 1 / 2λ and 1 / 4λ, respectively, if the resistance to radiation R is not overwhelmed by substantially larger ohmic losses ("O"). Ohmic losses can be minimized by using impedance matching networks, which can be expensive and difficult to use. But while even impedance-matched, small-loop antennas can exhibit efficiencies of 50% to 85%, their bandwidth is inherently narrow, with a very high Q, e.g. eg, Q> 50. As used here, Q is defined as (transmitted or received frequency) / (3 dB bandwidth).

As has been observed, it is well known experimentally that the resistance to radiation R drops rapidly with small area Euclidean antennas. However, the theoretical basis is not generally known, and any current understanding (or misunderstanding) seems to originate from the research of J. Kraus, mentioned in Antennas (Ed. 1), McGraw Hill, New York (1950), where examined a circular loop antenna with uniform current. The Kraus loop exhibited a gain with a surprising limit of 1.8 dB over an isotropic irradiator, as the area of ​​the loop fell below that of a loop having an aperture of 1 λ squared. For small loops of area A <λ²/ 100, the resistance to radiation R was given by:

R = K (A / λ²)² where K is a constant, A is the bounded area of ​​the loop, and λ is the wavelength. Unfortunately, the resistance to radiation R can all too easily be less than 1 Ω for a small loop antenna.

From his investigation of the circular loop, Kraus generalized that the calculations could be defined by antenna area, rather than antenna perimeter, and that his analysis should be correct for small loops of any geometric shape. Kraus's early research and conclusions, that small-sized antennas will exhibit a relatively large ohmic resistance O and a relatively small resistance to radiation R, such that the resulting low efficiency detracts from the use of the small antenna, have been widely accepted. Indeed, some researchers have actually proposed reducing the ohmic resistance O to 0 Ω, by building small antennas of superconducting material, in order to promote efficiency.

As noted, prior art antenna and resonator design has traditionally focused on geometry that is Euclidean. However, a non-Euclidean geometry is fractal geometry. Fractal geometry can be grouped into random fractals, which are also called chaotic or Brownian fractals, and include random noise components, such as that illustrated in Figure 3, or deterministic fractals such as those shown in Figure 1C.

In deterministic fractal geometry, a structure similar to itself is the result of the repetition of a design or motif (or "generator"), at a number of different size scales. A well-known treatise in this field

ES 2 236 745 T3 en Fractals, Endlessly Repeated Geometrical Figures, by Hans Lauwerier, Princeton University Press (1991).

Figures 1A-2D illustrate the development of some elementary fractal shapes. In Figure 1A, a base member 10 is shown as a straight line, although a curve could be used instead. In Figure 1B, a so-called Koch fractal, or generator 20-1 motif, a triangle in this case, is inserted into the base element 10, to form a first-order ("N") iteration pattern, e.g. eg, N = 1. In Figure 1C a second order iteration design, N = 2, is obtained by replicating the 20-1 triangular motif in each segment of Figure 1B, but where the 20-1 'version has been scaled different, here reduced in size. As observed in Lauwerier's treatise, in its replication, the motif can be subjected to rotation, translation, change of scale in its dimension, or a combination of any of these characteristics. Thus, as used here, the second order of iteration, or N = 2, means that the ground motif has been replicated, after rotation, translation, scaling (or a combination of these), in the pattern first-order iteration. A higher order iteration, p. eg, N = 3, means that a third fractal pattern has been generated, including another rotation, translation and / or scaling of the first order motif.

In Figure 1D, a portion of Figure 1C has been subjected to an additional iteration (N = 3), in which reduced-scale versions of the 20-1 triangular motif have been inserted into each segment of the left half of Figure 1 C. Figures 2A-2C continue what has been described with respect to Figures 1A-1C, except that a rectangular pattern 20-2 has been adopted. Figure 2D shows a pattern in which a portion of the left side is an N = 3 iteration of the rectangular motif 20-2, and in which the central portion of the figure now includes another motif, in this case, a triangular motif of type 20-1, and in which the right side of the figure remains in the iteration N = 2.

Traditionally, non-Euclidean designs that include random fractals have been understood to exhibit antiresonance characteristics with mechanical vibrations. It is known in the art to attempt to employ non-Euclidean random designs at lower frequency regimes in order to absorb or at least not reflect sound due to antiresonance characteristics. For example, M. Schroeder, in Fractals, Chaos, Power Laws (1992), WH Freeman, New York, reveals the use of supposedly random or chaotic fractals when designing sound-blocking diffusers for recording studios and auditoriums.

Experimentation with non-Euclidean structures has also been undertaken with respect to electromagnetic waves, including radio antennas. In one experiment, Y. Kim and D. Jaggard, in The Fractal Random Array, Proc. IEEE 74, 1278-1280 (1986), spread antenna elements in a sparse microwave array, in order to minimize side lobe energy without having to employ an excessive number of elements. But Kim and Jaggard did not apply a fractal condition to the antenna elements, and the test results were not necessarily better than any other techniques, including a totally random scattering of antenna elements. Most significantly, the resulting formation was no smaller than a conventional Euclidean design.

Spiral antennas, cone antennas, and V-shaped antennas of the prior art can be thought of as a continuous, deterministic, first-order fractal, the pattern of which continually expands with increasing distance from a central point. A logarithmic periodic antenna can be considered a type of continuous fractal in that it is manufactured from a radially expanding structure. However, logarithmic periodic antennas do not use the perimeter of the antenna for irradiation, but instead use an arc-shaped opening angle in the antenna geometry. Such opening angle is an angle that defines the scale of the size of the logarithmic periodic structure, which structure is proportional to the distance from the center of the antenna, multiplied by the opening angle. Furthermore, known logarithmic periodic antennas are not necessarily smaller than conventional controlled element-parasitic element antenna designs of similar gain.

First order fractals have been used, unintentionally, to distort the shape of dipole and vertical antennas in order to increase gain, the shapes being defined as a Brownian type of chaotic fractals. See F. Landstorfer and R. Sacher, Optimization of Wire Antennas, J. Wiley, New York (1985). Figure 3 illustrates three inclined-vertical antennas developed by Landstorfer and Sacher with the trial and error method, the graphs showing the actual vertical antennas as a function of x-axis and y-axis coordinates, which are a function wavelength. The nomenclature of "DE" and "DP" in Figure 3 refer, respectively, to the extreme shot and back shot irradiation patterns of the resulting tilt-vertical antennas.

First-order fractals have also been used to reduce the geometry of horn-type antennas, in which a double-peak horn configuration is employed to lower the resonant frequency. See J. Kraus in Antennas, McGraw Hill, New York (1985). The use of rectangle, square, and triangle shapes as impedance matching load elements, in order to shorten the dimensions of antenna elements, is also known in the art.

Whether intentional or not, such prior art attempts to use a quasifractal or fractal pattern in an antenna employ, at best, a first order iteration fractal. By first iteration it is meant that a Euclidean structure is loaded with another Euclidean structure in a repetitive way, using the same size

ES 2 236 745 T3 for repetition. Figure 1C, for example, is not first-order, because the triangles 20-1 'have shrunk relative to the size of the first pattern 20-1.

The prior art antenna design does not attempt to exploit the multi-scale self-similarity of true fractals. This is very unsurprising, in view of accepted conventional wisdom, in that, because such antennas would be anti-resonance, and / or, if conveniently shrunk, would exhibit such little resistance to radiation R, the ohmic losses Or substantially higher would result in too low antenna efficiency for any practical use. Furthermore, it is probably not possible to mathematically predict such an antenna design, and high-order iteration fractal antennas would be increasingly difficult to manufacture and erect, in practice.

Figures 4A and 4B respectively illustrate configurations of resonators, of series and parallel types, of the prior art, comprising capacitors C and Euclidean inductors L. In the series configuration of Figure 4A, a filter characteristic is presented -notch in that the impedance from port A to port B is high, except for frequencies approaching resonance, determined by 1 / γ (LC).

In the distributed parallel configuration of Figure 4B, a low pass filter characteristic is created in that, for frequencies below resonance, there is a relatively low impedance path from port A to port B, but for frequencies higher than the resonant frequency, signals at port A are shunted to ground (eg, common terminals of capacitors C), and a high impedance path is presented between port A and port B. Of course, a unique parallel LC configuration can also be created by removing (e.g., shorting) the rightmost inductor L and two right-hand capacitors C, in which case port B would be at the lower end of capacitor C of more to the left.

In Figures 4A and 4B, the inductors L are Euclidean, in that the increase in the effective area captured by the inductors increases with increasing geometry of the inductors, e.g. eg more numerous or larger inductive windings or, if they are not cylindrical, traces that include inductance. In prior art configurations such as Figures 4A and 4B, the presence of Euclidean inducers L guarantees a predictable relationship between L, C and the resonance frequencies.

Applicant's patent application FRACTAL ANTENNA AND FRACTAL RESONATORS, mentioned above, provided a design methodology to produce smaller-scale antennas that exhibit at least as much gain, steedability, and efficiency as more Euclidean counterparts. large. Such a design approach should exploit the multiple-scale self-similarity of true fractals, including fractals of iteration order N> 2. Furthermore, said application disclosed a non-Euclidean resonator whose presence in a resonant configuration can create resonance frequencies beyond those normally presented in serial and / or parallel LC configurations. Applicant's patent application TUNING FRACTAL ANTENNAS AND FRACTAL RESONATORS, mentioned above, provided devices and procedures for tuning and / or adjusting such antennas and resonators. Said application further revealed the use of non-Euclidean resonators, the presence of which in a resonant configuration could create resonance frequencies beyond those normally presented in series and / or parallel LC configurations.

However, such antenna design and tuning approaches should also be usable with vertical antennas, allowing downscaling of one or more radial ground plane elements, and / or ground planes, and / or ground counterweights. , and / or elements of load of top hat.

The present invention provides such antennas, radial ground plane elements, ground planes, ground counterweights and top hat loading elements, as well as methods for their design.

Summary of the invention

Patent document US-A-3079602 describes a fractal antenna system in which a first pattern, defined in at least two dimensions and in the shape of a tooth or a triangle, is replicated to scale. As is known from patent document US-A-3079602, the present invention provides an antenna system comprising:

an antenna arrangement with at least one part that is fractal in design and that includes a first element having a portion that includes at least one first motif defined in at least two dimensions, said portion further including at least one first replication of said first motif and a second replication of said first motif, in such a way that a point chosen on a geometric figure represented by said first motif results in a corresponding point on said first replication and on said second replication of said first motif, each in different spatial locations, and a conductive element, away from said fractal antenna, to influence at least one of the resonant frequency and the bandwidth of said antenna system.

In contrast to patent document US-A-3079602, and according to the invention, each of the replications is far from the first motif and geometrically defined by at least one set of operations, selected from a group consisting of (a) change scaling the size of said first pattern, (b) rotating said first pattern and (c) translating said first pattern; Y

Each operation that defines each replication excludes those operations that are a function of, and that allow reference to, the spatial location of a single point on said first motif.

The present invention also provides a method of making an antenna system, including an antenna arrangement comprising:

effecting the antenna arrangement to include a fractal antenna, the fractal antenna being arranged to include a first element having a portion that includes at least one first pattern defined in at least two dimensions, at least one first replication of said first pattern and a second replication of said first motif, such that a point chosen on a geometric figure represented by said first motif results in a corresponding point on said first replication and on said second replication of said first motif, each at different spatial locations; and coupling a conductive element, away from said antenna arrangement, to influence at least one of the resonant frequency and the bandwidth of said antenna system characterized in that each of the replications is away from the first motif and geometrically defined by al minus one set of operations selected from a group consisting of (a) downscaling the size of said first motif, (b) the rotation of said first motif and (c) the translation of said first motif, and each operation that defines each replication excludes those operations that are a function of, and admit reference to, the spatial location of a single point on said first reason.

As appears from the following, an antenna system constructed in accordance with the invention may have a ground plane or ground counterweight system, having at least one element whose shape, at least in part, is essentially a deterministic fractal of iteration of order N> 1. (The term "ground counterweight" will be understood to include a ground plane, and / or at least one ground element). Employing fractal geometry, the ground counterweight of the antenna has a self-similar structure that results from the repetition of a pattern or motif (or "generator") that is replicated using rotation, and / or translation, and / or reduction. to scale. The fractal element will have coordinates of the x axis and the y axis for a next iteration N + 1, defined by x_N+₁ = f (x_N, and b_N) yy_{N +}i = g (x_N,Y_N), where x_N, Y_Ndefine the coordinates for a preceding iteration, and where f (x, y) and g (x, y) are functions that define the fractal motif and behavior. A vertical antenna can be cup loaded with a so-called top-hat assembly, which includes at least one fractal element. A fractalized top hat assembly conveniently reduces the resonant frequency as well as the physical size and area required for the top hat assembly.

In contrast to an antenna design of Euclidean geometry, deterministic fractal elements according to the present invention have a perimeter that is not directly proportional to area. For a given perimeter dimension, the bounded area of ​​a multi-iteration fractal will always be as small as, or smaller than, the area of ​​a corresponding conventional Euclidean element.

A fractal antenna has a dimension D of the fractal ratio limit, given by log (L) / log (r), where L and r are one-dimensional lengths of antenna elements, before and after fractalization, respectively.

As used here, a Perimeter Compression (CP) parameter of a fractal antenna is defined as:

CP _ Total length of the antenna element

Reduced length per fractal of the antenna element where:

CP = A log [N (D + C)] in which A and C are constant coefficients for a given fractal pattern, N is an iteration number and D is the fractal dimension, defined above.

The radiation resistance R of a fractal antenna decreases as a small perimeter compression (CP) power, with a fractal loop or island always exhibiting significantly greater radiation resistance than a small Euclidean loop antenna of the same size. Deterministic fractals are used in the present invention in which A and C have large values ​​and thus provide the largest and fastest shrinkage of element size. A fractal antenna according to the present invention will exhibit an increased effective wavelength.

The number of resonant nodes in a loop-shaped fractal antenna increases with the iteration number N and is at least as large as the number of resonant nodes in a Euclidean island with equal area. Also, the resonant frequencies of a fractal antenna include frequencies that are not harmonically linked.

An antenna that includes a fractal ground counterweight according to the present invention is smaller than its Euclidean counterpart, but provides at least as much gain and as many resonance frequencies, and provides a

ES 2 236 745 T3 reasonable termination impedance at its lowest resonant frequency. Such an antenna system can exhibit non-harmonically resonant frequencies, a low Q and good bandwidth as a result, an acceptable Standing Wave Ratio (SWR) and a radiation impedance that is frequency dependent, and high efficiency values.

With respect to vertical antennas, the present invention allows such antennas to be made with a smaller vertical element, and / or with smaller ground counterweights, e.g. eg, radial ground plane elements and / or a ground plane. The ground counterweight element (s) are fractalized with N> 1. In a preferred embodiment, the vertical element is also a fractal system, preferably comprising spaced first and second fractal elements.

A fractal antenna system having a fractal ground counterweight and a fractal vertical element is tuned, preferably, by placing an active (or controlled) fractal antenna or resonator at a distance Λ from a second conductor. Such an arrangement of the antenna and second conductor conveniently lowers the resonant frequencies and widens the bandwidth for the fractal antenna. In some embodiments, the fractal antenna and the second conductor are non-coplanar and λ is the separation distance between them, preferably <0.05 λ for the frequency of interest (1 / λ). In other embodiments, the fractal antenna and the second conductive element may be coplanar, in which case λ is a separation distance, measured on the common plane. In another embodiment, an antenna is loaded with a fractal "top hat" assembly, which can provide a substantial reduction in antenna size.

The second conductor may, in fact, be a second fractal antenna of similar or different configuration to that of the active antenna. Varying the distance Λ fine-tunes the active antenna and therefore the overall system. Furthermore, if the second element, preferably a fractal antenna, is rotated angularly relative to the active antenna, the resonant frequencies of the active antenna can be varied.

Cutting the fractal antenna results in new and distinct resonant nodes, including resonant nodes that have perimeter compression parameters, defined below, ranging from about three to ten. If desired, a portion of a fractal antenna can be cut and removed to fine-tune the antenna, increasing the resonance (s).

Tunable antenna systems with a fractal ground counterweight need not be coplanar, according to the present invention. Fabrication of the antenna system around a shape such as a toroidal ring, or formation of the fractal antenna on a flexible substrate that is curved back on itself results in field self-proximity that produces resonant frequency offsets. A fractal antenna and a conductive element can be formed as a curved surface, or even a toroidal shape, and placed in close enough proximity to each other to provide a useful tuning and feature-altering mechanism for the system.

In the various embodiments, more than two elements can be used, and tuning can be accomplished by varying one or more of the parameters associated with one or more elements.

Other characteristics and advantages of the invention will emerge from the following description, in which the preferred embodiments have been pointed out in detail, in conjunction with the accompanying drawings.

Brief description of the drawings

Figure 1A illustrates a base element for an antenna or an inductor, according to the prior technology;

Figure 1B illustrates a triangular shaped Koch fractal motif, according to the prior art;

Figure 1C illustrates a second iteration fractal, employing the pattern of Figure 1B, according to prior technology;

Figure 1D illustrates a third iteration fractal, employing the pattern of Figure 1B, according to prior technology;

Figure 2A illustrates a base element for an antenna or an inductor, according to the prior technology;

Figure 2B illustrates a rectangular shaped Minkowski fractal motif according to prior art;

Figure 2C illustrates a second iteration fractal employing the pattern of Figure 2B, according to prior technology;

Figure 2D illustrates a fractal configuration that includes a third order employing the motif of Figure 2B, as well as the motif of Figure 1B, in accordance with prior technology;

Figure 3 illustrates inclined-vertical chaotic fractal antennas, according to the prior technology;

Figure 4A illustrates a serial LC resonator, according to the prior art;

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Figure 4B illustrates a parallel distributed LC resonator according to the prior art;

Figure 5A illustrates a Euclidean "quad" antenna system, according to prior technology;

Figure 5B illustrates a second order Minkowski island fractal "quad" antenna in accordance with the present invention;

Figure 6 illustrates a free space irradiation pattern generated by ELNEC for an MI-2 fractal antenna, according to the present invention;

Figure 7A illustrates a Cantor comb fractal dipole antenna according to the present invention;

Figure 7B illustrates a broken-frame fractal "quad" antenna in accordance with the present invention;

Figure 7C-1 illustrates a second iteration Minkowski printed circuit (MI-2) fractal antenna, in accordance with the present invention;

Figure 7C-2 illustrates a second iteration Minkowski slot fractal antenna (MI-2) according to the present invention;

Figure 7D illustrates a deterministic dendrite fractal vertical antenna, in accordance with the present invention;

Figure 7D-1A illustrates a vertical 0.25λ antenna with three radial 0.25λ ground elements, based on prior technology;

Figure 7D-1B illustrates the gain pattern for the antenna of Figure 7D-1A;

Figure 7D-2A illustrates a 0.25λ vertical antenna with three fractal radial ground elements in accordance with the present invention;

Figure 7D-2B illustrates the gain pattern for the antenna of Figure 7D-2A;

Figure 7D-3A illustrates a "top hat" loaded antenna, according to prior technology;

Figure 7D-3B illustrates the gain pattern for the antenna of Figure 7D-3A;

Figure 7D-4A illustrates a ternary fractal "top hat" loaded antenna, in accordance with the present invention;

Figure 7D-4B illustrates the gain pattern for the antenna of Figure 7D-4A;

Figure 7D-5 illustrates an antenna having a fractal vertical element and fractal radial ground elements, in accordance with the present invention;

Figure 7E illustrates a third iteration Minkowski Island (MI-3) fractal "quad" antenna, in accordance with the present invention;

Figure 7F illustrates a second iteration Koch fractal dipole according to the present invention;

Figure 7G illustrates a third iteration dipole, according to the present invention;

Figure 7H illustrates a second iteration Minkowski fractal dipole according to the present invention;

Figure 7I illustrates a third iteration multifractal dipole according to the present invention;

Figure 8A illustrates a generic system in which an electronic system, passive or active, communicates using a fractal antenna, according to the present invention;

Figure 8B illustrates a communication system in which several fractal antennas, including a vertical antenna with a fractal ground counterweight, are electronically selected for best performance, in accordance with the present invention;

Figure 8C illustrates a communication system in which electronically steerable arrays of fractal antennas are electronically selected for best performance, in accordance with the present invention;

Figure 9A illustrates the fractal antenna gain as a function of iteration order N, in accordance with the present invention;

Figure 9B illustrates perimeter compression CP as a function of iteration order N for fractal antennas, according to the present invention;

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Figure 10A illustrates a fractal inducer for use in a fractal resonator, in accordance with the present invention;

Figure 10B illustrates a credit card-sized security device utilizing a fractal resonator in accordance with the present invention;

Figure 11A illustrates an embodiment in which a fractal antenna is separated by a distance Λ from a conductive element in order to vary the resonant properties and irradiation characteristics of the antenna, in accordance with the present invention;

Figure 11B illustrates an embodiment in which a fractal antenna is coplanar with a ground plane and is separated by a distance Δ 'from a coplanar passive parasitic element, to vary the resonant properties and irradiation characteristics of the antenna, as herein invention;

Figure 12A illustrates the spacing of the first and second fractal antennas at a distance Λ to decrease resonance and create additional resonant frequencies for the active or controlled antenna, in accordance with the present invention;

Figure 12B illustrates the relative angular rotation between the first and second fractal antennas, separated by a distance Δ, to vary the resonant frequencies of the active or controlled antenna, according to the present invention;

Figure 13A illustrates the cutting of a fractal antenna or resonator to create different resonant nodes and to alter perimeter compression, in accordance with the present invention;

Figure 13B illustrates the formation of a non-coplanar fractal antenna or resonator on a flexible substrate that is curved to shift the resonant frequency, apparently due to electromagnetic fields in its own proximity, in accordance with the present invention;

Figure 13C illustrates the formation of a fractal antenna or resonator on a curved toroidal shape to shift the resonant frequency, apparently due to electromagnetic fields of its own proximity, according to the following invention;

Figure 14A illustrates the formation of a fractal antenna or resonator in which the conductive element is not attached to the coaxial of the system or to another power line, according to the present invention;

Figure 14B illustrates a system similar to Figure 14A, but demonstrates that the controlled fractal antenna can be coupled to the system coaxial or other feed line at any point along the antenna, in accordance with the present invention;

Figure 14C illustrates an embodiment in which a supplemental ground plane is provided adjacent a portion of the controlled fractal antenna and the conductive element, forming a sandwich-type system, in accordance with the present invention;

Figure 14D illustrates an embodiment in which a fractal antenna system is tuned by clipping a portion of the controlled antenna, in accordance with the present invention;

Figure 15 illustrates a communication system similar to that of Figure 8A, in which several fractal antennas are tunable and electronically selected for best performance, in accordance with the present invention.

Detailed description of the preferred embodiments

In an overview, the present invention provides an antenna system with a fractal ground counterweight, e.g. eg, a counterweight and / or ground plane and / or ground element having at least one element whose shape, at least in part, is essentially a fractal of iteration order N> 1. The resulting antenna is smaller than its Euclidean counterpart, provides about 50 Ω of termination impedance, exhibits at least as much gain and more resonance frequencies than its Euclidean counterpart, including non-harmonically linked resonance frequencies, exhibits a low Q and the consequently good bandwidth, an acceptable SWR, a radiation impedance that is dependent on frequency and high efficiency values.

In contrast to the Euclidean geometric antenna design, a fractal antenna ground counterweight, according to the present invention, has a perimeter that is not directly proportional to area. For a given perimeter dimension, the bounded area of ​​a multi-iteration fractal area will always be at least as small as any Euclidean area.

Employing fractal geometry, the soil element has a self-similar structure that results from the repetition of a design or motif (or "generator"), which motif is replicated using rotation, translation, and / or downscaling (or any combination). from the same). The fractal portion of the element has the coordinates on the x-axis and on the y-axis, for a next iteration N + 1, defined by x_N+₁ = f (x_N, and b_N) hey_{N +}i = g (x_N,Y_N), where x_N, Y_N are coordinates of a preceding iteration, and where f (x, y) and g (x, y) are functions that define the fractal motif and behavior.

ES 2 236 745 T3

For example, the fractals in the Julia set can be represented in the form:

9 ^xN + 1 = ^xN 'Yn + ^to

Yn + i - ^2xnYn - ^b

In complex notation, the above can be represented as:

^zN + 1 ^{- z}N + ^c

Although it is apparent that fractals can comprise a wide variety of shapes for the functions f (x, y) and g (x, y), it is the iterative nature and the direct relationship between structure or morphology at different size scales that uniquely distinguishes f (x, y) and g (x, h) from non-fractal forms. Many reference works, including Lauwerier's treatise, indicate suitable equations for f (x, y) and g (x, y).

Iteration (N) is defined as the application of a fractal pattern on a scale of sizes. Thus, a single-size scale repeat of a pattern is not a fractal, as that term is used here. Of course, multifractals can be implemented in which a motif is changed for different iterations, but eventually at least one motif is repeated in another iteration.

A general appreciation of the present invention can be obtained by comparing Figures 5A and 5B. Figure 5A shows a conventional Euclidean "quad" antenna 5 having a controlled element 10, each of whose four sides is 0.25 λ long, with a total perimeter of 1 λ, where λ is the frequency of interest.

The Euclidean element 10 has an impedance of about 130 Ω, an impedance that decreases if a parasitic “quad” element 20 is separated on a boom 30 at a distance B of between 0.1 λ and 0.25 λ. The parasitic element 20 is also sized S = 0.25 λ on one side, and its presence can improve the steedability of the resulting two-element "quad" antenna. Item 10 is illustrated in Figure 5A with thicker lines than item 20, solely to avoid confusion in interpreting the figure. Non-conductive extenders 40 are used to help hold element 10 and element 20 together.

Due to the relatively large control impedance, the controlled element 10 is coupled with an impedance matching device or network 60, the output impedance of which is approximately 50 Ω. A coaxial cable 50, typically 50 Ω, couples the device 60 with a transceiver 70 or other active or passive electronic equipment 70.

As used herein, the term "transceiver" shall mean a piece of electronic equipment that can transmit, receive, or transmit and receive an electromagnetic signal via an antenna, such as the "quad" antenna shown in Figure 5A or 5B. As such, the term "transceiver" includes, without limitation, a transmitter, a receiver, a transmitter receiver, a cellular telephone, a cordless telephone, a pager, a communicator of a computer wireless local area network ("LAN"), a unit. passive resonant used by stores as part of an antitheft system, in which the transceiver 70 contains a resonant circuit that is triggered, or is not triggered, by an electronic signal at the time of purchase of the item to which the transceiver 70 is attached, resonant sensors and transponders, and the like.

Furthermore, since the antennas according to the present invention can receive incoming radiation and couple the same as alternating current in a cable, it will be appreciated that fractal antennas can be used to intercept incoming light radiation and to supply a corresponding alternating current. For example, a photocell antenna defining a fractal, or even an array or array of fractals, would be expected to emit more current in response to incoming light than a photocell of the same overall array size. Figure 5B illustrates a fractal "quad" antenna 95, designed to resonate at the same frequency as the larger prior art antenna 5 shown in Figure 5A. The controlled element 100 is seen to be a second-order fractal, in this case a fractal called Minkowski's Island, although any other of numerous fractal configurations could be used instead, including, without limitation, Koch's geometry, that of Broken frame, Mandelbrot, Caley tree, Monkey swing, Sierpinski packing, and Cantor packing.

If the amount of conductive wire or conductive footprint that comprises the perimeter of element 40 were measured, it would be perhaps 40% greater than the 1.0 λ of the Euclidean "quad" antenna of Figure 5A. However, for the fractal antenna 95, the straight physical length of one side KS of the element will be significantly smaller, and for the fractal antenna of N = 2, shown in Figure 5B, KS ~ 0.13 λ (in air ), compared to K 0.25 λ for the prior art antenna 5.

However, although the effective perimeter length of element 100 is greater than the 1 λ perimeter of prior art element 10, the area within the antenna element 100 is significantly less than the area S²from element 10 of prior technology. As has been observed, this independence of area with respect to perimeter is a characteristic of a deterministic fractal. Boom length B for antenna 95 will be slightly different

ES 2 236 745 T3 to length B for the prior art antenna 5 shown in Figure 4A. In Figure 5B, a parasitic element 120, which is preferably similar to the controlled element 100, but need not be, may be attached to the boom 130. For ease of illustration, Figure 5B does not include the non-conductive extenders, such as extenders 40 shown in Figure 4A, which help hold element 100 and element 120 together. Furthermore, to facilitate understanding of the figure, element 10 is drawn with thicker lines than element 120, in order to avoid confusion in the portion of the figure in which elements 100 and 120 appear overlapping.

An impedance matching device 60 is conveniently unnecessary for the fractal antenna of Figure 5B, since the control impedance of element 100 is around 50 Ω, e.g. eg, a perfect match for wire 50 if reflector element 120 is absent, and around 35 Ω, still acceptable impedance match for wire 50, if element 120 is present. Antenna 95 can be powered by wire 50 from essentially any part of element 100, e.g. eg, including X, Y, Z locations, among others, without any substantial change in termination impedance. With wire 50 connected as shown, antenna 95 will exhibit horizontal polarization. If vertical polarization is desired, the connection can be made, as shown, with lead 50 '. If desired, both cables 50 and 50 'may be present, and an electronic switching device 75 at the antenna end of these cables can short-cut one of the cables. If cable 50 is shorted at the antenna, vertical polarization is obtained, and if, instead, cable 50 'is shorted at antenna, horizontal polarization is obtained.

As shown in Table 3 included here, the fractal “quad” antenna 95 exhibits a gain of around 1.5 dB with respect to the Euclidean “quad” antenna 10. In this way, the transmission power emitted by the transceiver 70 can be cut perhaps 40%, yet the system of Figure 5B will not perform worse than the prior art system of Figure 5A. Furthermore, as shown in Table 1, the fractal antenna of Figure 5B exhibits more resonance frequencies than the antenna of Figure 5B, and also exhibits some resonant frequencies that are not harmonically linked to each other. As shown in Table 3, antenna 95 has an efficiency in excess of 92%, and exhibits an excellent SWR of about 1.2: 1. As shown in Table 5, Applicant's fractal "quad" antenna exhibits a relatively low Q value. This result is surprising, in view of the conventional wisdom of the prior technology, in that small loop antennas will always exhibit a high Q.

In short, that the fractal 95 “quad” antenna works is already surprising, in view of the misunderstanding of the prior technology as to the nature of resistance to radiation R and ohmic losses O. In truth, the prior technology would predict that since the fractal antenna in Figure 5B is smaller than the conventional antenna in Figure 5A, the efficiency would be affected, due to a predicted decrease in resistance to radiation R. Also, Q would be expected to be unduly high for a fractal "quad" antenna.

Figure 6 is a free space radiation pattern generated by ELNEC for a second iteration Minkowski fractal antenna, an antenna similar to that shown in Figure 5B, with parasitic element 120 omitted. The frequency of interest was 42.3 MHz, and an SWR of 1.5: 1 was used. In Figure 6, the outer ring represents 2,091 dBi, and a maximum gain of 2,091 dBi. (ELNEC is a PC graphics version of MININEC, which is a PC version of NEC). In practice, however, the data shown in Figure 6 was cautious, in that a 4.8 dB gain was effectively obtained over a reference isotropic irradiator. The error in the profit figures associated with Figure 6, presumably, is due to rounding and other limitations inherent in the ELNEC program. However, Figure 6 is believed to accurately illustrate the relative gain irradiation pattern of a single element Minkowski (MI-2) fractal "quad" antenna, in accordance with the present invention.

Figure 7A illustrates a third iteration Cantor comb fractal dipole antenna in accordance with the present invention. Generation of a Cantor comb involves trisecting a basic shape, e.g. eg, a rectangle, and provide a rectangle of one third of the basic shape over the ends of the basic shape. The new, smaller rectangles are then trisected, and the process is repeated. Figure 7B is modeled after Lauwerier's treatise, and illustrates a single element broken-leaf fractal "quad" antenna.

As described hereinafter, the fractal element shown in Figure 7B can be used as a ground counterweight for an antenna system, eg, for a vertical antenna. In such an application, the center conductor of cable 50 would be coupled to the lower end of the vertical antenna element (not shown, but which may be a fractal itself), and the ground shield of cable 50 would be coupled with the fractal element shown. in Figure 7B. The fractal counterweight can be significantly smaller than a conventional 0.25 λ ground system, without detriment to the gain, coupling impedance, and vertical polarization characteristics of the antenna system.

Figure 7C-1 illustrates a printed circuit antenna, in which the antenna is manufactured using printed circuit or semiconductor manufacturing techniques. For ease of understanding, the etch-delineated, non-conductive portion of the printed circuit board 150 is shown scratched, and the copper or other conductive fingerprints 170 are shown without scratching.

Applicant notes that while various corners of the rectangular Minkowski motif may appear as touching in this, and perhaps other figures included herein, in fact no contact occurs. Furthermore, it is understood that it is sufficient if an element according to the present invention is essentially a fractal. By this is meant that a

ES 2 236 745 T3 deviation of less than, say, 10% from a perfectly drawn and implemented fractal will still provide adequate fractal-like performance, based on actual measurements made by the applicant.

Substrate 150 is covered by a conductive layer of material 170 that is etch delineated, or otherwise removed in areas other than the fractal pattern, to expose substrate 150. Remaining conductive footprint portion 170 defines a fractal antenna , a second iteration Minkowski slot antenna in Figure 7C-1. Substrate 150 can be a silicone wafer, a rigid or flexible plastic-like material, perhaps Mylar ™ material, or the non-conductive portion of a printed circuit board. Overlay material 170 may be doped polysilicon deposited for a semiconductor substrate 150, or copper, for a printed circuit board substrate.

If desired, the fractal structure shown in Figure 7C-1 could be used as a fractal ground counterweight for an antenna system, for example a vertical antenna. The fractal ground counterweight can be manufactured using smaller dimensions than a conventional prior art system, typically employing radials or 0.25 λ ground elements. If the structure shown in Figure 7C-1 was used as a ground counterweight, the center conductor of cable 50 would mate with the vertical member (not shown), and the ground shield would mate with the fractal structure shown.

Figure 7C-2 illustrates a slot antenna version of what was shown in Figure 7C-2, where the conductive portion 170 (shown hatched in Figure 7C-2) surrounds and defines a non-conductive substrate fractal shape. 150. The electrical connection to the slot antenna is made with a coaxial or other cable 50, the inner and outer conductors of which make contact as shown.

In Figures 7C-1 and 7C-2, the substrate or plastic-like material in such constructions can contribute a dielectric effect that can somewhat alter the performance of a fractal antenna, reducing the resonant frequency, increasing compression. perimeter CP.

Those skilled in the art will appreciate that by virtue of the relatively large amount of conductive material (in contrast to a thin wire), the efficiency of the antenna is promoted in a slot configuration. Of course, a printed circuit board or substrate type construction could be used to implement a non-slot fractal antenna, e.g. eg, one in which the fractal pattern is fabricated as a conductive fingerprint and the remainder of the conductive material is etched or otherwise removed. Thus, in Figure 7C, if the scratched surface now represents the non-conductive material, and the non-scratched material represents conductive material, the result is a wire-type, printed circuit board or substrate implemented fractal antenna.

Fractal antennas implemented with printed circuit board and / or substrate are especially useful at frequencies of 80 MHz or higher, where the fractal dimensions are effectively made small. A 2M MI-3 fractal antenna (eg, Figure 7E) will measure about 5.5 inches (14 cm) on one KS side, and a MI-2 fractal antenna (eg, Figure 5B) will be about 7 inches (17.5 cm) per KS side. As will be seen in Figure 8A, an MI-3 antenna suffers a slight loss in gain relative to an MI-2 antenna, but offers a significant reduction in size.

The applicant has fabricated a MI-2 Minkowski Island Fractal Antenna for operation in the 850-900 MHz cell phone band. The antenna was fabricated on a printed circuit board and measured about 1.2 inches (3 cm ) on the KS side. The antenna was small enough to fit inside the applicant's cell phone, and it worked just as well as if the normal attached semi-rigid rubber whip antenna was still attached. The antenna was placed on the side to obtain the desired vertical polarization, but could be fed anywhere on the element, with an impedance of 50 Ω still inherently present. The applicant also fabricated on a printed circuit board an MI-3 Minkowski island fractal "quad" antenna, the KS lateral dimension of which was about 0.8 inches (2 cm), the antenna being again inserted into the cell phone. . The MI-3 antenna seemed to work just as well as the normal whip antenna, which was not connected. Again, any slight loss of gain when going from MI-2 to MI-3 (eg, perhaps a 1 dB loss relative to a reference MI-0 “quad” antenna, or a 3 dB loss relative to an MI-2) is more than compensated for by the resulting size shrinkage. At satellite phone frequencies around 1650 MHz, the dimensions would again be reduced by about half. Figures 8A, 8B and 8C illustrate preferred embodiments for such antennas.

Figure 7D illustrates a deterministic 2M dendrite fractal antenna, which includes a slight measure of randomness. The vertical arrays of numbers illustrate the wavelengths with respect to 0 λ, at the lower end of the stem-aspect element 200. Eight radial-aspect elements 210 are arranged at 1.0 λ, and various other elements are arranged vertically at a plane along the length of element 200. The antenna was manufactured using 12 gauge copper wire, and was found to exhibit an astonishing 20 dBi gain, which is at least 10 dB better than any antenna twice the size of the one shown in Figure 7D. Although on the surface the vertical antenna of Figure 7D may appear analogous to a periodic-logarithmic antenna, a vertical fractal antenna according to the present invention is not based on an aperture angle, in stark contrast to prior art periodic-logarithmic designs.

Figures 7D-1A and 7D-1B illustrate a conventional vertical antenna 5, comprising a vertical element 195 0.25 λ long and three ground plane radials 205 0.25 λ long. Antenna 5 is powered using cable

ES 2 236 745 T3 coaxial 50 in the conventional manner, the impedance of the antenna being on the order of about 24 Ω. The efficiency of the antenna can be improved by adding additional radial elements 205; however, doing so often requires more space than is conveniently available. In other configurations, a ground plane or counterweight without radials can be used, e.g. eg, ground or the metal body of a car, in the case of an antenna mounted on a vehicle. The azimuth graph of elevation angle of 0 ° in Figure 7D-1B illustrates the undesirably large horizontal polarization components (the "figure eight" pattern) exhibited by this prior technology vertical system, with the vertical and total gain being around 1.45 dBi.

Figure 7D-2A illustrates an antenna system 5 according to the present invention, including a vertical element 195 and a fractalized ground counterweight system comprising, in this example, three ground radials 215 dendrite fractals. The ground radials mate with the ground shield on cable 50, while the center conductor of cable 50 couples with vertical element 195. Of course, other fractal configurations can be used instead, and a different number of ground radials can also be used.

On the azimuthal graph of Figure 7D-2B, the elevation angle is 0 °, and each radial fractal ground element is only about 0.087 λ. The maximum gain, in the outermost ring in the figure, is 1.83 dBi, and the input impedance is around 30 Ω. Note in Figure 7D-2B that relatively little energy radiates horizontally, and almost all energy radiates vertically, a desirable characteristic for a vertical antenna. It will be appreciated that the 0.087 λ dimensions of the fractal ground plane elements 215 are physically significantly larger than the 0.25 λ elements 205 in the prior art system of Figure 7D-1A. However, the irradiation pattern for the Figure 7D-2A system is effectively better than that of the larger, prior-technology system.

Figure 7D-3A illustrates a vertical loaded antenna 5, called a "top hat", according to prior technology. The antenna 5 includes a vertical element 195 and, in the example shown, a top hat assembly comprising three blades 207 located at the upper end of the antenna. The antenna is fed conventionally with coaxial cable 50. Figure 7D-3B illustrates the radiation pattern for the conventional top-hat loaded antenna of Figure 7D-3A.

Figure 7D-4A illustrates a "top hat" antenna 5 that includes a vertical element 195, the upper end of which is loaded with a top hat assembly, including the fractalized radial blades 215. The antenna 5 can be fed conventionally over the wire. coaxial 50. For the same vertical length of element 195 that was used in Figure 7D-3A, the use of radial fractal blades 215 conveniently decreases the resonant frequency by 20%. In addition, the size of the "top hat" assembly can be reduced by about 20%, and the area required for the "top hat" assembly can be reduced by about 35%. These reductions are desirable in that the fractalized top hat antenna of Figure 7D-4A may require less material to manufacture, thereby reducing manufacturing cost, weight, and wind resistance, relative to a top hat configuration. of previous technology. According to the present invention, it is sufficient that at least one of the elements in the top hat assembly has a physical shape defined, at least in part, by a fractal. Of course, more than three blades, or fewer, can be used, and other fractal configurations can also be used, including combinations of fractal and non-fractal elements, as well as different types of fractal elements.

Figure 7D-4B represents the radiation pattern for the fractalized top hat antenna of Figure 7D-4A. A comparison of Figures 7D-4B and 7D-3B confirms that there is no effective performance penalty associated with using the fractalized configuration. In this way, the aforementioned savings in cost, weight and wind resistance are essentially penalty free.

Figure 7D-5 illustrates an antenna system in accordance with the present invention, in which fractal ground elements 215 and a fractal vertical element 197 are employed. The fractal antenna elements 215 are preferably around 0.087 λ, and element 197 is around λ / 12. Fractal vertical element 197 preferably comprises a pair of spaced elements, such as those generally described with respect to Figures 11A, 12A, 12B, 13B, 14A, 14B, and 14C. It is to be understood, however, that the salient feature of element 197 in Figure 7D-3 is not its specific shape, but rather that it defines a fractal and, preferably, a pair of spaced apart fractal elements. It is only for ease of illustration that the fractal elements shown in Figures 7D-3, 11A, 12A, 12B, 13B, 14A, 14B, 14C, and 14D are drawn similarly. Furthermore, the fractal-fractal antenna system shown in Figure 7D-3 is refined, preferably by varying the separation distance Δ, and / or by rotating the spaced elements, each relative to the other, and / or by forming a "cut" in one element, as described hereinafter with respect to several of Figures 11a, 12A, 12B, 13B, 14A, 14B, 14C and 14D.

Figure 7E illustrates a third iteration Minkowski island "quad" antenna (denoted here as MI-3). The orthogonal line segments associated with the Minkowski rectangular motif make this configuration especially acceptable for numerical study, using ELNEC and other numerical tools that use moments to estimate energy patterns, among other modeling schemes. By testing various fractal antennas, the Applicant gained the opinion that the right angles present in the Minkowski pattern are especially suitable for electromagnetic frequencies.

With respect to the MI-3 fractal of Figure 7E, the applicant found that the antenna becomes a vertical if the center conductor of the coaxial cable 50 is connected anywhere with the fractal, but the braided shield is left

ES 2 236 745 T3 of the external coaxial disconnected at the end of the antenna. (At the end of the transceiver, the external shield is connected to ground.) Not only do the fractal antenna islands respond as vertical antennas when the center conductor of cable 50 is attached to all but one side of the island and the braid is left ungrounded at the antenna, but the resonance frequencies for the antenna coupled thus are significantly reduced. For example, a 2-inch (5 cm) sized MI-3 fractal antenna resonated at 70 MHz when coupled like this, which is equivalent to CP ~ 20 perimeter compression.

Figure 7F illustrates a second iteration Koch fractal dipole, and Figure 7G a third iteration dipole. Figure 7H illustrates a second iteration Minkowski fractal dipole, and Figure 7I a third iteration multifractal dipole. Depending on the frequencies of interest, these antennas can be made by bending wire, or etch-delineating or otherwise fingerprinting a substrate. Each of these dipoles essentially provides a terminating impedance of 50 Ω, to which the coaxial cable 50 can be directly coupled, without any impedance matching device. It is understood in these figures that the central conductor of cable 50 is attached to one side of the fractal dipole, and the outer braided shield to the other side.

A fractal ground counterweight can be fabricated using a fractal element, as shown in any (or all) of Figures 7E-7I. Thus, in Figures 7D-2A and 7D-3, it is understood that the fractal radial ground elements 215 illustrate any fractal of iteration order N> 1. Furthermore, such fractals can be defined, but need not be, by a opening angle.

Figure 8A illustrates a generalized system in which a transceiver 500 is coupled with a fractal antenna system 510 to send electromagnetic radiation 520 and / or receive electromagnetic radiation 540. A second transceiver 600, shown equipped with a vertical antenna 610 of Conventional whip type, it also sends 630 electromagnetic energy and / or receives 540 electromagnetic energy.

The fractal antenna system 510 may include a fractal ground counterweight and / or a fractal antenna element, as described hereinbefore. As has been observed in the case of a vertical antenna element, the overall size of the resulting antenna system is significantly smaller than can be achieved with a prior technology ground counterbalance system. In addition, the fractal ground counterbalance system can be fabricated on a flexible substrate that is rolled, or otherwise patterned, to fit within a housing such as that containing the transceiver 500. The resulting antenna ground system exhibits an improved pattern of energy efficiency and distribution over a prior technology system that can somehow be fitted into an area of ​​equivalent magnitude.

If the transceivers 500, 600 are communication devices, such as transceivers, cordless telephones, pagers, or the like, a communications repeater unit may be present, such as a satellite 650 and / or a ground-based repeater unit 660, coupled with an antenna 670, or with a fractal antenna according to the present invention.

Alternatively, antenna 510 on transceiver 500 could be a passive LC resonator fabricated on an integrated circuit microchip, or other similarly small sized substrate, attached to a valuable item to be protected. The transceiver 600, or unit 660, would then be an electromagnetic transmitter emitting energy at the resonant frequency, a unit typically located near the cash register control area of ​​a store or at an outlet.

Depending on whether the fractal antenna-resonator 510 is designed to "jump" (eg, become open circuit) or "clip" (eg, become closed circuit) on the transceiver 500, it will or will not reflect back electromagnetic energy 540 or 6300 to a receiver associated with transceiver 600. In this manner, unauthorized relocation of antenna 510 and / or transceiver 500 can be signaled by transceiver 600.

Figure 8B illustrates a transceiver 500 equipped with multiple fractal antennas, shown here as 510A, 510B, 510C, and 510D, coupled by respective leads 50A, 50B, 50C, 50D, with electronics 600 within unit 500. In the embodiment shown, one or more of these antenna elements is (are) fabricated on a compliant, flexible substrate 150, e.g. eg, Mylar ™ material or similar, on which the antennas can be implemented per se, by printing fractal patterns using conductive ink, or by copper deposits, among other methods that include printed circuit board and semiconductor manufacturing techniques. Such a flexible substrate can be shaped into a rectangular, cylindrical or other shape, as required.

In the embodiment of Figure 8B, unit 500 is a handheld transceiver, and antennas 510A, 510B, 510C, 510D are preferably powered for vertical polarization, as shown. Element 510D may, for example, be a fractal ground counterweight system for a vertical antenna element, shown in dotted line as element 193 (element which may itself be a fractal to further reduce dimensions).

An electronic circuit 610 is coupled by leads 50A, 50B, 50C with the antennas, and samples incoming signals to discern which fractal antenna system, e.g. eg, 510A, 510B, 510C, 510D, is currently optimally aligned with the transmitting station, perhaps a 600 or 650 or 670 unit, as shown in Figure 8A. This determination can be made by examining the signal strength of each of the antennas. An electronic circuit 620 then selects the currently best oriented antenna, and couples that antenna to the receiver input and the

ES 2 236 745 T3 output of the transmitting portion, collectively 630, of the unit 500. It is understood that the selection of the best antenna is dynamic and that it can change, for example, as a user of 500 walks holding the unit, or the Transmitting source is moving, or due to other changing conditions. In a cell phone or cordless application, the result is more reliable communication, with the advantage that the fractal antennas can be small enough in size to fit fully inside the 500 unit housing. Also, if used a flexible substrate, the antennas can wrap around portions of the inner shell, as shown.

A further advantage of the embodiment of Figure 8B is that the user of unit 500 can be physically away from the antennas, at a greater distance than if a conventional external whip antenna were employed. Although the medical evidence attempting to link cancer to exposure to electromagnetic radiation from handheld transceivers is not yet conclusive, the embodiment of Figure 8B appears to minimize any similar risks. Although Figure 8B illustrates a vertical antenna 193 and a fractal ground counterweight 510D, it is understood that the antenna 193 could represent a cellular antenna on a motor vehicle, the counterweight of which is the fractal unit 510D. Furthermore, as noted, vertical element 193 may itself be a fractal.

Figure 8C illustrates yet another embodiment, in which some or all of the antenna systems 510A, 510B, 510C may include electronically steerable arrays, including fractal antenna arrays of various sizes and polarization orientations. The 510C antenna system, for example, may include similarly designed fractal antennas, e.g. eg, antennas F-3 and F-4, which are oriented differently from each other. Other antennas within the 510C system may be of a different design than the F-3 and F-4. The fractal antenna F-1 can be a dipole, for example. The conductors of the various antennas in the 510C system can be coupled with an integrated circuit 690, mounted on the substrate 150. The circuit 690 can determine the relative optimal selection between the antennas that comprise the system 510C and transmit on the cable 50C to the electronics 600 associated with the transmitting and / or receiving portion 630 of the unit 630. Of course, the embodiment of Figure 8C could also include vertical antenna element 193 and fractal ground counterweight 510D, illustrated in Figure 8B.

Another 510B antenna system may include a steerable array of identical fractal antennas, including fractal antennas F-5 and F-6. An integrated circuit 690 is coupled to each of the antennas in the array, and dynamically selects the best antenna for its signal strength, and couples such antenna along wire 50B with electronics 600. A third antenna system 510A may be different. , or identical, to any of the 510B and 510C systems.

Although Figure 8C illustrates a unit 500 that may be handheld, unit 500 could, in fact, be a tabletop communications system, or a field mountable unit, perhaps unit 660, as shown. in Figure 8A.

To facilitate pairing of antennas with a transceiver load, the resonance of a fractal antenna was defined as a total impedance that falls between about 20 Ω and about 200 Ω, and the antenna was required to exhibit a medium to high Q, p . eg, frequency ^ frequency. In practice, Applicants' various fractal antennas were found to resonate at at least one position of the antenna feed point, e.g. eg, the point at which the coupling with the antenna was made. Furthermore, multiple iteration fractals, according to the present invention, were found to resonate at multiple frequencies, including frequencies that were nonharmonically linked.

Contrary to conventional wisdom, Applicant found that island-shaped fractals (eg, a closed-loop configuration) do not exhibit significant drops in resistance to R radiation with decreasing antenna size. As described here, fractal antennas were constructed with dimensions less than 12 inches long (30.48 cm) and yet resonated in a desired frequency band between 60 MHz and 100 MHz. The applicant further discovered that the antenna perimeters do not correspond to the lengths that could be anticipated from the measured resonant frequencies, the effective lengths being longer than expected. This increase in the length of the elements appears to be a property of fractals as radiators, and not a result of geometric construction. A similar elongation effect was reported by Pfeiffer when constructing a large “quad” antenna, using a first order fractal; see A. Pfeiffer, The Pfeiffer Quad Antenna System, QST, p. 28-32 (March 1994).

If L is the total initial one-dimensional length of a pre-pattern fractal application, and r is the one-dimensional length of the post-pattern application, the resulting fractal dimension D (in effect, a ratio limit) is:

D = log (L) / log (r)

Referring to Figure 1A, for example, the length of Figure 1A represents L, while the sum of the four line segments comprising the Koch fractal of Figure 1B represents r.

Unlike mathematical fractals, fractal antennas are not characterized solely by the ratio D. In practice D is not a good predictor of how much smaller a fractal design antenna can be, because D does not incorporate the perimeter elongation of an element antenna irradiator.

ES 2 236 745 T3

Since D is not a particularly useful predictive parameter in the design of fractal antennas, a new parameter of “perimeter compression” (“CP”) will be used, where

CP _ actual size antenna element length reduced antenna element length by fractal

In the preceding equation, measurements are made at the lowest resonant frequency of the fractal-resonant element. Thus, for a life-size antenna, according to the prior technology, CP = 1, while CP = 3 represents a fractal antenna, according to the present invention, in which one side of the element has been reduced by a factor of three.

Perimeter compression can be represented empirically using the fractal dimension D as follows: CP = A log [N (D + C)] where A and C are constant coefficients for a given fractal pattern, N is an iteration number, and D is the fractal dimension, defined above.

It is seen that for each fractal, CP becomes asymptotic to a real number and yet it does not approach infinity, even when the iteration number N becomes very large. Expressed differently, the CP of a fractal irradiator asymptotically approaches a non-infinite limit in a finite number of fractal iterations. This result is not a representation of a purely geometric fractal.

It follows that some fractals are better resonant elements than other fractals, because optimized fractal antennas approach their asymptotic CPs in fewer iterations than non-optimized fractal antennas. In this way, the best fractals for antennas will have large values ​​for A and C, and will provide the largest and fastest element size shrinkage. The fractal used can be deterministic or chaotic. Deterministic fractals have a pattern that replicates at a 100% level on all size scales, while chaotic fractals include a random noise component.

Applicant found that the radiation resistance of a fractal antenna decreases as a small perimeter compression (CP) power, with a fractal island always exhibiting significantly higher radiation resistance than a small Euclidean loop antenna of the same size.

Also, it seems that the number of resonant nodes of a fractal island increases with the iteration number (N) and is always greater than or equal to the number of resonant nodes of a Euclidean island with the same area. Finally, it appears that a fractal resonator has an increased effective wavelength.

The above findings will now be applied to experiments conducted by the applicant with fractal resonators, in the form of closed loops or islands. The antennas analysis of the previous technology would not predict any resonance points, but, as shown below, this is not the case.

A Minkowski motif is illustrated in Figures 2B-2D, 5B, 7C and 7E. The selected Minkowski motif was a three-sided enclosure (eg, 20-2 in Figure 2B) placed on top of a line segment. The sides of the enclosure can be any arbitrary length, eg. eg, an enclosure height and width of 2 units, with the remaining two base sides being three units long (see Figure 2B). For such a configuration, the fractal dimension D is as follows:

log (L) log (r) log (12) log (8)

1,08 090 _ 1,20

It will be appreciated that D = 1.2 is not especially high. compared to other deterministic fractals.

The application of the motif to the line segment can be expressed very simply by the function f (x) defined piecemeal by the following:

f (x) _ 0 ^vmax f (x) _

-max ^vmax max <x <

ES 2 236 745 T3 f (x) - 0 ^vmax

8 ~ max where xmax is the largest continuous value of x in the line segment.

A second iteration can be expressed as f (x) 2, linked to the first iteration f (x) 1 by:

f (x) 2 = f (x) 1 + f (x) where xmax is defined in the above-mentioned piecewise function. Notice that each individual horizontal line segment will have a different lower value of x and xmax. Relevant offsets from zero can be entered as needed, and vertical segments can be “locked in” by 90 ° rotation and application of the above methodology.

As shown in Figures 5B and 7E, a Minkowski fractal quickly begins to look like a Moorish design pattern. However, each successive iteration consumes more perimeter, thus reducing the overall length of an orthogonal line segment. Four quadrilateral or rectangular shaped fractals from the same N iteration can be combined to create a Minkowski fractal island and a resulting "fractalized" cubic "square."

An ELNEC simulation was used as a guide for far-field energy patterns, resonant frequencies, and SWR values ​​from fractal antennas from Minkowki Island, up to iteration N = 2. Analysis for N> 2 was not undertaken due to deficiencies in the test equipment available to the applicant.

The following table summarizes the applicant's ELNEC simulated fractal antenna designs, undertaken to derive the lower resonance frequencies and energy patterns, up to and including iteration N = 2. All designs were built on the x, y axes, and for each iteration the external length was kept at 42 inches (106.7 cm).

Table 1, below, summarizes the ELNEC-derived far-field radiation patterns for Minkowski island “quad” antennas, for each iteration, for the first four resonances. In Table 1, each iteration is designated MI-N, for the Minkowski Island of iteration N. Notice that the frequency of the lowest resonances decreased with fractal Minkowski Island antennas, compared to a “quad” antenna. ”From previous technology. In other words, for a given resonant frequency, a fractal Minkowski Island antenna will be smaller than a conventional “quad” antenna.

TABLE 1

<td>Antenna</td><td>Resonant Frequency (MHz)</td><td>Gain (dBi)</td><td>SWR</td><td>CP (for 1^to)</td><td>Direction</td>

<td>"Reference square</td><td> 76</td><td> 3,3</td><td> 2,5</td><td> 1</td><td>Long side</td>

<td></td><td> 144</td><td> 2,8</td><td> 5,3</td><td> —</td><td>Shooting extreme</td>

<td></td><td> 220</td><td> 3,1</td><td> 5,2</td><td></td><td>Shooting extreme</td>

<td></td><td> 294</td><td> 5,4</td><td> 4,5</td><td></td><td>Shooting extreme</td>

<td>MI-1</td><td> 55</td><td> 2,6</td><td> 1.1</td><td> 1,38</td><td>Long side</td>

<td></td><td> 101</td><td> 3,7</td><td> 1,4</td><td> —</td><td>Shooting extreme</td>

<td></td><td> 142</td><td> 3,5</td><td> 5,5</td><td></td><td>Shooting extreme</td>

<td></td><td> 198</td><td> 2,7</td><td> 3,3</td><td> -</td><td>Long side</td>

<td>MY 2</td><td> 43,2</td><td> 2,1</td><td> 1,5</td><td> 1,79</td><td>Long shot</td>

<td></td><td> 85,5</td><td> 4,3</td><td> 1.8</td><td> —</td><td>Shooting extreme</td>

<td></td><td> 102</td><td> 2,7</td><td> 4,0</td><td></td><td>Shooting extreme</td>

<td></td><td> 116</td><td> 1,4</td><td> 5,4</td><td> --</td><td>Long side</td>

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It appears from Table 1 that Minkowski Island fractal antennas are multi-resonant structures that have virtually the same gain as larger, full-size conventional "quad" antennas. The gain figures in Table 1 are for "free space", in the absence of any ground plane, but simulations on a perfect ground surface at 1 λ yielded similar gain results. Understandably, there will be some inaccuracy in the ELNEC results, due to rounding and insufficient pulse samples, among other factors.

Table 2 presents the ratio of resonant frequencies derived by ELNEC for the first four resonant nodes mentioned in Table 1.

TABLE 2

<td>Antenna</td><td>SWR</td><td>SWR</td><td>SWR</td><td>SWR</td>

<td>Reference “Square” (Ml0)</td><td> 1:1</td><td> 1:1,89</td><td> 1:2,89</td><td> 3,86:1</td>

<td>MI-1</td><td> 1:1</td><td> 1:1,83</td><td> 1:2,58</td><td> 3,6:1</td>

<td>MY 2</td><td> 1:1</td><td> 2,02:1</td><td> 2,41:1</td><td> 2,74:1</td>

Tables 1 and 2 confirm the shrinkage of an antenna designed with fractals, and the increase in the number of resonance points. In the preceding simulations, the MI-2 fractal antenna exhibited four resonance nodes before the prior technology reference “quad” antenna exhibited its second resonance. The near fields in the antennas are very important, since they are combined in antennas of multiple element to achieve high gain . Unfortunately, ELNEC's inherent scheduling limitations preclude serious investigations of upcoming fields. However, as described hereinafter, the applicant has designed and constructed several different high gain fractal formations that exploit the near field.

The applicant fabricated three Minkowski Island fractal antennas with # 8 aluminum and / or thinner # 12 galvanized ground wire. The antennas were designed so that the lowest operating frequency was close to a desired frequency in the 2M amateur radio band (144 MHz), to facilitate relative gain measurements using 2M FM repeater stations. The antennas were mounted for vertical polarization and positioned so that their center points were the highest practical point on the mounting platform. For gain comparisons, a vertical ground plane with three reference radials, and a reference “quad” antenna, were constructed using the same size wire as the fractal antenna under test. Measurements were made in the receiving mode.

Multi-way reception was minimized with careful antenna placement. Low height effects were reduced and clearance testing approaches were achieved by mounting the antenna test platform on the edge of a third floor window, providing 3.5 λ clearance above the ground, and in the line of sight to the repeater, 45 miles (28 km) distant. The antennas were pulled out the window at a distance of about 0.8 λ from any metallic object, and the test was repeated five times from different windows on the same floor, the test results being consistent within 1/2 dB for each attempt.

Each antenna was attached to a short piece of 50 Ω 9913 coaxial cable, connected at right angles to the antenna. A 2M transceiver, with 9913 coaxial cable, was coupled with two precision antenna attenuators under test. The transceiver S counter was coupled with a volt-ohm counter to provide signal strength measurements. The attenuators were used to insert an initial threshold, in order to avoid the problems associated with non-linear readings of the S-counter, and with the saturation of the S-counter in the presence of total attenuation of the background noise.

Each antenna was quickly activated for the volt-ohm counter measurement, with the attenuation added or removed to obtain the same counter reading as seen on the reference "quad" antenna. All readings were corrected for SWR attenuation. For the reference “quad” antenna, the SWR was 2.4: 1 for an impedance of 120 Ω, and for the fractal “quad” antennas, the SWR was less than 1.5: 1 at resonance. The lack of a suitable 2M noise bridge prevented efficiency measurements for the various antennas. Understandably, echoless chamber tests would provide even more useful measurements.

For each antenna, the relative leading gain and the optimized physical orientation were measured. No attempt was made to compensate for the launch angle, or to measure energy patterns other than to demonstrate the long-sided nature of the gain. A 1/2 dB difference produced noticeable S-counter drift, and differences of several dB produced significant counter drift. Removal of the antenna from the receiver resulted in a 20+ dB drop in received signal power. In this way, the system distortions in the readings were counteracted to provide more meaningful results. Table 3 summarizes these results.

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TABLE 3

<td>Antenna</td><td>CP</td><td>Length perimeter</td><td>SWR</td><td>Gain compensated (dB)</td><td>Side length (Λ)</td>

<td>"Quad"</td><td> 1</td><td> 1</td><td> 2,4:1</td><td> 0</td><td> 0,25</td>

<td>1/4 wave</td><td> 1</td><td> -</td><td> 1,5:1</td><td> -1,5</td><td> 0,25</td>

<td>MI-1</td><td> 1,3</td><td> 1,2</td><td> 1,3:1</td><td> 1,5</td><td> 0,13</td>

<td>MY 2</td><td> 1.9</td><td> 1,4</td><td> 1,3:1</td><td> 1,5</td><td> 0,13</td>

<td>MI-3</td><td> 2,4</td><td> 1,7</td><td> 1:1</td><td> -1,2</td><td> 0,10</td>

It appears from Table 3 that, for the vertical test configurations, a fractal "quad" antenna according to the present invention either exceeded the gain of the prior technology test "quad" antenna, or had a gain deviation of no more 1 dB relative to the test “quad” antenna. Clearly, the older technology "quad" cubic (square) antennas are not optimized for gain. Fractal shrinkage of a cubic “quad” antenna by a factor of two will increase the gain, and the additional shrink will exhibit slight losses of 1 to 2 dB.

MI-2 and MI-3 versions of fractal “quad” antennas were built for the 6M (50 MHz) amateur radio band. A receive 50 Ω noise bridge was attached between these antennas and a transceiver. The receiver was muted around 54 MHz and the noise bridge was calibrated with 5 Ω and 10 Ω resistors. Table 4 below summarizes the results, in which almost no reactance was observed.

TABLE 4

<td>Antenna</td><td>SWR</td><td>Ζ (Ω)</td><td>Ο (Ω)</td><td>Ε (%)</td>

<td>"Quad" (MI-0)</td><td> 2,4:1</td><td> 120</td><td> 5-10</td><td> 92-96</td>

<td>MY 2</td><td> 1,2:1</td><td> 60</td><td> &lt;5</td><td> &gt;92</td>

<td>MI-3</td><td> 1,1:1</td><td> 55</td><td> &lt;5</td><td> &gt;91</td>

In Table 4, the efficiency (E) was defined as 100% * (R / Z), where Z was the measured impedance, and R was Z minus the ohmic impedance and reactive impedances (O). As shown in Table 4, the MI-2 and MI-3 fractal antennas, with their low SWR values, &lt;1.2: 1, and their low ohmic and reactive impedances, provide extremely high efficiency values ​​of 90 +%. These findings are indeed surprising, in view of the teachings of the previous technology, originating in primitive small-loop Euclidean geometries. In fact, Table 4 strongly suggests that the prior technology associations of low radiation impedances for small loops should generally be abandoned, to be invoked only when exposing small Euclidean loops. Applicant's MI3 antenna was indeed very small in size, with dimensions of about 0.1 λ per side, an area of ​​about λ²/1,000 and yet it did not indicate the onset of the inefficiency that has long been thought to accompany smaller sized antennas.

However, the 6M efficiency data does not explain the fact that the MI-3 fractal antenna had a gain drop of almost 3 dB relative to the MI-2 fractal antenna. Low ohmic impedances of &lt;5 Ω strongly suggest that inefficiency is not the explanation, despite the small size of the antenna. It is quite possible that near field diffraction effects occur at higher iterations, resulting in loss of gain. However, the smaller antenna sizes achieved by the higher iterations seem to guarantee the small loss in gain.

Using fractal techniques, however, 2 M “squares” antennas with dimensions less than 3 inches (7.6 cm) per side, as well as 20 M (14 MHz) “squares” of less than 3 feet ( 1 m) per side. Of greater economic interest, fractal antennas built for cell phone frequencies (850 MHz) could have sizes smaller than 0.5 inches (1.2 cm). As shown in Figures 8B and 8C, several such antennas, each oriented differently, could be fabricated within the curved or rectilinear casing of a cellular or cordless phone, with the antenna outputs coupled with a circuit, to couple with the antenna optimally oriented for the signal currently being received. The resulting antenna system would be smaller than the semi-rigid rubber antennas now used by cell phones, and it would also have better features.

ES 2 236 745 T3

Similarly, fractal-designed antennas could be used in military handheld "walkie-talkie" transceivers, global location systems, satellites, transponders, wireless and computer communication networks, remote and / or robotic control systems, among other applications. .

Although the Minkowski island fractal antenna has been described here, other motifs are also useful, as well as non-island fractal configurations.

Table 5 shows bandwidths (“AB”) and multifrequency resonances of the described MI-2 and MI-3 antennas, as well as the Q values, for each node found for the 6 M versions between 30 MHz and 175 MHz. Regardless of resonant frequency SWR, displayed bandwidths are 3: 1 SWR values. The Q values ​​shown were estimated by dividing the resonant frequency by the AB of SWR 3: 1. The frequency ratio is the relative scale of the resonance nodes.

TABLE 5

<td>Antenna</td><td>Freq. (MHz)</td><td>Reason for Freq.</td><td>SWR</td><td>3: 1 AB</td><td>Q</td>

<td>MI-3</td><td> 53,0</td><td> 1</td><td> 1:1</td><td> 6,4</td><td> 8,3</td>

<td></td><td> 80,1</td><td> 1,5:1</td><td> 1,1:1</td><td> 4.5</td><td> 17,8</td>

<td></td><td> 121,0</td><td> 2,3:1</td><td> 2,4:1</td><td> 6,8</td><td> 17,7</td>

<td>MY 2</td><td> 54,0</td><td> 1</td><td> 1:1</td><td> 3,6</td><td> 15,0</td>

<td></td><td> 95,8</td><td> 1,8:1</td><td> 1,1:1</td><td> 7,3</td><td> 13,1</td>

<td></td><td> 126,5</td><td> 2,3:1</td><td> 2,4:1</td><td> 9,4</td><td> 13,4</td>

The Q values ​​in Table 5 reflect that the MI-2 and MI-3 fractal antennas are multiband. These antennas do not exhibit the very high Q values ​​observed in fine tuned Euclidean loops, and there does not appear to be a mathematical application to electromagnetism to predict these resonances or Q values. One approach could be to estimate scalar and vector potentials in Maxwell's equations, considering each iteration of Minkowski Island as a series of vertical and horizontal line segments with displaced positions. The sum of these segments will lead to a Poynting vector calculation and an energy pattern that can be especially useful to better predict the characteristics and optimized shapes of fractal antennas.

In practice, actual Minkowski Island fractal antennas appear to perform slightly better than their ELNEC predictions, almost certainly due to inconsistencies in ELNEC modeling, or to ratios of resonant frequencies, CP, SWR, and gains. .

Those skilled in the art will appreciate that multiband fractal antenna arrays can also be constructed. The resulting formations will be smaller than their Euclidean counterparts, will present less area to the wind, and will be mechanically rotatable by means of a smaller antenna rotor.

In addition, fractal antenna configurations can be implemented using means other than Minkowski islands or loops. Table 6 shows the highest iteration number N for other fractal configurations which, as found by Applicant, resonated on at least one frequency.

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TABLE 6

<td>Fractal</td><td>Maximum Iteration</td>

<td>Koch</td><td> 5</td>

<td>Broken Picture</td><td> 4</td>

<td>Minkowski</td><td> 3</td>

<td>Mandelbrot</td><td> 4</td>

<td>Caley Tree</td><td> 4</td>

<td>Monkey swing</td><td> 3</td>

<td>Sierpinski packing</td><td> 3</td>

<td>Cantor packing</td><td> 3</td>

Figure 9A illustrates the gain with respect to a Euclidean “quad” configuration (eg, an MI-0) as an iteration function of value N. (A Euclidean “quad” antenna is understood to exhibit a gain of 1 , 5 dB relative to a reference standard dipole). For first and second order iterations, the gain of a fractal “quad” antenna increases relative to a Euclidean “quad” antenna. However, beyond the second order, the gain declines with respect to a Euclidean “quad” antenna. The applicant believes that the diffraction-type cancellations of the near field electromagnetic energy may explain the loss of gain for N&gt; 2. Possibly, the much smaller areas found in the fractal antennas according to the present invention more sharply focus this phenomenon of diffraction. In practice, the applicant was unable to physically bend the wire for a 2M Minkowski fractal antenna of 4^to or 5^to iteration, although at lower frequencies the larger antenna sizes would not present this problem. However, at higher frequencies, printed circuit techniques, semiconductor fabrication techniques, as well as machine construction could readily produce fractal antennas of N = 4, N = 5, and higher order iterations.

In practice, a Minkowski Island fractal antenna could reach the theoretical limit of gain, of around 1.7 dB, observed for down-wavelength Euclidean loops, but N will be greater than 3. More cautiously However, a Minkowski Island fractal “quad” antenna with N = 4 should provide a CP = 3 value without exhibiting significant inefficiency.

Figure 9B illustrates perimeter compression (CP) as a function of iteration order N for a Minkowski island fractal configuration. A conventional Euclidean “quad” antenna (MI-0) has CP = 1 (eg, no compression) and as iteration increases, CP increases. Note that as N increases and approaches 6, the CP approaches an asymptotically finite real number, as predicted. In this way, the Minkowki Island fractal antennas, beyond the N = 6 iteration, can exhibit decreasing performance with each increase in the iteration.

It will be appreciated that the nonharmonic resonant frequency characteristic of a fractal antenna according to the present invention can be employed in a system in which the frequency signature of the antenna must be recognized to pass a safety test. For example, at conveniently high frequencies, perhaps several hundred MHz, a fractal antenna could be implemented inside a credit card with identification. When the card is used, a transmitter associated with a credit card reader can electronically sample the frequency resonance of the antenna within the credit card. If, and only if, the credit card antenna responds with the expected signature pattern of the appropriate frequency, the credit card can be used, eg. eg for acquisitions or to allow the owner to enter a protected area otherwise.

Figure 10A illustrates a fractal inducer L according to the present invention. In contrast to a prior art inductor, the winding or fingerprints from which L is made define, at least in part, a fractal. The resulting inductor is physically smaller than its Euclidean counterpart. Inductor L can be used to form a resonator, including resonators such as those shown in Figures 4A and 4B. As such, an integrated circuit or other suitably small container that includes fractal resonators could be employed as part of a safety system, in which electromagnetic radiation, possibly from transmitter 600 or 660 in Figure 8A, will excite, or perhaps not excite. , an LC resonator circuit containing the fractal antenna. Such applications are described elsewhere herein, and may include a credit card-sized unit 700, as shown in Figure 10b, in which an LC 710 fractal resonator is implemented. (Card 700 is illustrated in Figure 10b. Figure 10B as if its upper surface were transparent).

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The preceding description has largely reproduced what has been stated in US-A-6104309 and / or US-A-6140975. The next section will outline the procedures and techniques for tuning such antennae and fractal resonators. In the following description, although the expression "antenna" can be used when referring to a preferably fractal element, in practice, what is being described is an antenna or filter-resonator system. As such, an "antenna" can be made to act as if it were a filter, eg. eg, letting certain frequencies pass and rejecting other frequencies (or vice versa).

In one group of embodiments, Applicant has discovered that placing a fractal antenna at a distance Λ that is in the closest environment (eg, less than about 0.05 λ for the frequency of interest) to a conductor , the resonant properties and radiation characteristics of the antenna can be conveniently changed (with respect to such properties and characteristics when there is not such a close proximity, e.g. g., when the separation distance is relatively large). For example, in Figure 11A a conductive surface 800 is positioned at a distance Λ behind or below a fractal antenna 810, which in Figure 11A is a single arm of a fractal MI-2 antenna. Of course, other fractal configurations, such as those disclosed herein, could be used in place of the MI-1 configuration shown, and non-coplanar configurations can also be used. The fractal antenna 810, preferably, is fed with the power line 50 by coaxial cable, whose central conductor is attached to one end 815 of the fractal antenna, and whose external shield is connected to ground by the conductive plane 800. As described Here, there is great flexibility to connect the antenna system shown with a preferably coaxial feed line. The termination impedance is approximately similar in magnitudes to those described hereinabove.

In the configuration shown, the relatively close proximity between the conductive foil 800 and the fractal antenna 810 decreases the resonant frequencies and broadens the bandwidth of the antenna 810. Conductive foil 800 may be a metal plane, the upper copper surface of a printed circuit board, a region of conductive material, possibly sprayed onto the cover of a device employing the antenna, for example, the inside of a cover. transceiver 500, such as that shown in Figures 8A, 8B, 8C and 15.

The relationship between Δ, where Λ &lt;0.05 λ, and the resonant properties and radiation characteristics of a fractal antenna system is generally logarithmic. That is, the resonant frequency decreases logarithmically with decreasing separation Δ.

Figure 11B shows an embodiment in which a preferably fractal antenna 810 lies in the same plane as a ground plane 800, but is separated therefrom by an insulating region, and in which a passive or parasitic element 800 'is arranged " within ”and separated by a distance Δ 'from the antenna, also being coplanar. For example, the embodiment of Figure 11B can be fabricated from a single piece of printed circuit board material, in which the copper (or other conductive material) remains to define the ground plane 800, the antenna 810, and the parasitic element 800 ', the remaining portions of the source material having been etched to form the "pit-like" regions separating regions 800, 810, and 800'. Changing the shape and / or size of the element 800 'and / or the coplanar spacing distance Δ' fine-tunes the antenna system shown. For example, for a center frequency in the 900 MHz range, element 800 'measured about 63mm x 8mm, and each of elements 810 and 800 measured about 25mm x 12mm. In general, element 800 should be at least as large as antenna 810, preferably fractal. For this configuration, the system shown exhibited a bandwidth of about 200 MHz, and could be made to exhibit characteristics of a band pass filter and / or a band rejection filter. In this embodiment, a coaxial feed line 50 was used, in which the center conductor was coupled with the antenna 810, and the ground shield conductor was coupled with the ground plane 800. In Figure 11B, the inner perimeter of the ground plane region 800 is shown as rectangular in shape. If desired, this inner perimeter could be brought closer to the outer perimeter of the preferably fractal antenna 810, and could, in fact, define a perimeter shape that follows the perimeter shape of the antenna 810. In such an embodiment, the perimeter of the inner conductive region 800 'and the inner perimeter of the ground plane region 800 would follow the shape of the antenna 810. Based on experiments to date, Applicant's belief is that the Shifting the inner perimeter of the ground plane region 800 close enough to the antenna 810 could also affect the characteristics of the overall antenna / resonator system.

Referring now to Figure 12A, if the conductive surface 800 is replaced by a second fractal antenna 810 ', which is separated by a distance Δ, which preferably does not exceed about 0.05 λ, the resonances for the fractal antenna irradiating 810 decreases and new resonant frequencies conveniently emerge. To facilitate fabrication, it may be desired to build the antenna 810 on the top or first surface 820A of a substrate 820, and to build the antenna 810 'on the bottom or second surface 820B of the same substrate. The substrate could be double-sided printed circuit board type material, if desired, in which the antennas 810, 810 'are manufactured using printed circuit type techniques. The thickness Δ of the substrate is selected to provide the desired performance for the antenna 810 at the frequency of interest. The substrate 820 can, for example, be a non-conductive film, flexible or not. To avoid overloading Figures 12A and 12B, the substrate 820 is drawn with a dotted line, as if the substrate were transparent.

As noted above, the separate fractal structure illustrated in Figures 12A and 12B can instead be used to form a fractal element in a vertical antenna system, preferably including a fractal ground counterweight, as described. with respect to Figure 8D-3.

ES 2 236 745 T3

Preferably, the center conductor of the coaxial cable 50 is connected to one end 815 of the antenna 810, and the outer conductor of the cable 50 is connected to a free end 815 'of the antenna 810', which is considered to be ground, although they can be used. other power line connections. Although Figure 12A illustrates antenna 810 'as being essentially identical to antenna 810, the two antennas could, in fact, have different configurations.

The applicant has discovered that if the second antenna 810 'is rotated at an angle θ relative to the antenna 810, the resonant frequencies of the antenna 810 can be varied, analogously to the tuning of a variable capacitor. Thus, in Figure 12B, antenna 810 is tuned by rotating antenna 810 'relative to antenna 810 (or the other way around, or by rotating each of the antennas). If desired, substrate 820 could comprise two substrates, each Δ / 2 thick, pivotally connected to each other, e.g. eg, with a non-conductive rivet, in order to allow the rotation of the substrates and, thus, the relative rotation of the two antennas. Those skilled in the mechanical arts will appreciate that various "tuning" mechanisms could be implemented in order to allow fine control over the angle θ in response, for example, to rotation of a tunable shaft.

Referring now to Figure 13A, Applicant has discovered that creating at least one cutout or aperture 830 in a fractal antenna 810 (comprising here two legs of an MI-2 antenna) results in completely new, resonant nodes. different, for the antenna. Also, these nodes can have Edge Compression (CP) ranging from possibly three to around ten. The precise location of the 830 cutout on the fractal antenna or resonator does not appear to be critical.

Figures 13B and 13C illustrate a self-closeness characteristic of fractal antennas and resonators, which can be conveniently used to create a desired resonant frequency offset. In Figure 13B, a fractal antenna 810 is fabricated on a first surface 820A of a flexible substrate 820, the second surface 820B of which does not contain an antenna or other conductor in this embodiment.

The curvature of the substrate 820, which can be a flexible film, appears to cause the electromagnetic fields associated with the antenna 810 to be sufficiently self-approximated to shift the resonant frequencies. Such self-approximated antennas or resonators may be referred to as cil-com devices. The amount of curvature can be controlled where a flexible substrate or a fractal antenna without substrate and / or a conductive element is present, to control or fine-tune the frequency-dependent characteristics of the resulting system. Cyl-com embodiments could include a fractal antenna and conductive element disposed concentrically or eccentrically. Such embodiments can include telescopic elements, the amount of "overlap" of which can be telescopically adjusted by contracting or lengthening the overall configuration, in order to fine-tune the characteristics of the resulting system. Also, more than two items could be provided.

In Figure 13C, a fractal antenna 810 is formed on the outer surface 820A of a filled substrate 820, which may be a ferrite core. The resulting cil-com antenna appears to exhibit such self-proximity as to produce the desired offsets in resonant frequency. The Geometry of the Core 820, p. g., the magnitude of the curvature (p. g., the radius in this embodiment) relative to the size of the antenna 810 can be used to determine the frequency offsets.

In Figure 14A, an antenna or resonator system is shown, in which the uncontrolled fractal antenna 810 'is not connected to the feed line, preferably coaxial 50. The ground shield portion of the feed line 50 it is coupled to ground plane conductive element 800, but is not otherwise connected to system ground. Of course, the fractal antenna 810 'could be rotated angularly relative to the controlled antenna 810, it could be a different configuration than the antenna 810, even having a different N iteration, and it could, in fact, incorporate other features disclosed here. (eg, a cut).

Figure 14B demonstrates that controlled antenna 810 can mate with feed line 50 at any point 815 ', and not necessarily at an end point 815, as shown in Figure 14A.

In the embodiment of Figure 14C, a second ground plane element 800 'is provided, adjacent to at least a portion of the system comprising controlled antenna 810, passive antenna 810', and underlying conductive coplanar element 800. The presence , location, geometry and distance associated with the second ground plane element 800 'from the underlying elements 810, 810', 800, allow the tuning of characteristics of the general antenna or resonator system. In the multi-element configuration shown, in the form of a sandwich, the ground shield of conductor 50 is connected to a system ground, but not to any ground plane 800 or 800 '. Of course, more than three elements could be employed to form a tunable system in accordance with the present invention.

Figure 14D shows a single fractal antenna separated from an underlying ground plane 800 by a distance Δ, in which a region of the antenna 800 is clipped to increase resonance. In Figure 14D, for example, L1 indicates a clipping line, indicating that portions of antenna 810 upstream (in the drawn Figure) from L1 are trimmed and removed. Doing so will increase the resonant frequencies associated with the remaining antenna or resonator system. On the other hand, if portions of antenna 810 are clipped and removed above clipping line L2, even higher resonances will be obtained. Selective cropping or etching of portions of antenna 810 allows for fine-tuning of the characteristics of the remaining system.

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As noted, fractal elements similar to what is generically illustrated in Figures 14A-14D can be used to form a fractal vertical element in a fractal vertical antenna system, such as that described with respect to Figure 7D- 3.

Figure 15 illustrates an embodiment somewhat similar to what has been described with respect to Figure 8B or Figure 8C. Again, unit 500 is a handheld transceiver, and includes the fractal antennas 510A, 510B-510B ', 510C. It is again understood that a vertical antenna may be provided such as elements 193 and fractal counterweight 510D (shown in Figure 8b). The antennas 510B-510B 'are similar to what has been described with respect to Figures 12A-12B. The antennas 510B-510B 'are fractal antennas, not necessarily of the MI-2 configuration, as shown, and are separated by a distance Λ and, in Figure 13, are offset in the direction of rotation. Collectively, the separation distance and relative rotational displacement allow fine-tuning of the characteristics of the controlled antenna, in this case the 510B antenna. In Figure 14, antenna 510A is drawn with dotted lines to better distinguish it from separate antenna 510B. Of course, the passive conductor 510B 'could instead be a solid conductor, such as that described with respect to Figure 11A. Such a conductor can be implemented by spraying the inside surface of the cover for the adjacent antenna 510B of the unit 500 with conductive paint.

In Figure 15, the antenna 510C is similar to what has been described with respect to Figure 13A, in that a cut 830 is made in the antenna, for tuning purposes. Although antenna 510A is shown as similar to what was shown in Figure 8B, antenna 510A could, if desired, be formed on a curved substrate similar to Figures 13B or 13C. While Figure 15 shows at least two different techniques for tuning antennas in accordance with the present invention, it will be understood that a common technique could be employed instead. By that is meant that any one, or all, of the antennas 510A, 510B-510B ', 510C could include a cut-off, or be separated by a controllable distance Δ, or be rotatable with respect to a spaced conductor.

As described with respect to Figure 8B, an electronic circuit 610 may be coupled, via leads 50A, 50B, 50C, to the antennas, and sample the incoming signals in order to discern which fractal antenna, e.g. e.g. 510A, 510B-510B ', 510C (and, if present, 510D-197 antenna) is currently optimally aligned with the transmitting station, possibly a 600 or 650 or 670 unit, as shown in Figure 8A. This determination can be made by examining the signal strength of each of the antennas. An electronic circuit 620 then selects the best-oriented antenna at that time, and couples that antenna to the input of the receiver and the output of the transmitter portion, collectively 630, of the unit 500. It is understood that the selection of the best antenna is dynamic and may change depending on, for example, a 500 user walks holding the unit, or as the transmitting source moves, or due to other changing conditions. In a cell phone or wireless application, the result is more reliable communication, with the advantage that the fractal antennas can be small enough in size to fit fully within the housing of the unit 500. Also, if a flexible substrate is used, the antennas can be wrapped around portions of the inner shell, as shown. A further advantage of the embodiment of Figure 8B is that the user of unit 500 can be physically away from the antennas, at a greater distance than if a conventional external whip antenna were employed. Although the medical evidence attempting to link cancer to exposure to electromagnetic radiation from handheld transceivers is not yet conclusive, performing Figure 8B appears to minimize any similar risks.

Modifications and variations may be made to the disclosed embodiments without departing from the scope of the invention, as defined by the following claims.