Communication station, method and network for power control in spread spectrum systems

IS 2 154 247 T3

DESCRIPTION

Communication station, method and network for power control in spread spectrum systems.

Background of the invention

This invention relates to spread spectrum communications, and more particularly to a multipath processor, a variable bandwidth device, and a power control system.

Description of the corresponding technique

Spread spectrum modulation provides means for communications in which a spread spectrum signal occupies a bandwidth greater than the minimum bandwidth necessary to send the same information. Band expansion is carried out by modulating a data or information signal with a fragmentation sequence signal, which is independent of a data or information signal. The data or information signal may come from a data device, such as a computer, or from an analog device that outputs an analog signal that has been digitized into a data or information signal, such as voice or video. The fragmentation sequence signal is generated by a fragment code, in which the duration of time C_T of each chunk is substantially less than one data bit or data symbol. A synchronized reception of the information data signal with the fragmentation sequence signal at a receiver is used to de-spread the spread spectrum signal, and subsequently recover the data from the spread spectrum signal.

Spread spectrum modulation offers many advantages as a communication system for an office or urban environment. These benefits include reducing intentional or unintentional interference, combating multipath problems, and providing multiple access to communication systems shared by multiple users. Commercially, these applications include, but are not limited to, local area networks for computers, and personal communication networks for telephones, as well as other data applications.

A cellular communication network that uses spread spectrum modulation for communication between a base station and a multiplicity of users requires control of the power level of a particular mobile user station. Within a particular cell, a mobile station near the base station of the cell may be required to transmit at a lower power level than required when the mobile station is near the outer perimeter of the cell. This adjustment in power level is made to ensure that a constant power level is received at the base station from each mobile station.

In a first geographic area, such as an urban environment, the cellular architecture may have small cells, in which the respective base stations are close to each other, requiring a low power level from each mobile user. In a second geographic area, such as a rural environment, the cellular architecture may have large cells in which the respective base stations are separated from each other, requiring a relatively high power level from each mobile user. A mobile user moving from the first geographic area to the second typically adjusts the power level of his transmitter in order to cope with the requirements of that particular geographic area. If such adjustments are not made, a mobile user moving from an area sparsely populated with larger cells, using the relatively higher power level with its spread spectrum transmitter, to a densely populated area with many small cells, without reducing the original power level of your spread spectrum transmitter will produce undesirable interference within the smallest cell it has traveled to, and / or adjacent cells. Similarly, if a mobile user stands behind a building, and has his signal to the base station blocked by said building, he will have to increase his power level. These adjustments must be made quickly, with a high dynamic range and in such a way as to ensure an almost constant received power level with root mean square error and low peak deviations with respect to the constant level.

Accordingly, there is a need for a spread spectrum system and a method for automatic control of the power level of a user's mobile spread spectrum transmitter, when working in a cellular communication network.

Document EP-A-0 682 471 describes a system having a closed loop power control scheme, in which transmit power control is achieved by transmitting and receiving a power control bit . If a station receives a "1" bit, the transmit power is increased by a fixed amount of 1 dB. If the power control bit is a "0", the transmission power is decreased by an amount that depends on the number of "0" bits received successively up to a maximum of 5, varying from 1 dB to 5 dB. The document EPA-0 682 417 is not part of the state of the art according to Article 54 (2) of the EPC, but may be part of the prior art according to Article 54 (3) of said EPC.

Summary of the invention

An object of the invention is to provide an apparatus and a method that controls the power level of a station. Accordingly, there is provided a communication station according to claim 1, and a method for

ES 2 154 247 T3 adjusting the transmission power level according to claim 6. Furthermore, a cellular communication network is provided according to claim 11. Other preferred aspects of the invention are provided according to the dependent claims.

In a multipath environment, a spread spectrum signal is reflected from multiple surfaces, such as buildings, and is supposed to generate a multiplicity of spread spectrum signals. Said multiplicity of spread spectrum signals typically appears in a plurality of groups of spread spectrum signals and each group of them has a plurality of said signals. The plurality of groups of said spread spectrum signals are the result of the spread spectrum signal reflected in a multi-path environment.

A multipath processor is provided for tracking a spread spectrum signal arriving in a plurality of groups. The multipath processor includes a first plurality of correlators, a second plurality of correlators, a first adder, a second adder, and a selector device or a combiner device. The first adder is coupled between the first plurality of correlators and the selector device or the combiner device. The second adder is coupled between the second plurality of correlators and the selector device or the combiner device.

The first plurality of correlators de-spread a first plurality of spread-spectrum signals within a first group, to generate a first plurality of de-spread signals. The first adder adds or combines the first plurality of de-spread signals to generate a first combined-de-spread signal.

The second plurality of correlators de-spread a second plurality of spread-spectrum signals within a second group, to generate a second plurality of de-spread signals. The second adder adds or combines the second plurality of de-spread signals to generate a second de-spread combined signal.

The selector device selects the first combined de-spread signal or the second combined de-spread signal. The selected combined de-spread signal is output from the decision device as a de-spread output signal. Alternatively, the combiner device may combine or add the first despread-combined signal to the second combined despread signal, to generate the output despread signal.

The present invention also includes a variable bandwidth spread spectrum device for use with a spread spectrum transmitter. The variable bandwidth spread spectrum device generates a spread spectrum signal having a spread bandwidth. The variable bandwidth spread spectrum device uses a fragmentation sequence signal having a certain fragmentation rate, the rate of which is less than the spread bandwidth.

The variable bandwidth spread spectrum device includes a splicing sequence generator, spread spectrum processing means, a pulse generator, and a filter. The spread spectrum processing means is coupled to the fragmentation sequence generator. The pulse generator is coupled to the spread spectrum processing means. The filter is coupled to the pulse generator.

The fragmentation sequence generator generates a fragmentation sequence signal with a certain fragmentation rate. The spread spectrum processing means treats a data signal with the fragmentation sequence signal to generate a spread data signal. The pulse generator, which responds to each fragmentation in the expanded data signal, generates a pulse signal. The filter filters out a spectrum of each pulse signal with the expanded bandwidth.

The spread spectrum processing means may be embodied as an EXCLUSIVE OR gate, a produced device, or other device as is well known in the art of signal processing of spread spectrum data with fragmentation sequence signals. The filter may include a variable bandwidth filter. Said variable bandwidth filter can be used to vary or adjust the expanded bandwidth of the spectrum for each pulse signal. Accordingly, a spread spectrum signal having the chosen bandwidth can be designed based on the bandwidth of the chosen bandwidth filter. The bandwidth can be variable or adjustable, as required according to the requirements of a particular system. As used in this patent, a variable bandwidth is one that is capable of varying according to weather conditions or other requirements of a particular system. An adjustable bandwidth would be similar to a variable bandwidth, but is used to refer to a bandwidth that can be adjusted to stay at a chosen setting.

A system for adaptive power control (APC) of a spread spectrum transmitter is also provided. A plurality of mobile stations work in a cellular communication network with the use of spread spectrum modulation. A mobile station transmits a first spread spectrum signal. The base station transmits a second spread spectrum signal.

The base station includes automatic gain control (AGC) means, base correlator means, comparator means, power means, transmission means, and an antenna. The base correlator means is coupled to the AGC means. The power means are coupled to the correlator means

ES 2 154 247 T3 and the comparator means. The comparator means are coupled to the power means. The antenna is coupled to the transmission media.

Each mobile station includes de-spreading means and variable gain means.

A received signal is defined herein as including the first spread spectrum signal and an interference signal. The interference signal is defined herein as including noise and / or other spread spectrum signals, and / or other undesirable signals that are coexistent in frequency with the first spread spectrum signal.

For each received signal, the AGC means generates an AGC output signal. The base correlator means de-spreads the AGC output signal. The power means processes the de-spread AGC output signal and generates a received power level. The comparator means generates a power signal commanded by comparing the received power level with a threshold level. The commanded power signal may be an analog or digital data signal multiplexed with information data bits. The base station transmitting means transmits the power command signal as the second spread spectrum signal, or as a data signal multiplexed with the information data bits.

At each mobile station, the de-spreading means de-spreads the sent power signal from the second spread-spectrum signal as a power adjustment signal. The variable gain means uses the power setting signal as a basis for setting the transmitter power level of the first spread spectrum signal transmitted from the mobile station transmitter. The transmitter power level can be adjusted linearly or non-linearly.

The present invention also includes a method for automatic power control of a spread spectrum transmitter for a mobile station working in a cellular communication network with the use of spread spectrum modulation. A mobile station transmits a first spread spectrum signal. The base station performs the operations of acquiring the first spread spectrum signal transmitted from the mobile station, and detecting a received power level of the first spread spectrum signal plus any other interfering signals, including noise. The operations also include generating an AGC output signal from the received signal, and de-spreading the AGC output signal. The de-spread AGC output signal is processed to generate a received power level. The method also includes comparing the received power level with the threshold level, to generate a power command signal. Said power command signal is transmitted from the base station as part of the second spread spectrum signal.

At the mobile station, the method de-spreads the power command signal from the second spread spectrum signal, and adjusts a transmitter power level of the first spread spectrum signal in response to the power command signal.

Additional objects and advantages of the invention are set forth in the description that follows, and in part are obvious from that description, or may be appreciated from practice. The objects and advantages of the invention are also appreciated and achieved by means of the instrumentations and combinations that are particularly pointed out in the appended claims.

Brief description of the drawings

The accompanying drawings, which are incorporated in and constitute a part of the specification, illustrate preferred embodiments of the invention, and together with the description serve to explain the principles of the invention.

- fig. 1 illustrates the channel impulse response that gives rise to various multipath signals;

- fig. 2 illustrates the conditions that lead to two groups of various multipath signals;

- fig. 3 is a block diagram of a multipath processor using two sets of correlators to de-spread a received spread spectrum signal as two groups of spread spectrum signals;

- fig. 4 is a block diagram for generating delayed fragmentation sequence signals;

- fig. 5 is a gradual delay line model of a communication channel;

- fig. 6 is a block diagram of a correlator:

- fig. 7 is an autocorrelation function diagram generated from the correlator of FIG. 6;

- fig. 8 is a block diagram for tracking a received signal;

- fig. 9 is a block diagram for combining a test signal from a received spread spectrum signal;

IS 2 154 247 T3

- fig. 10 is a block diagram for tracking a test signal embedded in a pilot channel of a spread spectrum signal;

- fig. 11 illustrates a cross-correlation between a received signal and a referenced fragmentation sequence signal, as a function of referenced delay;

- fig. 12 illustrates the center of gravity of the cross-correlation function of FIG. eleven;

- fig. 13 is a block diagram of a multipath processor that uses two matching filter sets to de-spread a received spread spectrum signal as two groups of spread spectrum signals;

- fig. 14 is a block diagram of a multipath processor using three sets of correlators to de-spread a received spread spectrum signal as three groups of spread spectrum signals;

- fig. 15 is a block diagram of a multipath processor that uses three matching filters to de-spread a received spread spectrum signal as three groups of spread spectrum signals;

- fig. 16 is a block diagram of a variable bandwidth spread spectrum device;

- fig. 17 illustrates the fragmentation of an expanded data signal:

- fig. 18 illustrates the pulse signals corresponding to the expanded data signal fragmentation of FIG. 17;

- fig. 19 is an alternative block diagram of the variable bandwidth spread spectrum device of FIG. 16;

- fig. 20 is a block diagram of a base station;

- fig. 21 is a block diagram of a mobile station;

- fig. 22 illustrates a non-linear power setting;

- fig. 23 illustrates linear and non-linear power adjustment;

- fig. 24 illustrates fading during transmission of multiple signals of equivalent power, received at the base station;

- fig. 25 illustrates a broadcast power, adaptive power control signal for a fixed pitch algorithm;

- fig. 26 illustrates the de-spread output power for a fixed-step algorithm;

- fig. 27 illustrates a broadcast power adaptive power control signal, for a variable pitch algorithm; Y

- fig. 28 illustrates the de-spread output power for a variable pitch algorithm.

Detailed description of the preferred embodiments

Reference is now made in detail to the now preferred embodiments of the invention, examples of which are illustrated in the accompanying drawings, in which similar elements are indicated by the same reference numerals throughout.

Multipath processor

In a multi-path environment, a signal is reflected from various buildings or other structures. Multiple reflections from various buildings can result in multiple signals. or several groups of them, which reach the receiver. Fig. 1 illustrates a signal that arrives in time as several signals. Fig. 2 illustrates a signal arriving in time as two groups of various signals. Multiple signals arriving at the receiver generally do not arrive with uniform spread over time. Thus, in a multipath environment, a received signal r (t) can include two or more groups of spread spectrum signals.

In the multipath environment, it is assumed that a spread spectrum signal generates a plurality of groups of spread spectrum signals, and each group has a plurality of spread spectrum signals. The plurality of groups is the result of the spread spectrum signal that is reflected in a road environment

ES 2 154 247 T3 multiple. As a means of responding to and dealing with this plurality of groups, the multipath processor is an improvement over a spread spectrum receiver system.

In the arrangement example shown in fig. 3, a multipath processor is shown for tracking a spread spectrum signal. The multipath processor is used as part of a spread spectrum receiver system.

The multipath processor includes first de-spreading means, second de-spreading means, first combining means, second combining means, and output selecting or combining means. The first combination means is coupled between the first de-expansion means and the selector means or output combination signal. the second combining means is coupled between the second de-expansion means and the selection means or combining and output means.

The first de-spreading means de-spreads a received signal having a first plurality of spread-spectrum signals within a first group. The first de-spreading means thus generates a first plurality of de-spreading signals. The first combining means combines, or adds together, the first plurality of de-spread signals to generate a first combined and de-spread signal.

The second de-spreading means de-spreads the received signal having a second plurality of spread spectrum signals within a second group. The second de-spreading means a second plurality of de-spread signals. The second combining means combines, or adds together, the second plurality of de-spread signals as a second de-spread and combined signal.

The selection means selects the first combined de-spread signal or the second combined de-spread signal. The selected combined de-spread signal is output from the selection means as a de-spread output signal. The selection means may act in response to the strongest signal of the combined first despread signal and the combined second despread signal, the least root mean square error, the greatest possibility, or other selection criteria. Alternatively, with the use of the output combining means instead of the selection means, the outputs of the first combining means and the second combining means can be coherently combined or summed together, after suitable weighting.

As shown in fig. 3, the first de-spreading means may include a first plurality of correlators for de-spreading, respectively, the first plurality of spread spectrum signals. The first plurality of correlators is illustrated, by way of example, as first multiplier 111, second multiplier 112, third multiplier 113, first filter 121. second filter 122, third filter 123, first fragmentation sequence signal g (t), second fragmentation sequence signal g (tT₀). and third fragmentation sequence signal g (t-2T₀). The second fragmentation sequence signal g (tT₀) and the third fragmentation sequence signal g (t-2T₀) are the same as the first fragmentation sequence signal g (t), but delayed by a time T₀ and a 2T0 time, respectively. The delay between each fragmentation sequence signal is preferably a fixed delay

At the input the signal r (t) is received. The first multiplier 111 is coupled between the input and the first filter 121, and to a source of the first fragmentation sequence signal g (t). The second multiplier 112 is coupled between the input and the second filter 122, and to a source of the second signal g (tT₀) of fragmentation sequence. The third multiplier 113 is coupled between the input and the third filter 123, and to a source of the third signal g (t-2T₀) of fragmentation sequence. The outputs of the first filter 121, the second filter 122, and the third filter 123 are coupled to the first adder 120.

Circuits and apparatus are well known in the art for generating fragmentation sequence signals with various delays. With reference to fig. 4, a fragmentation sequence generator 401 is coupled to a controlled voltage oscillator 402 and a plurality of delay devices 403, 404, 405, 406. The controlled voltage oscillator receives a group delay signal. Said group delay signal corresponds to the time delay that the group of fragmentation sequence signals used to de-spread a particular group of received signals. The controlled voltage oscillator 402 generates an oscillator signal. The fragmentation sequence generator 401 generates the first fragmentation sequence signal g (t) from the oscillator signal, with an initial position of the first fragmentation sequence signal g (t) determined from the delay signal. group. The first fragmentation sequence signal g (t) is delayed by the plurality of delay devices 403. 404, 405, 406, to generate the second g fragmentation sequence signal (t-T0), the third g fragmentation sequence signal (t-2T₀), the fourth signal of fragmentation sequence g (t-3T_0-), etc. Thus, the second fragmentation sequence signal g (t-T0) and the third fragmentation sequence signal g (t-2T0) can be generated as delayed versions of the first fragmentation sequence signal g (t). Additionally, the acquisition and tracking circuit is part of the receiver circuit for acquiring a particular fragmentation sequence signal incorporated into a received spread spectrum signal.

Optionally, the multipath processor of FIG. 3 may include a first weighting device 131, a second weighting device 132, and a third weighting device 133. The first weighting device 131 is coupled to the output of the first filter 121, and to a source of a first weighting signal.

IS 2 154 247 T3

W₁. The second weighting device 132 is coupled to the output of the second filter 122, and to a source of the second weighting signal W₂. The third weighting device 133 is coupled to the output of the third filter 123 and to a source of the third weighting signal W₃. The first signal of weighting W₁, the second weighting signal W₂, and the third weighting signal W₃ they are optional, and may be predisposed within the first weighting device 131, the second weighting device 132, and the third weighting device 133, respectively. Alternatively, the first weighting signal W₁, the second weighting signal W2, and the third weighting signal W3, can be controlled by a processor or other control circuit. The outputs of the first filter 121, the second filter 122, and the third filter 123 are coupled through the first weighting device 131, the second weighting device 132, and the third weighting device 133, respectively, to the first adder 120. .

Similarly, the second de-spreading means may include a second plurality of correlators for de-spreading the second plurality of spread spectrum signals. Said second plurality of correlators is illustrated, as an example, as fourth multiplier 114, fifth multiplier 115, sixth multiplier 116, fourth filter 124, fifth filter 125, sixth filter 126, fourth fragmentation sequence signal g (tT_D1), fifth signal of fragmentation sequence g (tT₀-T_D1), and sixth signal of fragmentation sequence g (t-2T₀-T_D1). The fourth multiplier 114 is coupled between the input and the fourth filter 124, and a source of the fourth fragmentation sequence signal g (tT_D1). The fifth multiplier 115 is coupled between the input and the fifth filter 125 and a source of the fifth fragmentation sequence signal g (tT₀-T_D1). The sixth multiplier is coupled between the input and the sixth filter 126, and a source of the sixth fragmentation sequence signal g (t-2T₀-T_D1). The fourth fragmentation sequence signal g (tT_D1), the fifth signal of fragmentation sequence g (tT₀-T_D1), and the sixth fragmentation sequence signal g (t-2T₀-T_D1). are equal to the first fragmentation sequence signal, but delayed by time T_D1. time T₀+ T_D1, and time 2T₀+ T_D1, respectively. The second plurality of correlators thus generates the second plurality of de-spread signals. The outputs of the fourth filter 124, the fifth filter 125, and the sixth filter 126 are coupled to the second adder 130.

At the output of the fourth filter 124, the fifth filter 125, and the sixth filter 126, there may optionally be a fourth weighting device 134, a fifth weighting device 135, and a sixth weighting device 136. Said fourth weighting devices 134, fifth 135, and sixth 136, are coupled to a source that generates a fourth weighting signal W₄, a fifth weight signal W₅. and a sixth weight signal W₆, respectively. Said signals weighting fourth W₄, fifth W₅, and sixth W₆, are optional, and can be preset within the fourth 134, fifth 135, and sixth 136 weighting devices, respectively. Alternatively, the fourth weighting signals W₄. fifth W₅, and sixth W₆, can be controlled by a processor or other control circuit. The outputs of the fourth filters 124, fifth 125, and sixth 126 are coupled through the fourth weighting devices 134, fifth 135, and sixth 136, respectively, to the second adder 130. The outputs of the first adder 120 and the second adder 130 are coupled to decision device 150. Decision device 150 may be a selector or a combiner.

Weighting devices can be embodied as amplification or attenuation circuits, which change magnitude and phase. Amplification or attenuation circuits can be implemented with analog devices or with digital circuits. The amplification or attenuation circuit may be adjustable, with the gain of the amplifier or attenuation circuit controlled by the weighting signal. The use of a weighting signal with a particular weighting device is optional. A particular weighting device may be designed with a fixed value or a preset amount, such as a fixed amount of amplifier gain.

Fig. 5 is an adjustable delay line model of a communications channel. A signal s (t) that penetrates the communication channel through a plurality of retarders 411, 412, 413, 414, modeled with a time T₀. The signal s (t), for each retarder, is attenuated 416, 417, 418 by a plurality of complex attenuation factors hⁿ and adder 419. The output from adder 419 is the output from the communication channel.

A given communications channel has a frequency response that is the Fourier transform of the impulse response.

H (f) = Σ a. ei = 1

-} 2πτ i

where to. represents the complex gains of the multipaths of the communication channel, and τ. represents the multipath delays of the communication channel.

Let us consider the frequency response of the communication channel H_c (F). This response has a band of interest, B. Then, this band of interest is fixed. and the frequency response of the communication channel H_c (f) is the equivalent low pass filter function. The frequency response of the communications channel is expanded in the Fourier series as:

Hc (f) = Σ hn e ^{- B}

ES 2 154 247 T3 where h_n represents Fourier coefficients. This is an adjustable delay line model of the communication channel for which the receiver of FIG. 3 acts as a match filter when T₀ = 1 / B, and the W values_n are set for the complex conjugate of the h values_n. Namely. what w_n = h_n.

Preferably, each correlator of the first plurality of them is de-spread with a fragmentation sequence signal g (t) having a different time delay to each time delay of the fragmentation sequence signal used, respectively, with each of the other correlators of the first plurality of them. Said first plurality of correlators uses fragmentation sequence signals g (t), g (t-T0), g (t-2T0), where T0 is the time delay between the fragmentation sequence signals. The time delay T0 can be the same or different between each fragmentation sequence signal. For illustrative purposes the time delay T0 is assumed to be the same.

Likewise, each correlator of the second plurality of them de-spreads with a fragmentation sequence signal having a different time delay from each time delay of each of the other fragmentation sequence signals used, respectively, with each of the other correlators of the second plurality of them. Likewise, each correlator of said second plurality of them de-spreads with a fragmentation sequence signal having a time delay T_D1 different from each time delay of each fragmentation sequence signal used with each respective correlator of the first plurality of correlators. Therefore, the second plurality of correlators uses fragmentation sequence signals g (tT_D1), g (tT₀-T_D1), g (t-2T₀-T_D1), in which the time delay T_D1 is the time delay between the first plurality of correlations and the second plurality of them. The time delay T_D1 it is also approximately the same time delay as between the first received group of spread spectrum signals and the second received group of spread spectrum signals.

Fig. 6 illustrates a correlator in which an input signal s (t) is multiplied by multiplier 674 by a delayed version of the input signal s (tT). The product of the two signals is filtered by filter 675, and the output is the autocorrelation function R (T). Said autocorrelation function R (T) for a square wave input signal s (t) is shown in fig. 7. In a time of fragment C_T, the correlation function R (T) is maximized when points A and B are of equal amplitude. A circuit, well known in the art to perform this function is shown in FIG. 8. In said fig. 8, the de-spread signal s (t) is delayed by half a CT / 2 fragment time and is advanced by said CT / 2 fragment time. Each of the three signals is multiplied by the received signal r (t). The multiplied lead and lag signal outputs are filtered, and then amplitude sensed. The two filtered signals are combined by subtracting the delayed version from the advanced version, and the difference or error signal is used to adjust the timing of the fragmentation sequence signal used to de-spread signal s (t). Accordingly, if the delayed version were ahead of the leading, the fragmentation sequence signal for the despread signal s (t) would be delayed. Similarly, if the early version were ahead of the delayed version, then the fragmentation sequence signal to de-spread the s (t) signal would be advanced. These techniques are well known.

A similar technique is used for estimating an indicator signal from a received signal r (t), which has passed through a multi-path environment. With reference to fig. 9, the lower part of the diagram shows correlators corresponding to the correlators previously shown in FIG. 3. The upper part of the diagram shows the received signal treated by delayed versions of the fragmentation sequence indicator signal g_p(t). In fig. 9, the received signal r (t) is multiplied by the indicator signal g_p(t), and a plurality of delayed versions of the indicator signal g_p(tT₀), ..., g_p(t-kT₀) by a plurality of multipliers

661, 651, 641. The outputs of the plurality of multipliers 661, 651, 641 are filtered by a plurality of filters

662, 652,642, respectively. The outputs of the plurality of filters 662,652,642 are multiplied by a second plurality of multipliers 663, 653, 643, and are filtered respectively by a second plurality of filters 664, 654, 644. The outputs of the second plurality of filters 664, 654 , 644 are treated through a plurality of complex conjugate devices 665, 655, 645. The outputs of the plurality of complex conjugate devices 665, 655, 645 are the plurality of W values₁, W₂, W_k, respectively. The plurality of values ​​are multiplied by the output of the first plurality of filters 662, 652, 642, by a third plurality of multipliers 666, 656, 646, and then combined by the combiner 667. At the output of the combiner 667 there is a signal combined de-expansion indicator.

Each of the second plurality of indicator filters 664, 654, 644 has a bandwidth that is approximately equal to the attenuation bandwidth. This bandwidth is typically very narrow, and can be on the order of several hundred Hertz.

With reference to fig. 10, the output of the combiner 667 is multiplied by a fourth multiplier 668, and is passed through an imaginary device 669 to determine the imaginary component of the complex signal from the fourth multiplier 668. The output of the imaginary device 669 passes through a loop filter

672 to a voltage controlled oscillator 673 or a numerically controlled oscillator (NCO). The oscillator output

673 of controlled voltage passes to the fourth multiplier 668 and to each of the second plurality of multipliers 663, 653, 643.

With reference to fig. 11, the exposed circuits can generate a cross-correlation function between the received signal and a referenced fragmentation indicator signal, as a function of the referenced delay, or

ES 2 154 247 T3 phase shift. As shown in fig. 11, these transverse correlation points may have a center of gravity. The center of gravity is determined when the left mass equals the right mass from the correlation function, as shown in fig. 12. A circuit similar to that shown in fig. 8, coupled to the output of the fourth multiplier 68, can be used to align a fragmentation sequence signal from the reporter channel.

As an alternative embodiment, and as shown in fig. 13, the first de-spreading means may include a first plurality of matching filters for de-spreading the received signal r (t), having the first plurality of spread spectrum signals. At the output of the first plurality of matching filters is the first plurality of de-spread signals. Each matching filter of the first plurality of them has an impulse response h (t), h (tT₀), h (t-2T₀), etc., with a time delay T₀ displaced from the other matching filters. With reference to fig. 13, as an example, a first matching filter 141 is coupled between the input and through the first weighting device 131 to the first adder 120. A second matching filter 142 is coupled between the input and through the second weighting device 132 to the first adder 120. A third matching filter 143 is coupled between the input and through the third weighting device 133 to the first adder 120. As mentioned above, the first weighting device 131, the second weighting device 132, and the third weighting device 133 , are optional. Said first 131, second 132, and third 133 weighting devices are generally connected to a source of the first weighting signal W_B of the second weighting signal W₂, and of the third weighting signal W₃, respectively. The first plurality of matching filters generates the first plurality of de-spread signals.

Likewise, the second de-spreading means may include a second plurality of matching filters, for de-spreading the received signal r (t) having the second plurality of spread spectrum signals. Accordingly, at the output of the second plurality of matching filters is the second plurality of de-spread signals. Each matching filter of the second plurality of them has an impulse response, h (tT_D1), h (tT₀-T_D1), h (t-2T₀-T_D1), etc., with a time delay T₀ deviated from the other matching filters, and with a time delay T_D1 deviated from the first plurality of matching filters. A fourth matching filter 144 is coupled between the input and through the fourth weighting device 134 to the second adder 130. A fifth matching filter 145 is coupled between the input, and through the fifth weighting device 135 to the second adder 130. A sixth matching filter 146 is coupled between the input and through the sixth weighting device 136 to the second adder 130. As mentioned before, the fourth weighting device 134, the fifth weighting device 135, and the sixth weighting device 136 are optional. The fourth 134, fifth 135, and sixth 136 weighting devices are coupled, respectively, to a source to generate the fourth weighting signal W₄, the fifth weighting signal W₅. and the sixth weighting signal W₆. Likewise, as in the correlator embodiment, the first adder 120 and the second adder 130 are coupled to the decision device 150. This device 150 can be embodied as a selector or a combiner.

The present arrangement may also include de-spread spread spectrum signals located within a third group. Accordingly, this invention may include third de-expansion means and third combining means. The third combining means are coupled between the third de-spreading means and the selection means.

As shown in fig. 14, the third de-spreading means de-spreads the received signal r (t) as a third plurality of spread-spectrum signals within a third group. Accordingly, the third de-spreading means generates a third plurality of de-spreading signals. The third combining means combines the third plurality of de-spread signals as a third combined de-spread signal. The selecting means selects one of the combined de-spread signals, the first, the second, or the third. The output of the selector means is the de-spread output signal.

As shown in fig. 14, the third de-spreading means may include a third plurality of correlators for de-spreading the third plurality of spread spectrum signals. Said third plurality of correlators is illustrated, as an example, with seventh 117, eighth 118, and ninth 119 multipliers, seventh 127, eighth 128, and ninth 129 filters, and a source for generating the seventh g fragmentation sequence signals ( tT_D2), eighth g (tT₀-T_D2), and ninth g (t-2T₀-T_D2). The seventh multiplier 117 is coupled between the input and the seventh filter 127. The eighth multiplier 118 is coupled between the input and the eighth filter 128. The ninth multiplier 119 is coupled between the input and the ninth filter 129. The seventh multipliers 117, eighth 118, and ninth 119, are coupled to the source to generate the seventh, eighth, and ninth fragmentation sequence signals, respectively. Optionally, at the output of the seventh filters 127, eighth 128, and ninth 129, there may be, respectively, the seventh weighting devices 137, eighth 128, and ninth 139. Accordingly, the output of the seventh filter 127 is coupled through the seventh weighting device 137 to the third adder 140. The output of the eighth filter 128 is coupled through the eighth weighting device 138 to the third adder 140. The output of the ninth multiplier 129 is coupled through the ninth weighting device 139 to the third adder 140. The third adder is coupled to the decision device 150. At the output of the third plurality of correlators is the third plurality of despread signals, respectively .

Preferably, each correlator of the third plurality of correlators de-spreads with a fragmentation sequence signal g (t-TD2), g (t-T0-TD2), g (t-2T0-TD2), having a time delay T0 different from each time delay of each fragmentation sequence signal used with other correlators of the third plurality of them. Similarly, each correlator of the third plurality of correlators de-expands with

ES 2 154 247 T3 a fragmentation sequence signal having a time delay different from each time delay of each fragmentation sequence signal used, respectively, with each correlator of the second plurality of them. Similarly, each correlator of the third plurality of correlators de-spreads with a fragmentation sequence signal having a time delay of 2T_D different from each fragmentation sequence signal used with each correlator of the first plurality of them.

Alternatively, the third de-expansion means may include, as shown in FIG. 15, a third plurality of matching filters for de-spreading the third plurality of spread spectrum signals. Said third plurality of matching filters includes a seventh 147, an eighth 148, and a ninth 149 matching filter. The seventh matching filter is coupled between the input and through the seventh weighting device 137 to the third adder 140. The eighth matching filter 148 is coupled between the input and through the eighth weighting device 138 to the third adder 140. The ninth matching filter 140 is coupled between the input and through the ninth weighting device 139 to the third adder 140. The third adder 140 it is coupled to decision device 150. At the output of the third plurality of matching filters is the third plurality of de-spread signals.

The present arrangement may include fourth de-expansion means and fourth combination means, with the latter coupled between the fourth de-expansion means and the selection means. The fourth de-spreading means would de-spread a fourth plurality of spread spectrum signals within a fourth group. The output of the fourth de-spreading means would be a fourth plurality of de-spreading signals. The fourth combining means would combine the fourth plurality of de-spread signals as a combined fourth de-spread signal. The selection means selects one of the combined first, second, third, or fourth de-spread signals as the output de-spread signal.

Similarly, the fourth de-spreading means includes a fourth plurality of correlators, or a fourth plurality of matching filters, for de-spreading the fourth plurality of spread spectrum signals and generating the fourth plurality of de-spread signals. Each correlator of the fourth plurality of them de-spreads with a fragmentation sequence signal having a different time delay from each time delay of each fragmentation sequence signal used, respectively, with other correlators of the fourth plurality of them. Likewise, the fragmentation sequence signal will be different from the fragmentation sequence signals used with each correlator of the third plurality of them, from each fragmentation sequence signal used with each correlator of the second plurality of them, and from each signal of fragmentation sequence used with each correlator of the first plurality of them. Based on this discussion, a person skilled in this art will readily appreciate how to extend the concept to a fifth group of spread spectrum signals. or more generally, to a plurality of groups of spread spectrum signals.

Each of the matched filters can be obtained with the use of surface acoustic wave (SAW) devices, digital matched filter, or embodied in a specific integrated circuit (ASIC) chip application, or a digital signal processor chip ( DSP). The design of matching filters using these devices is well known in the art.

A multipath processor can single out individual paths from a group of rays. The value for each weighting device is calculated by sets of correlators, and with a reference code it is possible to follow the fragmentation sequence signal in each ray.

Alternatively, a method using a multipath processor can be employed to track a spread spectrum signal within a plurality of groups. The method comprises the steps of de-spreading the signal r (t) received as the first plurality of spread spectrum signals within a first group, to generate a first plurality of de-spread signals. The first plurality of de-spread signals are then combined as the first combined de-spread signal. The method includes de-spreading the received signal r (t) as a second plurality of spread spectrum signals within a second group, to generate a second plurality of de-spread signals. The second plurality of de-spread signals would be combined as the second combined de-spread signal. The method includes selecting either the first or the second combined de-spread signal as the output de-spread signal.

The step of de-spreading the first plurality of spread spectrum signals may include the step of coincidentally correlating or filtering the first plurality of spread spectrum signals, using a first plurality of correlators or a first plurality of matching filters, respectively. The step of de-spreading the second plurality of spread spectrum signals includes the step of matching or filtering the second plurality of spread spectrum signals with the use of a second plurality of correlators or a second plurality of match filters, respectively.

The method may also include de-spreading a third plurality of spread spectrum signals within a third group, to generate a third plurality of de-spread signals. Said third plurality of despread signals would be combined as a third combined despread signal. The selection operation would also include selecting one of the first, second, or third combined de-spread signals as the output de-spread signal. Similarly, the step of de-spreading the third plurality of spread spectrum signals may include the step of coincidentally correlating or filtering the third plurality of spectrum signals ex10.

ES 2 154 247 T3 paired with the use of a third plurality of correlators or a third plurality of matching filters, respectively.

The operation of de-spreading each of the first plurality of spread spectrum signals would include the operation of de-spreading with a fragmentation sequence signal having a different time delay than each time delay of each fragmentation sequence signal used to de-spread. other spread spectrum signals from the first plurality of these. Similarly, the operation of de-spreading each of the second plurality of spread spectrum signals would include the operation of de-spreading with a fragmentation sequence signal having a different time delay than each time delay of each sequence signal of fragmentation used to de-spread other spread spectrum signals from the second plurality of spread spectrum signals. Similarly, the operation of de-spreading each of the second plurality of spread spectrum signals would include the operation of de-spreading with a fragmentation sequence signal having a different time delay from each time delay of each fragmentation sequence signal used to de-spreading other spread spectrum signals from the first plurality of spread spectrum signals.

In the event that the method includes the operation of de-spreading a third plurality of spread spectrum signals, said method would include the operations of de-spreading with a fragmentation sequence signal that has a different time delay for each time delay of each signal. of fragmentation sequence used to de-spread other spread spectrum signals from the third plurality of spread spectrum signals. Similarly, the time delay would be different for each fragmentation sequence signal used to de-spread the spread spectrum signals of the second plurality of these, and different for each time delay of each fragmentation sequence signal used to de-spread the signals of spread spectrum of the first plurality of them.

The method can be extended to a fourth, fifth, or a plurality of groups of spread spectrum signals.

Variable bandwidth filter

The present arrangement includes a variable bandwidth spread spectrum device for use with a spread spectrum transmitter. The variable bandwidth spread spectrum device generates a spread spectrum signal having an expanded bandwidth. The term "spread bandwidth" used herein refers to the bandwidth of the transmitted spread spectrum signal. The variable bandwidth spread spectrum device uses a fragmentation sequence signal that has a fragmentation rate less than the spread bandwidth. The term "fragmentation rate" used herein indicates the bandwidth of the fragmentation sequence signal.

The variable bandwidth spread spectrum device includes first generating means, second generating means, spread spectrum processing means, and filter means. The spread spectrum processing means are coupled to the first generating means. The second generator means is coupled between the spread spectrum treatment means and the filter media.

The first generating means generates the cleavage sequence signal with the cleavage rate. The spread spectrum processing means treats a data signal with the fragmentation sequence signal, to generate a spread data signal. The second generating means generates a pulse signal in response to each fragment of the expanded data signal. The filter media filters the spectrum of each pulse signal with a bandpass equal to the expanded bandwidth.

As illustratively shown in FIG. 16, the first generating means may be embodied as a fragmentation sequence generator 161, the second generating means may be embodied as a pulse generator 165, the spread spectrum processing means may be embodied as an EXCLUSIVE-OR gate produced device 164 , or other device known to those skilled in the art for mixing a data signal with a fragmentation sequence signal, and the filter media may be embodied as a filter 166.

Produced device 164 is coupled to fragmentation sequence generator 161. Pulse generator 165 is coupled between produced device 164 and filter 166.

The cleavage sequence generator 161 generates the cleavage sequence signal with the cleavage rate. The produced device 164 processes the data signal with the fragmentation sequence signal, thereby generating an expanded data signal as shown in FIG. 17. The pulse generator 165 generates a pulse signal, as shown in FIG. 18, in response to each fragment in the expanded data signal shown in FIG. 17. Each pulse signal of FIG. 18 has a pulse bandwidth. The expression "pulse bandwidth" used herein indicates the bandwidth of the pulse signal. Although theoretically, a pulse signal has an infinite bandwidth, in practice the pulse signal has a bandwidth that is greater than the expanded bandwidth.

IS 2 154 247 T3

Filter 166 has a bandwidth adjusted to the expanded bandwidth. Thus, filter 166 filters a spectrum of each pulse signal from the expanded data signal with the expanded bandwidth. Filter 166 does this for each pulse signal.

Filter 166 preferably includes a variable bandwidth filter. Said filter can be used to vary or adjust the expanded bandwidth of the spectrum for each pulse signal. Accordingly, a spread spectrum signal can be designed having a chosen bandwidth, based on the bandwidth of the variable bandwidth filter. The bandwidth can be variable or adjustable, as required for a particular system. As used in this patent, variable bandwidth is one that is capable of varying according to weather conditions, background signals or interference, or other requirements of a particular system. An adjustable bandwidth would be similar to a variable bandwidth, but is used to refer to a bandwidth that can be adjusted to stay in a chosen arrangement.

The first generating means, as shown in fig. 19, may include a frequency domain fragmentation sequence generator 161, and an inverse Fourier transform device 162. Frequency domain splicing sequence generator 161 can be used to generate a frequency domain representation of a splicing sequence signal. The inverse Fourier transform device 162 transforms the frequency domain representation of the fragmentation sequence signal to the fragmentation sequence signal.

The first generating means may also include a memory 163 for storing the fragmentation sequence signal.

The present arrangement also includes a variable bandwidth spread spectrum method for use with a transmitter. The method includes the steps of generating the fragmentation sequence signal with the fragmentation regime and spread spectrum, treating a data signal with the fragmentation sequence signal to generate an expanded data signal. Each chunk in the spread spectrum signal is used to generate a pulse signal. Each pulse signal is filtered with the expanded bandwidth, to generate the desired bandwidth signal.

Therefore, the variable bandwidth spread spectrum device uses a lower fragmentation rate, but provides a higher bandwidth signal. The power spectral density at the output of filter 166, or of the expanded and filtered data signal s (t), is proportional to the frequency response H (f) of the filter.

PSDs (t) = k | H (f) |²

Thus, filter 166 controls the spectrum shape of the expanded and filtered data signal.

The Processing Gain (PG) is the bandwidth W of the filtered and expanded data signal, divided by the fragmentation rate R_b of the expanded and filtered data signal.

PG = W / R_b

The capacity N of the expanded and filtered data signal is

PG

Eb / N0 +1

The capacity does not depend on the fragmentation regime, but on the bandwidth. An upper limit on the capacity can be achieved if the fragmentation rate is greater than the bandwidth, but if the rate is lower, power consumption can be saved, that is, the use of a lower clock rate of CMOs, which determines power consumption.

Adaptive power control system

The present arrangement assumes that a plurality of mobile stations work in a cellular communication network through the use of spread spectrum modulation. The cellular communication network has a plurality of geographical areas, with a multiplicity of cells within each of these areas. The size of cells in a first geographical area may differ from the size of said cells in a second geographical area. In a first geographical area, such as an urban environment, the cellular architecture can have a large number of cells, each of them with a small area, and in them the corresponding base stations are located close to each other. In a second geographic area, such as a rural environment, the cellular architecture may have a smaller number of cells, each with a larger area. Furthermore, the size of cells can vary, even within a specified geographic area.

IS 2 154 247 T3

A mobile station, while in the urban environment of the first geographical area, may be required to transmit at a lower power level than when in the second geographical area of ​​rural environment. This requirement could be due to a decrease in the distance from the mobile station to the base station. Within a particular cell, a mobile station near the base station of the cell may be required to transmit at a lower power level than required when the mobile station is near an outer perimeter of the cell. This adjustment in power level is necessary to ensure that a constant power level is received at the base station from each mobile station.

The adaptive power control works by measuring the received signal-to-noise ratio (SNR) by each user, and to make the power transmitted by a user vary in a way that makes all the SNRs of the users are equal to a common value, which will be adequate for reliable communication if the total number of users and interference are less than the system capacity. Although this assumes that all users are getting the same service, eg 32 kbs voice data, a feature of the described system is that different service options are maintained for requesting users. This is done by setting the reference value for each user independently.

There are two issues that arise when it comes to the basic performance of an adaptive power control system. The first is the common value obtained from the SNR versus the load and its cost to the transmitters, in terms of transmitted power, and the second is the stability of the system. Stability refers to the fact that a disturbance of the system from its rest state produces a reaction of said system to restore said rest condition. It is highly desirable that there is only one point of rest, as otherwise "vibrations" or oscillations can occur. Stability must be dealt with with any control system, but in the present case the situation is somewhat complicated due to the fact that the users affect each other, which makes the control variables, the transmitted power and the SNRs, s resulting are dynamically coupled. Coupling is evident when one takes into account that all signals are handled by a common AGC function, which does not discriminate individual user signals from each other or from other sources.

The power control scheme of one embodiment of the present invention is a closed loop scheme. The system measures the correlator output power for each user, and compares the measured value with a set or reference value. This measured power includes both the desired signal component and the unwanted power or noise.

The AGC keeps the total power in each correlator at the preset level. This level does not vary as a function of the action of the APC; that is, this role of the AGC is independent of the APC. Furthermore, an increase in received power from any user or subset of users will be "attacked" by the AGC. This is possible because the AGC time constant is less than the APC time constant, that is, the AGC is faster than the APC. Since the total power available outside the AGC is fixed, an increase in the portion due to one user results at the expense of all other users. Although this can go against the apparent stability of the system, the AGC sensor, which measures the AGC control signal, and therefore the total received power, makes the system look for a state of rest corresponding to the minimum power received by Username. It is desired that the transmitted power be minimized because this will minimize inter-cell interference, and save battery power. Excess transmitter power will dissipate within the AGC if all users transmit excessive power.

The implementation shown in the figures is to be considered representative. In particular, the method of remote transmitter power control via variable gain attenuators and amplifiers is perhaps redundant. One or both of these means may be employed, depending on the required (dynamic) control field. Likewise, the control can be carried out on any of the IF or RF frequencies.

For the purposes of discussion, a mobile station within a particular cell transmits a first spread spectrum signal, and the base station transmits a second spread spectrum signal.

In the example arrangement shown in FIG. 20, a base station block diagram is provided as part of a system for adaptive power control of a spread spectrum transmitter.

Fig. 20 illustrates the base station adaptive power control system, with automatic gain control (AGC) means, power means, comparator means, transmitting means, and an antenna. The AGC means is shown as an automatic gain control (AGC) amplifier 228, the correlator means is shown as a de-expander 231, and the power means is shown as a power meter device 233. The comparator means is shown as a comparator 239, and the transmitting means is shown as a power amplifier 237 coupled to the antenna 226. Also illustrated is a delta modulator 235 coupled between the comparator 239 and the power amplifier 237.

The AGC amplifier 228 is coupled to the de-expander 231. The power measurement device 233 is coupled to said de-expander 231. The comparator 239 is coupled to the output of the power measurement device 233 and the AGC amplifier 228. Multiplexer 234 is coupled between comparator 239 and power amplifier 237. Delta modulator 235 is coupled between power amplifier 237 and multiplexer 234. Said power amplifier 237 is coupled to antenna 56.

IS 2 154 247 T3

A threshold level is used by comparator 239 as a comparison to the received power level measured by power meter device 233.

For each received signal, the AGC amplifier 228 generates an AGC output signal and an AGC control signal. The AGC output signal is de-spread to obtain the signal from a first user using the de-spreader 231. The de-spread AGC output signal from the de-expander 231 is combined with the AGC control signal from the AGC amplifier 228, by middle of combiner 241. The AGC control signal from AGC amplifier 228 can be shifted one level of shift S1 with the use of combiner 242, and weighted by weighting device 243. Such weighting device 243 can be an amplifier or attenuator.

The received power level from the power device 233 can be shifted by an offset level S2 by using the combiner 244, and weighted by the weighting device 245. The weighting device 245 can be an amplifier or an attenuator. Combiner 241 combines the AGC control signal with the received level signal to generate an adjusted received power level. Comparator 239 generates a comparison signal by comparing the adjusted received power level with the threshold level. The comparison signal can be an analog or digital data signal. The comparison signal indicates whether the mobile station is to increase or decrease its power level. If the adjusted received power level is for example higher than the threshold, the comparison signal then sends a message to the mobile station to decrease its transmitter power. If the adjusted received power level is below said threshold, the comparison signal then sends a message to the mobile station to increase its transmitter power. The comparison signal is converted to a commanded power signal by means of the delta modulator 235.

The commanded power signal can be transmitted with the second spread spectrum signal, or separated from it. For example, a spread spectrum signal that employs a first fragment sequence can be considered a first spread spectrum channel, and a signal that uses a second fragment sequence can be considered a second spread spectrum channel. The power command signal can be transmitted on the same spread spectrum channel, that is, on the first spread spectrum channel, as the second spread spectrum signal, in which case the power command signal is transmitted in an interval different from that in which the second spread spectrum signal is transmitted. This format allows the mobile station to achieve synchronization with the first sequence, with the use of the second spread spectrum signal. The power command signal can also be transmitted on a second spread spectrum channel, different from the second spread spectrum signal. In this case, the second spread spectrum signal having the power command signal will be acquired by the second splicing sequence generator, and the second device produced. The power command signal may be time division multiplexed or frequency division multiplexed with the second spread spectrum signal.

The base correlating means is shown in fig. 20 as the first de-spreader 231. As an example, the system may have the base correlator means embodied as a produced device, a fragmentation sequence generator, and a band pass filter. Alternatively, the base correlator means may be embodied as a matching filter, such as a surface acoustic wave device, or as a matched digital filter, embodied in a digital signal processor. In general, the base correlator means use or are adapted to the fragmentation sequence of the spread spectrum signal that is being received. Matching correlators and filters for de-spreading a spread spectrum signal are well known in the art.

Typically, the AGC circuit 228 is coupled to a low noise amplifier 227, through an isolator 225 to the antenna 226. In FIG. 20, a plurality of de-spreaders, de-spreader 229 to de-spreader 231, are shown to de-spread a plurality of spread spectrum channels, which can be received from a plurality of mobile stations. Likewise, the output of each de-spreader 229 through de-spreader 231 is coupled to a plurality of demodulators, illustrated as demodulator 230 to demodulator 232, respectively, to demodulate data from the de-spread AGC output signal. Accordingly, a plurality of data outputs are available at the base station.

For a particular spread spectrum channel, first de-spreader 231 is shown coupled to power device 233 and multiplexer 234. Said power device 233 is typically a power measurement circuit that treats the de-spread AGC output signal as a power level. the received power. Power device 233 could include an analog-to-digital converter circuit to output a digital level of received power. The comparator means, embodied as the comparator circuit 239, compares the received and processed power level with a threshold. Multiplexer 234 is coupled to the output of power device 233 through comparator circuit 239. Multiplexer 234 can insert appropriate framing bits, as required.

The transmitting means can be embodied as a Quadrature Phase Deviation Keying (QPSK) modulator or a delta modulator 235 coupled to a power amplifier 237. In FIG. 20, the input to the delta modulator 235 will typically have the power command signal from the power device 233 multiplied by the data from the nth channel k. A plurality of spread spectrum channels will have their data and the appropriate power order signals combined by combiner 236 and amplified by power amplifier 237. The output of power amplifier 237 is coupled to antenna 226 through isolator 125 .

IS 2 154 247 T3

The power command signal is transmitted periodically. The period T could be chosen to be 250 microseconds, in order to ensure a low mean square error as well as a low peak error between the received instantaneous signals and the desired constant signal.

A mobile station is illustratively shown in FIG. 21. The movable de-spreading means is illustrated as the de-spreader 334, and the variable gain means is illustrated as a variable gain device 341. Variable gain device 341 is coupled between transmitter 342 and through isolator 336 to antenna 335. De-expander 334 is coupled to isolator 336 and demultiplexer 339. The output of de-spreader 334 is also coupled to a demodulator 340. De-spreader 334 can be embodied as an appropriate correlator or coincidence filter, for de-spreading of the nth channel k. Additional circuitry may be used, such as radio frequency (RF) amplifiers and filters, or intermediate frequency (IF) amplifiers and filters, as is well known in the art.

A second spread spectrum signal received at antenna 335 passes through isolator 336 to de-spreader 334. This de-spreader 334 is matched to the desired spread spectrum channel chipping sequence. The output of de-spreader 334 passes through demodulator 340 to demodulate data from the desired spread spectrum channel. Additionally, demultiplexer 339 demultiplexes the power command signal from the de-spread signal exiting de-spreader 334. The power command signal drives variable gain device 341.

A decision device 345 and an accumulator 346 may be coupled between the demutiplexer 339 and the variable gain device 341. A stepper algorithm device 344 is coupled to the output of decision device 345 and accumulator 346.

The progressive step algorithm device 344 stores an algorithm for adjusting the power level of the variable gain device 341. A non-linear algorithm that could be used is shown in fig. 22. fig. 23 compares a nonlinear algorithm with a progressive step linear algorithm.

The power command signal from demultiplexer 339 causes decision device 345 to increase or decrease the power level of variable gain device 341, based on the threshold of the forward-stepping algorithm shown in FIG. 23. The accumulator follows the previous power levels as a means to determine the necessary adjustments in the progressive step of the power level and to proceed with the algorithm as shown in FIG. 2. 3.

Variable gain device 341 can be embodied as a variable gain amplifier, variable gain attenuator, or any device that performs the same function as variable gain device 341 described herein. Said device 341 increases or decreases the power level of the remote station transmitter, based on the power command signal.

As illustratively shown in FIG. 20, a block diagram of a power meter circuit includes an interference eliminator for use with the base station. As shown in said fig. 20, the AGC amplifier 228 is connected to the de-expander 231, and the output of the latter is connected to the power measurement circuit 233. Additionally, the AGC amplifier 228 is connected to the combiner 236 through the comparator 239.

A received signal includes a first spread spectrum signal with a power P_C, and the other input signals that are considered to be interference signals with power Pj, at the input to the AGC amplifier 228 of FIG. 20. The interference signal can come from one or more undesirable signals, noise, multipath signals, and any other source that serves as the interference signal for the first spread spectrum signal. The received signal is normalized by means of the AGC amplifier 228. Therefore and as an example, the AGC amplifier 228 may have a power output, P_C + P_J = 1. The received and normalized signal is de-spread by de-spreader 231, to receive a particular mobile user signal. The clearer sequence generator 231 generates a fragment sequence signal, using the same cleavage sequence employed by the first spread spectrum signal. Alternatively, the matched filter, if used, of the despread 231 may have an impulse response matched to the same clipping sequence used by the first spread spectrum signal. The output of the spreader 231 is the normalized power of the first spread spectrum signal plus the normalized power of the interface signal divided by the treatment gain, PG, of the spread spectrum system. Power measurement circuit 233 generates a received power level from the first spread spectrum signal. Comparator 239 processes the received despread signal with the AGC control signal, and outputs the power control signal of the first spread spectrum signal. The power level of the interfering signal is reduced by the treatment gain, PG.

Comparator 239 processes the AGC control signal with the received, normalized, and de-spread signal by multiplying the two signals together, or by logarithmic processing of the AGC control signal with the received signal de-spread. In the latter case, the logarithm of the received signal power is taken, PC + PJ, and the logarithm of the received signal, normalized and de-spread, is taken. The two logarithms are added to produce the received power level.

IS 2 154 247 T3

For the present invention to work effectively, the despread signal must be kept nearly constant, and independent of variations in other signals or obstructions. A preferred implementation for accomplishing this is shown in the circuit of FIG. 20. FIG. 20 shows a means for determining at the base station the power of the first spread spectrum signal, when the received signal includes multiple signals and noise. If the circuit of fig. 20 was not used, then it is possible that the interfering signal, which can include noise, multipath signals, and other undesirable signals, could raise the power level measured at the input of the base station receiver, thereby the first spread spectrum signal is suppressed. The measured undesirable power level can cause the remote station to transmit more power than required, which increases the amount of power received at the base station.

As mentioned before, the APC system is closed-loop. The APC loop works by command generation to increase or decrease the transmitter power at the updated speed. This is actually a quantization procedure, which is done to limit the amount of information that must be returned to the remote transmitter. The amount of the increase or decrease can be set in advance, or it can be adapted in response to the characteristics of the channel, measured locally at the remote terminal, which is controlled. In particular, the remote terminal can examine the sequence of the orders received by it. For example, a long sequence of increase commands implies that the progressive step can be increased. A typical scheme increases the progressive step by a fixed amount or a fixed percentage, as long as two successive bits are equal. For example, the progressive step can be increased by 50% if two bits in a row are the same, and decreased by 50% if they are different. This constitutes a fairly rough change in step forward, and is intended to be adaptable to local or immediate variations in time in the required transmitted power. This procedure results in a large variation in uptime over time.

An adaptive phasing algorithm can also be considered in a different context. Specifically, the progressive step can be considered almost constant, or that it does not respond to localized variations in the transmitted power demanded, but the value can be adjusted automatically based on the global characteristics of the induced control action of the channel. Therefore, in an almost static environment, a small constant progressive step should be used, while in a mobile environment, the progressive step should be greater.

The remote station transmitter power level adjustment can be performed both linearly and non-linearly. The following algorithm will cause the progressive step to be set at a constant value close to optimal. The receiver examines successive APC bits and increases the forward step by the factor (1 + x) if they agree, and decreases the forward step by the factor (1 + x) if they are not. Here, the parameter x is small (for example, x = 0.01). Although this procedure does not allow local adaptation (because x is small), it will result in adaptation to global conditions. Specifically, if the transmitted APC bit stream shows a trend toward agreement in successive bits (i.e., passes of 1s or 0s are evident) this implies that the system is not tracking changes in channel conditions ( that is, the system is slow speed limited), and the progressive step must be increased. On the other hand, if successive bits tend to be opposite, the system is trying to “hunt” a value between two excessively far apart values. According to statistics, what is expected to be optimal are intermediate values ​​between these extremes. That is, the APC bit stream is likely to also contain the patterns (0, 0), (0, 1), (1, 0), and (1, 1) in any pair of successive bits. The above algorithm drives the behavior of the system towards this.

The above algorithm (global adaptation) works particularly well when the system employs a wide update rate relative to the channel dynamics.

As illustrated in fig. 23, for example, to increase the power level with the use of linear adjustment, the transmitter power is increased in regular increments of one volt, or other unit, as instructed by the base station, until the received power level at said base station is strong enough. Linear adjustment can be time consuming if the required power adjustment is substantial.

As shown in fig. 22, to increase power by using non-linear adjustment, the transmitter voltage can be increased, for example, geometrically until the transmitted power is higher than the desired level. The transmitter power can then be geometrically reduced until the transmitted power is below the desired level. A preferred method is to increase the step voltage by a factor of 1.5, and decrease the step voltage by a factor of 0.5. Other non-linear algorithms can be used. As shown in fig. 23, this procedure is repeated with a reduction in the error margins both in excess and in insufficient power, until the desired signal level is obtained. Non-linear adjustment provides significantly faster raise and lower time than linear adjustment, and may be preferable if power is to be adjusted significantly.

The system determines the error status (APC bit) every T sections, with 1 / T being the update rate of the control. This update rate can vary from 100 Hz, which is a low value, to 100 kHz, which is quite high. The opportunity to measure the error state of the system arises with each receipt of a new symbol. Therefore, the update rate can be equal to that of the symbol. If this update rate is not sustained, it is beneficial to make use of the error measurements available by combining them (or averaging them) between updates. This minimizes the possibility of power trimming in the wrong direction, which can occur due to noise in the error signals themselves.

IS 2 154 247 T3

The choice of an update rate depends on factors other than the operation of APC, namely, the amount of capacity and the method of allocating said capacity to the transport of the APC bits over the channel. In general, a faster update will produce a better performance, even if the increased update rate is obtained by allowing the APC bits to be occasionally received in error. Working at a 1 kHz update rate without channel induced errors will be less effective than at a 100 kHz update rate with a 25% error rate. This is due to the self-correcting behavior of the control loop. A faster update rate eliminates control time-out, which is a key performance limiting phenomenon.

A spread spectrum base station receives all incoming signals simultaneously. Therefore, if a signal were received with a higher power level than others, then that signal receiver has a higher signal-to-noise ratio, and therefore a lower bit error ratio. The base station ensures that each mobile station transmits with the correct power level, informing the remote station every 500 microseconds whether to increase or decrease the power of the mobile station.

Fig. 24 shows a typical fading signal that is received at the base station along with ten other independent fading and thermal noise signals having the same power as one of the signals. Note that the duration of the fade is approximately 5 milliseconds, which corresponds to a vehicle speed greater than 96 km per hour. Figs. 25 and 26 illustrate the results obtained when using a particular adaptive power control algorithm. In this case, whenever the power of the received signal changes, the base station informs the remote station, and the remote station changes its power by ± 1 dB. Fig. 25 shows the adaptive power control signal at the remote station. Fig. 26 shows the received power at the base station. Note that adaptive power control follows deep fades, resulting in 9 dB fades. This reduced power level results in a 1.4 x 10 bit error rate.².

For the same fading of fig. 24, it is assumed that a different adaptive power control algorithm is employed, as shown in Figs. 27 and 28. In this case the control voltage results in the remote unit changing its power by a factor of 1.5 in the same direction, or by a factor of 0.5 in the opposite direction. In this particular implementation, the minimum forward step was 0.25 dB, and the maximum forward step was 4 dB. Note that the error is normally limited to ± 2 dB, and occasional decreases in power from 5 dB to 6 dB result in a BER »8 x 10^-4, a significant improvement compared to the previous algorithm. The use of leading and interleaved error correction codes can generally correct any errors that result in the rarely observed power drops.

During operation, a mobile station in a cell may transmit the first spread spectrum signal on a continuous basis, or on a repetitive periodic basis. The base station inside the cell receives the first spread spectrum signal. Said first received spread spectrum signal is acquired and de-spread with the fragmentation sequence signal from the fragmentation sequence generator and the produced device. The first de-spread spread spectrum signal is filtered through a band pass filter. The base station detects the first de-spread spread spectrum signal with the use of an envelope detector, and measures or determines the received power level of the first spread spectrum signal. The base station generates the power command signal from the received power level.

The present arrangement also includes a method for the automatic power control of a spread spectrum transmitter, for a mobile station working in a cellular communication network using spread spectrum modulation, whose mobile station transmits a first spread spectrum signal. In use, the method includes the operation of receiving a signal, generating the AGC output signal, de-spreading the AGC output signal, processing the de-expanded AGC output signal to generate a received power level, generating a signal of power order, transmit the power order signal as the second spread spectrum signal, de-spread the power order signal from the second spread spectrum signal as the power setting signal, and adjusting a power level of the first spread spectrum signal.

The received signal includes the first spread spectrum signal and the interference signal, and is received at the base station. The AGC output signal is generated at the base station and is de-spread as the AGC output signal de-spread. Said de-spread AGC output signal is processed at the base station to generate a received power level.

The received power level is compared to a threshold, and the comparison is used to generate a power command signal. If the received power level were greater than the threshold, the power command signal would command the mobile station to reduce the transmitter power. If the received power level were less than the threshold, the power command signal would command the mobile station to increase the transmitter power.

The power command signal is transmitted from the base station to the mobile station as a second spread spectrum signal. In response to reception of the second spread spectrum signal, the mobile station de-spreads the power command signal as a power adjustment signal. Depending on whether the power command signal ordered the mobile station to increase or decrease the power of the transmitter, said mobile station, in response to the power adjustment signal, increases or decreases respectively the power level of the transmitter of the first spread spectrum signal.

IS 2 154 247 T3

The method may further include generating from a received signal, an AGC output signal, and de-spreading said AGC output signal. The received signal includes the first spread spectrum signal and an interference signal. The received signal is treated with the AGC output signal de-spread, to generate the received power level. The method then generates a comparison signal by comparing the received power level and the threshold level. While transmitting a second spread spectrum signal, the method adjusts the transmitter power level of the first spread spectrum signal from the transmitter with the use of the power adjustment signal.

Those skilled in the art will appreciate that various modifications may be made to the spread spectrum system and method of the present invention, and the present invention is intended to cover such modifications and variations of said spread spectrum method and system.