Radar sensor comprising two oscillators, two i/q transmit mixers and two i/q-receive mixers

State of the art

The invention relates to a radar sensor with a transmission part, which has two oscillators and a 90 ° phase shifter for generating a transmission signal, a first comparison signal and a second comparison signal which is 90 ° out of phase with the first comparison signal, and a receiving part with an I mixer for mixing a received signal with the first comparison signal and a Q mixer for mixing the received signal with the second comparison signal, wherein that the transmitter part has a first transmitter mixer, the inputs of which are connected directly to the two oscillators.

Driver assistance systems are increasingly being used in motor vehicles in which a radar sensor is used to locate objects in the vicinity of the vehicle. Depending on the functional principle and purpose of the radar sensor, it may be necessary to evaluate not only the amplitude but also the phase of the received radar signals. For example, for an accurate distance measurement in the close range with the help of a CW radar (Continuous Wave) a precise knowledge of the phase of the received signal is required. In the case of an angle-resolving radar sensor, for example an FMCW radar (Frequency Modulated Continuous Wave) with several transmission and reception channels, an evaluation of the phase differences between the different channels allows a more precise angle determination. With a radar sensor of the type mentioned above, the phase can be determined from the ratio of the I and Q signals supplied by the two mixers of the receiving part.

For example, this sensor can be a heterodyne sensor in which one oscillator is used to generate the transmission signal and the other oscillator is used to generate the comparison signals. If an oscillator with a fixed frequency is used to generate the comparison signals, a simple and precise generation of the 90 ° phase shift can be achieved. A disadvantage of this type of sensor, however, is that the phase noise of the two oscillators is not correlated, so that relatively complex, low-phase oscillators are required.

In contrast to this, in the case of a homodyne sensor concept, the transmission signal and the two comparison signals are generated by one and the same oscillator, so that the transmission and comparison signals have a correlated phase noise, which makes it easier to suppress the effects caused by the phase noise when evaluating the received signals. Here, however, there is the disadvantage that it cannot be checked, or only with great effort, whether and to what extent the phase shift produced by the phase shifter deviates from the ideal value of 90 °. Such deviations lead to phase errors in the received signal which cannot be determined or corrected, or only with great effort.

Out <patcit id="pcit0001" dnum="WO2005098471A2"><text>WO 2005/098471 A2</text></patcit> a radar sensor according to the preamble of claim 1 is known.

<patcit id="pcit0002" dnum="US7012561B2"><text>US 7 012 561 B2</text></patcit> describes a radar sensor based on the heterodyne principle, with a second oscillator, which is connected together with the first oscillator to a single transmitter mixer, with no phase shifter being interposed.

Disclosure of the invention

The object of the invention is to provide a radar sensor which enables a simpler and more precise determination of the phase of the received signal.

This object is achieved in that the transmitter part has a second transmitter mixer, one input of which is connected directly to a first of the two oscillators and the other input of which is connected to the other oscillator via the phase shifter.

Advantages of the invention

The two transmitter mixers can be used to generate low-frequency 1 and Q reference signals, which can be used to monitor and regulate the phase shifter and / or to reduce the phase noise of one or both oscillators. In addition, in the case of a sensor with a frequency-modulated transmission signal, these reference signals allow simple and precise control of the frequency modulation without falsification of the control loop by additional frequency divider chains.

In particular, this enables an accurate measurement at close range. The radar sensor is therefore also suitable, for example, for measuring the tank level in a motor vehicle or for applications outside of motor vehicles, for example for distance measurements in industrial technology.

Advantageous refinements and developments of the invention are specified in the subclaims.

In the case of a homodyne radar, the transmission signal and the first comparison signal can be formed by the sum frequency component of the mixed product of the first transmission mixer (i.e. the frequency component of the mixed product whose frequency is equal to the sum of the frequencies of the two oscillators), and the second comparison signal can be formed by the sum frequency component of the mixed product of the second transmitter mixer. The difference frequency components of the two transmitter mixers then form the I and Q reference signals. The function of the phase shifter can then be monitored and, if necessary, regulated by means of a phase comparison of these reference signals.

Alternatively or additionally, it is possible to use an oscillator with a variable frequency for the oscillator whose signal is fed to the phase shifter, the frequency of which is linked to a fixed reference frequency. Since the frequency of this oscillator is precisely known and the difference frequency of the two oscillators is given by the frequencies or I and Q reference signals, the frequency (sum frequency) of the transmission signal generated by the first transmission mixer can also be determined precisely.

Furthermore, the frequency of the I or Q reference signal can be regulated in a phase-locked loop, which drives the associated oscillator, to a modulation frequency, which then determines the frequency modulation of the transmission signal.

In the case of a hetorodyne radar, the first oscillator can be used to generate the transmission signal, while the second oscillator generates the two comparison signals. The two transmitter mixers then generate I and Q reference signals, the frequency of which is lower, the smaller the frequency difference between the two oscillators and which are preferably in an intermediate frequency band. The I and Q signals which are obtained as mixed products of the 1 and Q mixers then also lie in the same intermediate frequency band, and the phase noise of these signals is correlated with the phase noise of the reference signals. By mixing or I and Q reference signals with the intermediate frequency signals I and Q, corresponding signals are then obtained in the baseband, from which the phase of the received signal can be determined with high accuracy.

In <patcit id="pcit0003" dnum="DE102004052518A1"><text>DE 10 2004 052 518 A1</text></patcit> and <patcit id="pcit0004" dnum="WO2008006656A1"><text>WO 2008/006656 A1</text></patcit> describes angle-resolving (homodyne) radar sensors whose multiple transmission and reception channels each have a specific frequency offset. This suppresses interference effects caused by interference between the different channels and also enables the evaluation of so-called cross echoes, ie the evaluation of radar signals that were transmitted in one channel and received in another channel after being reflected on the object.

The radar sensor proposed here, in particular in the homodyne configuration, allows simple adjustment of the frequency offsets between the different channels and simple and uniform control of the frequency modulation in these channels.

Brief description of the drawings

Embodiments of the invention are shown in the drawings and explained in more detail in the following description.

Show it:<dl id="dl0001"><dt>Figure 1</dt><dd>a block diagram of a radar sensor according to the invention according to a first embodiment;</dd><dt>Figure 2</dt><dd>a block diagram of a development of the radar sensor <figref idref="f0001">Fig. 1</figref>;</dd><dt>Figure 3</dt><dd>a block diagram for another variant of the radar sensor <figref idref="f0001">Fig. 1</figref>;</dd><dt>Figure 4</dt><dd>a block diagram of a radar sensor according to a second embodiment; and</dd></dl><dl id="dl0002" compact="compact"><dt>Figure 5</dt><dd>a block diagram of a radar sensor with two transmit and receive channels.</dd></dl>

Embodiments of the invention

The in <figref idref="f0001">Fig. 1</figref> The radar sensor shown has a transmitting part 10 and a receiving part 12, which can be installed on a common circuit board 14. The transmission part 10 has a first oscillator 16 and a second oscillator 19, the oscillation frequency of which corresponds in each case to approximately half of the desired transmission frequency. For example, at a transmission frequency of 122 GHz, the first oscillator 16 has a frequency f₁ of 62 GHz and the second oscillator 18 a frequency f₂ of 60 GHz. The signals of the two oscillators 16, 18 are mixed with one another in a first transmission mixer 20. A signal i (t) of the form is thus obtained at the output of the first transmitter mixer 20<maths id="math0001" num=""><math display="block"><mi mathvariant="normal">i</mi><mfenced><mi mathvariant="normal">t</mi></mfenced><mo mathvariant="normal">=</mo><mi>cos</mi><mfenced open="[" close="]" separators=""><mn mathvariant="normal">2</mn><mo>⁢</mo><mi mathvariant="normal">π</mi><mo>⁢</mo><mfenced separators=""><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">1</mn></msub><mo mathvariant="normal">-</mo><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">2</mn></msub></mfenced><mo>⁢</mo><mi mathvariant="normal">t</mi></mfenced><mo mathvariant="normal">+</mo><mi>cos</mi><mfenced open="[" close="]" separators=""><mn mathvariant="normal">2</mn><mo>⁢</mo><mi mathvariant="normal">π</mi><mo>⁢</mo><mfenced separators=""><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">1</mn></msub><mo>+</mo><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">2</mn></msub></mfenced><mo>⁢</mo><mi mathvariant="normal">t</mi></mfenced><mn mathvariant="normal">.</mn></math><img file="EP2483706B1_D0001.tif" /></maths>

In a duplex filter 22, the frequency components with the sum frequency (f₁ + f₂) and the difference frequency (f₁ - f₂) separated from each other. At the output of a high-pass path of the duplex filter 22, a transmission signal S (with the sum frequency (f₁ + f₂), which is emitted via an antenna 24.

The transmitting part 10 also contains a phase shifter 26, with which the phase of the signal generated by the second oscillator 18 is shifted by 90 °. In a second transmitter mixer 28, this phase-shifted signal is mixed with the signal of the first oscillator 16, and a signal q (t) of the form is obtained at the output<maths id="math0002" num=""><math display="block"><mi mathvariant="normal">q</mi><mfenced><mi mathvariant="normal">t</mi></mfenced><mo mathvariant="normal">=</mo><mi>sin</mi><mfenced open="[" close="]" separators=""><mn mathvariant="normal">2</mn><mo>⁢</mo><mi mathvariant="normal">π</mi><mo>⁢</mo><mfenced separators=""><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">1</mn></msub><mo mathvariant="normal">-</mo><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">2</mn></msub></mfenced><mo>⁢</mo><mi mathvariant="normal">t</mi></mfenced><mo mathvariant="normal">+</mo><mi>sin</mi><mfenced open="[" close="]" separators=""><mn mathvariant="normal">2</mn><mo>⁢</mo><mi mathvariant="normal">π</mi><mo>⁢</mo><mfenced separators=""><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">1</mn></msub><mo>+</mo><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">2</mn></msub></mfenced><mo>⁢</mo><mi mathvariant="normal">t</mi></mfenced><mn mathvariant="normal">.</mn></math><img file="EP2483706B1_D0002.tif" /></maths>

In a further duplex filter 30, the signal q (t) is also converted into a frequency component with the sum frequency (f₁ + f₂) and a frequency component with the difference frequency (f₁ - f₂) split.

The receiving part 12 contains an I mixer 32, a Q mixer 34 and a receiving amplifier 36, the input of which is connected to an antenna 38. The antenna 38 can optionally be identical to the antenna 24 (monostatic antenna concept) if the received signal is separated from the transmission signal using a circulator, for example.

The radar radiation emitted by the antenna 24 is reflected by an object 40, and the radar echo generated in this way is received by the antenna 38 and, after amplification in the amplifier 36, is fed as a received signal E to the I and Q mixers 32 and 34. The received signal E has a frequency shift compared to the transmission signal S, which is dependent on the relative speed of the object 40 (Doppler effect) and, if the transmission signal is frequency-modulated, also on the signal transit time and thus the distance of the object 40.

In the I mixer 32, the received signal E is compared with a first comparison signal V₁ mixed, which is tapped (for example with the aid of a coupler, not shown in detail) from the high-pass output of the duplex mixer 22, that is to say is identical to the transmission signal S (homodyne mixing concept). An I signal I is thus obtained at the output of the I mixer 32_BB, whose frequency is in a so-called baseband and is equal to the frequency difference between the received signal E and the comparison signal V₁ is. The frequency of this I signal thus indicates the frequency shift of the radar echo. In the Q mixer 34, the received signal E is compared with a second comparison signal V₂ mixed, which is supplied from the high-pass output of the duplex mixer 32. This comparison signal V₂ thus has the sum frequency f₁ + f₂However, due to the effect of the phase shifter 26 compared to the first comparison signal V₁ 90 ° out of phase. A Q signal Q is thus obtained at the output of the Q mixer_BB, which is also in the baseband and has the same frequency as the I signal I_BB. In a downstream

The evaluation stage can now determine the phase ϕ of the radar echo from the I and Q signals: <maths id="math0003" num=""><math display="block"><mi mathvariant="normal">ρ</mi><mo mathvariant="normal">=</mo><mi>arctan</mi><mfenced separators=""><msub><mi mathvariant="normal">I.</mi><mi>BB</mi></msub><mo mathvariant="normal">/</mo><msub><mi mathvariant="normal">Q</mi><mi>BB</mi></msub></mfenced><mn mathvariant="normal">.</mn></math><img file="EP2483706B1_D0003.tif" /></maths>

The oscillators 16, 18 inevitably have a certain phase noise. However, since the comparison signals V₁ and V₂ are formed in the same way as mixed products from the signal of both oscillators, the phase noise of the second comparison signal V₂ with the phase noise of the first comparison signal V₁ correlated so that this phase noise in the further evaluation of the I and Q signals I_BB, Q_BB is not a significant source of interference.

A certain phase error can, however, result from the fact that the phase shift generated by the phase shifter 26 is not exactly 90 °. However, the radar sensor proposed here offers the possibility of determining this phase error and correcting it if necessary. For this purpose, an I reference signal I is at the output of the low-pass part of the duplex mixer 22_ref formed, the frequency of which is equal to the difference frequency f₁ - f₂ is. Accordingly, a Q reference signal Q is output at the low-pass portion of the duplex filter 30_ref formed, which is also the difference frequency f₁ - f₂ Has. The phase difference between the I and Q reference signals represents the phase error generated by the phase shifter 26. Consequently, these reference signals can be used to determine and correct the phase error and / or to further suppress the phase noise. Various options for this are described below using the<figref idref="f0002">Fig. 2</figref> and <figref idref="f0003">3</figref> illustrated.

In <figref idref="f0002">Fig. 2</figref> A phase-locked loop 40 is first shown, which has a phase comparator 42 and a downstream low-pass filter 44. The phase comparator 42 compares the phases of the I and Q reference signals I._ref, Q_ref. After low-pass filtering in the low-pass filter 44, the comparison result is fed to the phase shifter 26 as a control signal. In this way, the phase shifter 26 can be continuously controlled so that deviations in the phase shift from the ideal value of 90 ° are suppressed. It is essential that the I and Q reference signals have a comparatively low frequency (2 GHz in this example), so that their phases can be determined and compared with sufficient accuracy.

Furthermore, in <figref idref="f0002">Fig. 2</figref> a frequency control loop 46 is shown, which is also formed by a phase comparator 48 and a downstream low-pass filter 50. A reference frequency f_ref supplied, which is generated by a frequency-stable reference oscillator and 1/32 of the desired frequency f₂ of the second oscillator 18. The other input of the phase comparator 48 is supplied with a signal which is formed from the output signal of the second oscillator 18 by its frequency f₂ is divided using a frequency divider 52 in a ratio of 1/32. In this case, the second oscillator 18 is a frequency-controllable oscillator, to which the comparison result after low-pass filtering in the low-pass filter 50 is fed as a control signal. In this way, the frequency f₂ of the second oscillator 18 continuously controlled so that they are exactly 32 times the reference frequency f_ref is.

A major advantage here is that the frequency f₂ of the second oscillator 18 only about half the transmission frequency f₁ + f₂ is so that a frequency division in a ratio of 1/32 is sufficient to obtain a sufficiently low frequency for an accurate phase comparison in the phase comparator 48. With known frequency f₁ of the first oscillator 16, the transmission frequency f₁ + f₂ determine with high accuracy. If necessary, the difference frequency f₁ - f₂ be monitored using the I or Q reference signal to f₁ based on the difference frequency f₁ - f₂ and the stabilized frequency f₂ to determine.

If the first oscillator 16 can also be regulated, the frequency of the transmission signal can be modulated in a very simple manner with a (modulation) frequency control loop 54. For this purpose, a low-frequency signal, the frequency f_mod is modulated in the desired manner (for example, in the case of an FMCW radar with frequency modulation in accordance with a linear ramp), is applied to an input of a phase comparator 56 which compares the phase of this signal with the phase of the I reference signal I present at the other input_ref compares. The comparison result is fed to the oscillator 16 as a control signal via a low-pass filter 58. In this way, the frequency f₁ of the first oscillator 16 controlled so that the frequency f₁ - f₂ of the I reference signal with f_mod is kept in agreement. From f₁ - f₂ = f_mod then follows: <maths id="math0004" num=""><math display="block"><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">1</mn></msub><mo mathvariant="normal">=</mo><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">2</mn></msub><mo mathvariant="normal">+</mo><msub><mi mathvariant="normal">f</mi><mi>mod</mi></msub></math><img file="EP2483706B1_D0004.tif" /></maths> and for the frequency of the transmission signal E: <maths id="math0005" num=""><math display="block"><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">1</mn></msub><mo mathvariant="normal">+</mo><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">2</mn></msub><mo mathvariant="normal">=</mo><mn mathvariant="normal">2</mn><mo>⁢</mo><msub><mi mathvariant="normal">f</mi><mn mathvariant="normal">2</mn></msub><mo mathvariant="normal">+</mo><msub><mi mathvariant="normal">f</mi><mi>mod</mi></msub><mn mathvariant="normal">.</mn></math><img file="EP2483706B1_D0005.tif" /></maths>

<figref idref="f0003">Fig. 3</figref> illustrates a variant in which the Q reference signal Q_ref is used to reduce the phase noise of the oscillator system. The Q reference signal Q_ref is formed by mixing the signal of the first oscillator 16 with the signal of the phase shifter 26 which is phase-shifted by 90 ° in the second transmitter mixer 28 and has the frequency f₁ - f₂. Its phase therefore reflects the phase noise of both oscillators as well as any phase errors of the phase shifter 26. In the example considered here, f₁ and f₂ chosen so that their difference f₁ - f₂ just 1/32 of f₂ is. In a phase locked loop 60, the phase of the Q reference signal is compared with the 1/32 frequency-divided signal of the second oscillator 18, and the comparison result is fed to the second oscillator 18 as a control signal via a low-pass filter 64. In this way the phase noise is clearly suppressed.

If the frequency of an oscillator is halved, there is generally a 6 dB reduction in phase noise at the output of the frequency divider. Since the frequency in the frequency divider 52 is 1/32 (1/2⁵) is divided, there is arithmetically a reduction of the phase noise by about 30 dB.

<figref idref="f0004">Fig. 4</figref> shows as a further embodiment a radar sensor configured as a hetorodyne sensor. The signal of the first oscillator 16 directly forms the transmission signal S, while the first comparison signal V₁ by the signal of the second oscillator 18 and the second comparison signal V₂ is formed by the signal of the second oscillator 18 which is phase-shifted by 90 ° in the phase shifter 26. As an example, it can be assumed that the first oscillator 16 has a frequency of 122 GHz (transmission frequency), while the second oscillator 18 has a frequency f₂ of 120 GHz. In this case, the I and Q mixers 32, 34 deliver I and Q signals I_IF, Q_IFwhich are in an intermediate frequency band and have a frequency on the order of about 2 GHz.

The transmission mixers 20 and 28 mix the transmission signal S with the comparison signals V₁ and V₂ (The term "transmission mixer" only serves to differentiate between mixers of the transmitting part 10 and the receiving part 12 and does not necessarily mean that these mixers are also involved in the generation of the transmission signal or the comparison signals). The I and Q reference signals I obtained as mixed products_ref and Q_ref lie in the intermediate frequency band and have the frequency f₁ - f₂ = 2 GH_e.g..

The I and Q reference signals can be used in a similar manner as in the previously described embodiment. For example, a frequency divider 66 enables the frequency f₂ of the second oscillator 18 divides in a ratio of 1/64, a phase noise suppression similar to the phase locked loop 60 in <figref idref="f0003">Fig. 3</figref>. A control input 68 is also provided for the phase shifter 26, so that the phase shift can also be controlled in a similar manner to that with the phase locked loop 40 in<figref idref="f0002">Fig. 2</figref>.

Since the transmission signal S and the comparison signals V₁ and V₂ are generated here by different oscillators, the phase noise of the comparison signals is not correlated with the phase noise of the transmitted signal, which leads to corresponding noise effects in the I and Q signals I_IF and Q_IF leads. However, the I and Q reference signals contain the same noise effects because they are created by mixing the transmission signal with the same comparison signals. Hence the phase noise or I and Q signals is I_IF and Q_IF with the phase noise or I and Q reference signals I_ref and Q_ref correlates, and if these signals are mixed together to obtain corresponding I and Q signals in the baseband, the errors caused by the phase noise are largely eliminated and the phase of the radar echo can be determined with high accuracy. Optionally, the I and Q signals I_IF and Q_IF also directly with the associated reference signals I_ref and Q_ref are compared to determine the phase difference.

<figref idref="f0005">Fig. 5</figref> illustrates the use of transmitting and receiving parts with the in <figref idref="f0002">Fig. 2</figref> shown construction in a radar sensor with several transmit and receive channels. For simplicity, are in<figref idref="f0005">Fig. 5</figref> only two channels 70, 70 'are shown, the transmitting and receiving parts of which are each on a circuit board, but the principle illustrated here can easily be extended to radar sensors with more than two channels. The structure of the transmitting and receiving parts in<figref idref="f0005">Fig. 5</figref> differs from the structure in <figref idref="f0002">Fig. 2</figref> only in that <figref idref="f0005">Fig. 5</figref> a monostatic antenna concept is implemented and accordingly a circulator 72 is provided to separate the received signal E from the transmitted signal S.

The transmission signals S and S 'in the two channels 70, 70' have frequencies around a fixed frequency f_shift are shifted against each other. The frequency f_shift is generated with a fixed frequency oscillator 74. In addition, both transmission signals S and S 'are synchronously frequency modulated. For this purpose, a variable oscillator 76 generates the modulation frequency f_mod.

The modulation frequency f_mod fed directly to the frequency control loop 54 to the frequency generated by the variable oscillator 16 (here with f_v designated) with the modulation frequency f_mod to lock. The first oscillator 16 generates a fixed frequency f. The sum frequency f + f_v of the transmission signal S is thus in the by f_mod predetermined way modulated.

In contrast, the frequency control loop 54 in the second channel 70 'does not directly have the frequency f_mod fed, but rather the frequency f_mod + f_shiftwhich are obtained by mixing the signals of the oscillators 74 and 76 in a mixer 78 and then filtering them in a high-pass filter 80.

For the sake of simplicity, it should first be assumed that the frequency f that the second oscillator 18 generates in the second channel 70 ′ is identical to the frequency f of the second oscillator in the first channel 70. The frequencies of the transmission signals S and S 'then differ only by the frequency f_shift and also have an identical frequency modulation.

In the first channel 70, the I and Q signals give I_BB and Q_BB, after low-pass filtering in a filter 82, the frequency shift of the radar echo caused by the distance and the relative speed of the object. These signals in a baseband BB₁₁ are digitized, recorded as a time signal and then broken down into a spectrum by fast Fourier transformation, which is then further evaluated in a known manner.

It is true that signals are also received in the first channel 70, which in the second channel 70 'with the f_shift shifted frequency were sent, but these signals are outside the baseband BB₁₁, so that the signal evaluation in this baseband BB₁₁ are not disturbed by the signals originating from other channels.

In the construction shown here, however, mixers 84 are provided for the I and Q signals I_BB and Q_BB with the frequency f_shift to mix, so that after low-pass filtering in filters 86 in turn signals in a baseband BB₂₁ receives. These signals permit the evaluation of the amplitude and phase of cross echoes, that is to say of signals which were transmitted in the second channel 70 ′ and then received in the first channel 70.

The second channel 70 'is constructed symmetrically to the first channel 70. The transmission signal and the comparison signals have the frequency f + f '_v. So you get signals in a baseband BB in this channel₂₂, which represents the radar echoes transmitted and received in this channel, and signals in a baseband BB₁₂which represents the cross echoes transmitted in the first channel 70 and received in the second channel 70 '.

If the fixed frequencies f and f are not exactly the same, but a known difference frequency f_diff distinguish, so the signals in the basebands BB₁₁ and BB₂₁ and accordingly the signals in the base bands BB₂₂ and BB₁₂ based on the difference frequency f_diff correlate with each other, and one can thus determine and evaluate the phase differences between the direct radar echoes and the cross echoes, which are caused by the differently long signal paths from the antenna 24 of the one channel to the object and back to the antenna of the same channel and from the antenna of the one channel to Object and back to the antenna of the other channel.