Method and apparatus for frequency conversion, in particular for controlling the emitting power of a cellular mobile telephone

The invention relates to the frequency of transposition and applies advantageously but not exclusively in the field of radio frequency, for example in mobile telephony, in which RF circuits widely used devices frequency conversion or frequency mixers, both transmission and on reception.

At transmission, the frequency mixers, which are in this case frequency boosting circuit, are intended to transpose Information baseband around the carrier emission.

Figure 1 schematically illustrates the structure usually used for transposition devices frequency of the prior art.

On the upper part of Figure 1, the reference designates a MIX frequency transposition means, or mixer (here elevator frequency) having an input terminal for receiving a signal BES SI incident, such as baseband or at a frequency intermediate. The mixer MIX also has a terminal BCO order to receive a local oscillator signal LO by example around 2 GHz in a mobile phone application using CDMA systems (multiple access systems division code: "Code Division Multiple Access," in English), and BS output terminal for outputting the output signal STR which is a transposed signal whose frequency spectrum is located around of the fundamental frequency of the local oscillator signal and odd harmonics. The amplitude of these harmonics decreases in proportion to their rank, that is to say, it decreases in dB with a slope of -20 dB per decade.

Thus, in practice, the mixer MIX is followed by a filter bandpass centered about the fundamental frequency the local oscillator, so as to retain only the part of the spectrum centered around the fundamental frequency.

The mixers usually used for these structures is a Gilbert-type structure, usually differential, such illustrated diagrammatically in the middle part of Figure 1.

Such a cell is well known to those skilled in the art and are recalls the essential characteristics here.

More specifically, such a cell comprises a block BTC differential transducer for converting the input signal (Voltage) present on the terminals BES into a differential current. Here BTC block comprises a stage constituted by a differential pair of transistors whose respective bases are connected to the input terminals by two capacitors. The collectors of the two transistors of this stage form the output terminals of this CS transducer block. Good Naturally, the BTC block can have multiple floors.

The transistors of the floor of the block BTC are biased by conventional biasing means comprising in particular MPL resistors and a voltage source.

At the output of transducer block BTC, that is to say the collectors of the transistors of this block BTC, is connected to a block BCC current switching directing the current alternately to one or other of the two output terminals BS to the signal frequency LO LO received at terminals BCO. This block BCC conventionally comprises two pairs of transistors.

Each resistance ZL connected between the output terminals BS BCC block and the Vcc supply, represents the output load MIX mixer.

The BTC block converts the power or the voltage applied to BES into a differential current which is a supposed picture linear input signal. This linear signal is then cut by a non-linear square function (+1, -1, +1, -1, ...) carried out by the Double BCC switch, the frequency of the LO signal, this double switch router acting as dynamic current. The output signal is collected at the terminals of the differential charge 2ZL.

In other words, as shown schematically also on the lower part of Figure 1, the RTS signal at the terminal BS output of the mixer MIX matches multiplied incident signal SI by one (that is to say non-inverting) and -1 (that is to say reversed) at a rate the periodic local oscillator signal LO, usually clipped to 1 and -1.

Thus, in such a conventional mixer, the control input or local oscillator input, receives a transposition signal periodic (local oscillator signal) having the frequency transposition desired, and with a fixed power level required to drive the transistors of the cell BCC switching block Gilbert.

Furthermore, the power of the transposed signal is equal to losses close to the power of the incident signal.

Under these conditions, and assuming that it has a low power incident signal that should pass after transposition, with high power, it is necessary performing an amplification signal, the latter comprising usually an amplification of the incident signal before transposition and amplification after transposition.

However, when a strong amplification of the incident signal is necessary in certain applications, such as telephony mobile, it is particularly difficult to achieve because the linearity of the incident signal is to be preserved during the amplification in order not to lose information when mixing (transposition).

The invention aims to provide a solution to this problem.

An object of the invention is to provide a transposition frequency integrating somehow an amplification function, without affecting the linearity of the incident signal to be transposed.

The invention also aims a power control emission of a signal transmitted by a component of a system wireless communication, for example a cellular mobile telephone.

The invention provides a frequency transposition process of an incident signal, comprising a modulation of the incident signal delta-sigma type over one bit so as to generate a binary signal representative of the incident signal, a generation of an auxiliary signal periodic frequency equal to the desired transposition frequency, and succession of inversions and non-inversions signal auxiliary, controlled by the successive states of the binary signal, of to generate a transposed signal.

Thus according to the invention, the inlet of the usual signal from a mixer receives here not to transpose the incident signal as in the prior art, but an auxiliary signal or carrier signal, periodic and frequency equal to the transposition frequency desired (this signal is therefore the transposition signal), while the usual control input of a mixer is not receiving the transposition signal (local oscillator signal) as is the case in the prior art, but a binary signal (that is to say coded on two states +1 and -1, for example) from a modulation of sigma-delta 1-bit of the incident signal.

And, the power of the output signal thus depends on the power of the incident signal but also the amplitude of the signal auxiliary.

Moreover, the issue of linearity when signals high power is more critical here according to the invention, since there is no Information loss of the incoming signal. Indeed, the signal at the output delta-sigma modulator, although it has a constant amplitude, contains all the information of the input signal (amplitude and phase) encoded on the time axis. Thus, the delta-sigma modulator allows encode the signal with one bit while rejecting quantization noise outside the useful band of the signal.

It is furthermore particularly advantageous that the amplitude the auxiliary signal is variable, which then allows an adjustment of this amplitude as a function of the desired power for the signal transposed.

The invention also provides a transposition device frequency comprising<ul><li>an input terminal for receiving an incident signal,</li><li>a delta-sigma modulator type a bit connected to said input terminal,</li><li>means for generating a periodic auxiliary signal frequency equal to the desired transposition frequency, and</li><li>frequency transposition means of the type of cell Gilbert having a signal input connected to the output of generating means, a control input connected to the output the delta-sigma modulator, and an output delivering a transposed signal.</li></ul>

According to one embodiment of the invention, the amplitude of the auxiliary signal is variable and the device then further comprises adjustment means of this amplitude as a function of an output desired for the transposed signal.

The incident signal may be an analog or digital signal.

The invention also provides a component of a system wireless communication, for example a cellular mobile telephone, comprising a transmission channel comprising a processing stage digital delivering two digital signals in baseband quadrature, and an output amplifier stage. The transmission channel then comprises at least one frequency transposition device such as defined above, disposed between the digital processing stage and the output amplifier stage.

The invention also provides a method of controlling the power of a signal transmitted by the transmission channel of such a component, this method comprising an adjustment of the amplitude of the signal Auxiliary applied to the input of the signal translation device frequency.

Other advantages and features of the invention apparent on examining the detailed description of mode realization and implementation, in no way limiting, and drawings accompanying, in which: <ul><li>Figure 1, already described, illustrates a transposition means according to the prior art;</li><li>Figures 2 and 3 illustrate very schematically a device frequency transposition according to the invention;</li><li>4 schematically illustrates a delta-sigma modulator 1-bit embedded in a transposition device frequency according to the invention;</li><li>5 schematically illustrates the structure of a cellular mobile telephone according to the invention, incorporating in its path transmitting at least one frequency transposition device according the invention;</li><li>6 illustrates in more detail one embodiment of the transmission channel of a cellular mobile telephone according to the invention; and</li><li>7 schematically illustrates another mode realization of a transmission channel of a cellular mobile telephone according to the invention.</li></ul>

In Figure 2, the reference DTF denotes a device frequency transposition with transposition means or frequency mixer MIX, the type Gilbert cell. Thus, the MIX mixer of Figure 2 may be structurally identical to the Mixer MIX of the middle part of Figure 1.

This mixer MIX and has an input signal of BES, BCO a control input and an output delivering a signal BS STR transposed.

The frequency transposition device DTF has also generating means MGN capable of generating a signal Auxiliary SAX. This auxiliary signal or carrier signal is a periodic signal having a frequency equal to the frequency of desired transposition. In this case, this auxiliary signal is a signal Acos2πcf form₀t, where f₀ refers to the frequency of transposition (ω₀ = 2.pi.f₀).

The amplitude A of the auxiliary signal is advantageously SAX adjustable and will allow adjustment to vary the power of output signal, that is to say the power of the transposed signal STR.

Such generation means are of conventional structure and known per se.

Another essential element of the invention is a type delta sigma modulator, referenced MDU, input with the incident IF signal to be transposed and outputting at its terminal BSM a binary signal, that is to say with two states, for example +1 and -1, IF signal representative of the incident. In other words, by the delta-sigma modulation on 1 bit, the signal SCB output from MDU modulator has a constant amplitude and contains all information of the input signal SI, both amplitude and phase, this information being encoded in the time axis. Furthermore, the delta-sigma modulator to encode not only the IF signal 1 bit, but also to reject the quantization noise outside of the useful band of the incident signal SI.

Although the mixer MIX is structurally example identical to a mixer of the prior art, the nature of the signals that receives at its two inputs, namely the input signal on the one hand, and the second control input, is totally different and unusual with respect to the mixer of the prior art.

Specifically, while the inlet of a mixing signal in the prior art receives the signal to be transposed, the input signal BES MIX mixer of the invention here receives the auxiliary signal, that is to say the carrier signal, which will set the frequency of transposition.

Furthermore, while in the prior art input transposition control receives the signal, the mixer MIX according the invention receives its BCO control input coded signal SCB bit from the signal to be transposed after modulation type delta-sigma of 1 bit.

STR signal, which is the transposed signal delivered to terminal BS output of the mixer MIX, contains the spectrum of the incoming signal SI transposed around the transposition frequency F₀.

In addition, the output power of the signal STR depends both the power of the incident IF signal, but also the amplitude A the auxiliary signal SAX. Thus, it is possible to control the power STR signal by acting on the amplitude A of the carrier signal SAX.

In Figure 3, which illustrates another representation schematic of the mixer MIX according to the invention, we see that is carried successions of inversions and non-inversions of auxiliary signal SAX (that is to say, successions of multiplications by +1 and -1), these successions of inversions and non-inversions being controlled by the successive states of the binary signal SCB. Thus, for example, each time one has a 1 state for the signal SCB, it will perform a non-inverting SAX signal, while one reverse the SAX signal at a -1 state of the CBS signal.

The structure of a type of delta-sigma modulator is classic and well known in the art. We recall here briefly the main features, with particular reference to Figure 4.

The modulator MDU comprises an adder S1 head (Subtracting) receiving the incident IF signal. This subtractor is followed here an operator INT1 whose output is connected to the input means quantization QTZ (sampler) whose output constitutes the output of the modulator MDU. The output of quantization means QTZ is fed back to the negative input of the adder S1 by via a gain G.

Modulation "delta" is based on the quantitation of modification of the sample to the sample signal, rather than the quantification of the absolute value of the signal at each sample.

The presence of an integrator (sigma) in the modulator gives the modulator the modulator naming "delta-sigma".

The output of the delta-sigma modulator is at a frequency of oversampling very high Fs. It is a characteristic fundamental of delta-sigma modulators because they use high frequency portion of the spectrum to repel the major part of quantization noise.

Indeed, it is recalled that a delta-sigma modulator is designed to push the quantization noise out of the signal band useful.

The delta sigma modulator MDU described herein is a modulator order 1 because it has only one subtraction and one integrator. However, one can use a delta-sigma modulator higher order. Furthermore, since the modulation is performed on a bit, the quantization means here is a simple comparator delivers two levels, +1 or -1, for example.

Finally, because of the presence of an integrator, the modulator delta-sigma here is a delta-sigma modulator low pass. However, the invention is also compatible with delta-sigma modulators bandpass which include, instead of an integrator, a filter bandpass. These modulators, although they do not possess integrator (sigma), always keep the abuse of language name "delta-sigma modulator."

There will now be described with particular reference Figures 5 and following, an application example of such a frequency transposition in a mobile telephone system.

In Figure 5, the reference TP designates a remote terminal, such a cellular mobile phone, which is in communication with a base station BS1, for example according to a communication scheme of CDMA.

The cellular mobile telephone includes, in conventional manner, a radio frequency analog stage ERF connected to an antenna ANT via a duplexer DUP for receiving a signal input.

Conventionally, the stage ERF comprises a low amplifier noise and two processing pathways including mixers, filters and conventional amplifiers. Both mixers receive respectively from a two phase locked loop signals mutually having a difference of 90 ° phase. After frequency transposition in the mixers, the two channels treatment respectively define two streams I (direct stream) and Q (Quadrature stream) according to terminology well known to the man of career.

After digital conversion in converters analog / digital, the two streams I and Q are delivered to a floor ETNR reception processing part of a module digital processing baseband BB.

This ETNR processing stage comprises, conventionally, a receiver commonly referred to as the art "receiver Rake ", followed by conventional demodulation means performing demodulation of the constellation delivered by the receiver Rake.

The baseband processing block BB comprises, besides ETNR treatment beaches, a transmission processing stage ETNE that performs, in a conventional manner, including coding treatments source, spreading symbols, modulation, to deliver two streams I and Q baseband analog block transmission CHM conventional structure. The CHM block is conventionally followed one FTE power amplifier stage connected to the antenna the duplexer DUP.

In CDMA systems, in particular systems WCDMA, in which the transmission and reception are simultaneously, we must control the power of the transmitted signal on a wide range, typically about 80 dB, the signal strength to vary from -50 dBm to +24 dBm.

Currently, this power control requires the use several amplifier stages, in general variables, which is General complex to achieve, particularly from the standpoint of linearity amplification.

The use in the process of a mobile phone transmits cell, of at least one frequency transposition device according the invention solves this problem.

More specifically, in Figure 6, a delta-sigma modulator lowpass a bit referenced MDUI is connected within the block processing baseband BB, the output of the processing stage Digital ETNE towards I.

Similarly, an analog modulator is connected in MDUQ off the floor on NSEL channel Q.

These two modulators are sigma-delta modulators digital, that is to say that the integration function as well as summing functions are performed digitally.

The two baseband I and Q signals thus undergo delta-sigma modulation on one bit and, after digital conversion analog, they are issued respectively on the inputs Control of the two mixers MXI and MXQ.

Furthermore, the means for generating the auxiliary signal MGN delivers an auxiliary signal or carrier signal type cosine, to a radio frequency transposition frequency F_RF.

The auxiliary signal SAX is then delivered on the signal input MIXI the mixer, as well as the mixer signal input MIXQ, after a phase shift of π / 2.

The two transposed signals to the transposition frequency, respectively from the mixers and MIXI MIXQ, are summed in an adder S10, then the resulting signal is filtered in a filter bandpass FLT centered around the transposition frequency F_RF. The filtered signal is then delivered to an output amplifier PA the amplification stage ETP.

Information relating to the power of the transmission signal output of the amplifier PA, is returned to the processing stage NSEL and digital embedded processor in this stage can then, based on this information, adjusting the amplitude A of the signal auxiliary SAX, so as to control the power of the output signal mixers, and therefore the power of the transmission signal.

In the example transmitter architecture of Figure 6, the I and Q signals from the digital processing stage in strip base are encoded by delta-sigma modulators lowpass 1 bit before to be applied to two quadrature mixers for the to transpose the emission frequency. However, it is also consider an exemplary architecture as illustrated in Figure 7, wherein it first passes through an intermediate frequency, which may be performed in digital or analog manner, combined with the use of a delta-sigma modulator 1 bit bandpass.

More specifically, as illustrated in Figure 7, the signals I and Q, from the stage ETNE undergo a delta-sigma modulation in delta-sigma modulators lowpass 1 MDUI and MDUQ bit, before being transposed in two MX1I MX1Q and mixers, a intermediate frequency F_IF, Using a carrier signal cosine generated by first generating means MGN1.

The two transposed signals thus obtained are summed in a summing S10, then the summed signal is filtered in a first filter FLT1 bandpass centered around the intermediate frequency F_IF.

The filtered signal is then subjected to a delta-sigma modulation 1-bit MDUPB in a modulator that is a 1-bit delta-sigma modulator bandpass.

The modulated signal is applied to the control input of a MIX3 mixer, which will translate the signal to the transmission frequency RF F_RF using a cosine carrier signal having a frequency F_RF-F_IF, Applied to the input signal MIX3 mixer.

The output of the transposed signal is filtered in a second filter FLT2 passband centered around the transmit frequency RF F_RFBefore being amplified by the output amplifier PA.

Again, the transmit power of the output signal is transmitted to the digital processing stage ETNE, so that the processor baseband can optionally act on means of MGN1 and MGN2 generation, so as to adjust the amplitude A1 of first carrier signal and / or the amplitude A2 of the second signal carrier.