Method and circuit for limiting an overvoltage

Translation of description of equivalent WO 2004008601 A1

description

Method and circuit for limiting an overvoltage

The invention relates to a method and a circuit arrangement for limiting an overvoltage which reacts by switching an inductive load, such as an electric motor to a load driving.

When inductive loads are switched, so arise when switching high inductive reverse voltages, which can destroy the Lasttrei over. In order to reduce the overvoltages, it is common for so-called free-wheeling diodes in parallel to the load driver, the power surges or peaks derived from the load drivers.

One application in which an inductive load is controlled, for example, is an inverter circuit, with which a Elect is operated romotor. In Figure 6, a known converter circuit is shown schematically. A z. B. three-phase AC voltage of an AC voltage source 1 is first rectified by a rectifier stage 2, with the aid of load drivers 3 into an AC voltage variable out - put frequency converted and a load 4 is supplied (a three-phase electric motor M here). The signals for generation of the output voltage provides a microcontroller fifth These are converted into suitable pulses for driving the load driver 3 using gate drivers GD. A basic element of such an arrangement (shown framed in Figure 6) a half-bridge. 6

In such a converter for electrical drives the load 4 has in the form of an electric motor M inductive components. With the inverter in the three phases of the motor M nearly sinusoidal currents through pulweitenmoduliertes generates switching load drivers. 3 To protect against induced surges, each load driver 3 each a parallel arranged to him freewheeling diode 7. Due to the inductive components in the load 4 while the current between the load driver 3 and the corresponding freewheeling diode 7 commutated within a half-bridge 6 back and forth (see FIGS. 6 and 7).

Such half-bridge 6 is shown separately in figure 7 and in each case has two series-connected load driver 3 that between a supply potential V +_C and a reference potential are arranged 0V. Each load driving is respectively connected a freewheeling diode 7 parallel. The load 4 is closed to the common terminal of the load driver 3 other. The result is a bridge circuit, which allows to drive the load 4 with an AC voltage.

The supply voltage V_cc is temporarily stored in an intermediate circuit capacitor 8, the supply voltage V_Cc provides for half-bridges. 6 An intermediate circuit is formed by the intermediate circuit capacitor 8 and a half-bridge. 6

In each half-bridge 6 component internal Streuinduktivitä- act th L_σ, Int and due deiϊ construction external leakage inductances L_σ._Θχt in addition to the inductance of the load 4. The two load driver 3 are driven so that only one of the two load driving forwards, while the other is blocked.

In Figure 7, the fundamental current waveform is also in a half-bridge 6 during commutation of a current from the inductive load L_L represented by the upper freewheeling diode 7 in the bottom, switching-on load driver 3 (the Lasttrei- about 3 switch alternately; while one is turned on, the other is turned off). Before switching on the freewheeling through the upper freewheeling diode 7 current flows I_v through the load L_L, The upper freewheeling diode 7 and on to inherent stray inductances L_σ, ι_nt and L_σ._βχt - Upon completion of the commutation, d. H . (Commonly as a load driver 3 switches in semiconductor design used and therefore also the power semiconductor or semiconductor circuit - breaker called) when the lower load driver 3, the current flowing through the load L_L and through the lower load driver 3 and the lower stray inductance L_σ, i_n and L_σ, Ext. shown in dashed lines is the current contribution I_z by the transition of the freewheeling diode 7 from the conducting to the blocking state during the shift.

This proportion causes in Figures 8A to 9B gezeig- th reverse current spikes. The component internal and construction-related external leakage inductances L_σ, i_nt or . L_σ._ext can during the decay of the reverse current spike, an overvoltage greater than the intermediate circuit voltage or supply voltage V_Cc induce at the freewheeling diode 7, the u. U. the for load - exceeding driver 3 or the freewheeling diode 7 maximum permissible voltage (freewheeling diode 7 can be destroyed).

When transferring the current from the freewheeling diode 7 in the load driver 3, the slope of the current rise in the load driver 3 by the actuation of the load driver 3 is determined. In addition to the current from the load L creates a so-called reverse current peak, which is due to the storage charge of the freewheeling diode. 7 The additive reverse current spike is fed from the intermediate circuit capacitor 8 and flows via the leakage inductance L_σ, ι_nt or L_σ._ext in the intermediate circuit. Depending on the optimization of the freewheeling diode 7 this reverse current peak is more or less rapidly. Height and decay are determined by the steepness di / dt of the current rise in the load driver 3 beeinf lussbar.

Depending on the size of the leakage inductance L_σ, ι_nt and L_σ._ext / Dt in the DC and current steepness di arises at the rise of Current in the load driver 3 a voltage drop at the half-bridge 6 and the load driver 3, as amended by Law u (L_σ) = L_σ* Di / dt is determined. When the reverse current peak decay occurs at these stray inductances L_σ, i_nt and L_σ._eχt a reverse voltage generated at the freewheeling diode 7, a voltage peak. For very fast-ending freewheeling diodes 7 can by high leakage inductance L_σ, i_nt and L_σ._ext overvoltage spikes occur until the destruction limit.

In the figures 8A and 8B, the voltage on the component V_ce during the switching process, the switched load current I_c through the load driver 3 and the gate voltage V_GE at the switching load driver 3 '(Figure 8A) and at the freewheeling diode 7 (with the parallel, blocking load driver 3' '; Figure 8B) shown schematically.

When switching the gate voltage V rises_GE at the switching load driver 3 'from -15 V to +15 V, the voltage V_ce driver at load 3 drops of + V_Cc (Positive potential of the supply voltage or intermediate circuit voltage V_Cc) to approximately 0 V.

The current through the load driver (load current I_c) Increases rapidly and, after a brief current spike its nominal value I_c on.

At the blocking load driver 3 '' parallel freewheeling diode 7 ' ^■ decreases the freewheeling from face value I_c (Also referred to as reverse current peak with a brief overshoot or negative current peak) to zero. The local gate voltage V_GE the suspended load driver 3 '' remains at -15 V, while the voltage V_ce the load driving from 0 V to + V_Cc increases (also a short, strong overshoots; also known as Surge).

Such commutation (current reversal or changeover) can also be found in other applications with inductive loads and comparable switching operations take place (for example, in switching power supplies). To protect the freewheeling diodes 7 before such lightning has been (at least known internally) the controlling load driver 3 slowly turns the commutation Community, whereby the reverse current peak decreases and the freewheeling diode 7 less steep tear off (see FIGS. 9A and 9B).

In the figures 9A and 9B the influence of overvoltage and reverse current spike by slower switching the load is's operator by using a larger gate resistor R_G shown (Figure 9A shows the conditions at the switching load driver 3 'and 9B the conditions at the the blocking load driver 3' 'parallel freewheeling diode 7' '). For comparison, shown in FIGS 8A and 8B, current / voltage curves are shown in dotted lines respectively. In particular, the losses in the switching load driver are thereby, as can be seen from the current / voltage profiles in FIG 8A, significantly increased.

It is - internally at least - known effective

stray inductance L_σ to reduce by a so-called SnubberbeSchaltung. If the current rate of rise reduced by slower driving the load driver 3, this leads to increased, not acceptable switching losses in the load driver 3. A Snubberbeschaltung generates increased costs and also additional losses in the snubber circuit.

It is an object of the invention to provide a method and a circuit arrangement for limiting an overvoltage can be reduced by the additional losses. In particular, strong overvoltage and reverse current spikes to be reduced.

This object is achieved by a method for limiting an overvoltage with the features of patent claim 1 and by a circuit arrangement having the features of claim 7 and 13 respectively. Here, a lying parallel to a freewheeling diode (semiconductor) switch is power utilized during the commutation voltage (hereinafter referred to as load drive) as a delimiter for the free-wheeling diode in the pending immediate (over-). Limiting the overvoltage by a plane parallel to load drive is achieved by this load driving is briefly, just turned on so far during the decay of a reverse current spike that an addi- tive falling current in the semiconductor switch Freilaufdioden- pair arises. The additive, falling current thus reducing the current rise in circuit construction related stray inductances and reduces the induced voltages. The brief switching is timed with the occurrence of the reverse current peak.

Advantageous developments of the invention are given in the subclaims.

So can be used as a so-called semiconductor switches IGBTs in which the control terminal and the output terminal of the blocked semiconductor switch are high impedance interconnected, whereby when a current peak occurs, a temporary activation of the semiconductor switch is reached. This is a very easy way to counteract the voltage spike to a certain extent.

It is also advantageous to subject the control terminal of the blocked semiconductor switch with an approximately rechteckför strength voltage pulse of predetermined duration and an amplitude less than a predetermined threshold value during the occurrence of a current spike or at least during the decay of current spike.

Even more advantageous it is, rather than a simple rectangular pulse to use a step-shaped voltage pulse, the largest amplitude momentarily at least during the ex- sounding the reverse current peak is present, in order to counteract this reverse current peak.

The voltage at the output terminal of the other, locked semiconductor switch may additionally (or alone) are fed back via a feedback path to its control terminal. Thus, the blocked semiconductor switch during the presence of the current peak or inductive voltage is switched on briefly. This reduces the current gradient in the waste of the current peak and thus the surge.

This effect can be further increased, if - in addition to the feedback - the control terminal of the semiconductor switch with a voltage pulse during the presence o the decay of the current peak is applied, whereby the access blocked semiconductor switch is partially turned on.

The feedback path advantageously has at least such a device by which the feedback path is activated only when the voltage exceeds a threshold. So can it be used nerdioden two series-connected Zener. It is thereby achieved that only when a predetermined voltage is exceeded, a driving and thus a partial turn on the blocked semiconductor switch per se is achieved, and this timed with the occurrence of overvoltage. Alternatively or additionally, the voltage increase can, for example, at the Freilaufeinrich- device with the aid of a capacitor are fed back to the control terminal of the load driver. Thereby, the commutation process is so influenced that height and slope of the peak reverse recovery current decrease. Thus, it is possible with simple inexpensive means an effective reduction of an inductive overvoltage and reducing harmful current peaks with a large current rise. In addition, the surge or the reverse current peak is entgegenwirken- de activation of the semiconductor switch in time parked in the presence of the peak reverse recovery current. The threshold at which the feedback is to be activated, can also be easily adjusted coupling branch by appropriate dimensioning of the components in the rear.

Embodiments of the invention will be explained below with reference to the schematic drawings. Show it:

Figures 1A and IB current / voltage profiles at a switching-on semiconductor switch or on a one blocking semiconductor switch connected in parallel freewheeling diode,

2 shows a first embodiment of an inventive circuit arrangement with a feedback branch with Zener diodes,

Figure 3A shows another embodiment of an inventive circuit arrangement having a feedback branch with a separate condenser,

Figure 3B shows another embodiment of an inventive circuit arrangement with a branch Rückkoppel- proportional for a signal to the current rise in the diode,

Figures 4 and 5 current / voltage profiles at a switching-on semiconductor switch according to two exemplary embodiments of a method,

Figure 6 is a known converter circuit for controlling a three-phase electric motor,

7 shows a half-bridge inverter circuit of the known GE measure Figure 5 with the therein occurring electrical

Stream, Figures 8A and 8B current / voltage erlaufe at a switching-on semiconductor switches of the half-bridge of Figure 6 and

Figures 9A and 9B current / voltage waveforms at a switching-on semiconductor switches of the half-bridge of Figure 6 with the use of a larger gate resistance.

In the figures and the embodiments, like parts have harmful ionic nell given identical reference numerals.

To control or switching of inductive loads L_LSuch as an electric motor (see FIG. 6), are often used as power semiconductor devices drivers (referred to as a power switch, as a load driver 3 or as a semiconductor switch). The load driver 3 itself are usually controlled by a control unit with a microcontroller. 5 Since the switching of inductive loads L_{L /} especially when switching off the loads L_L, Undesirable, high voltages are induced, are in parallel with each load driver 3 are each a free-wheeling diode 7 is connected, which is to reduce an induced surge.

It is usual, a half-bridge is 6 for controlling an inductive load L_L to use, with two load drivers 3 are connected with each parallel freewheeling diodes 7 consecutively and alternately turn (the switched load driver 3 'and the suspended load driver 3' 'hereinafter). Between the two load drivers 3 is the output terminal for an inductive load L_L arranged.

As load driver 3 here two IGBT power transistors are used (see FIG. 2, there is a complete half-bridge 6 shown). Before the control terminal (gate) of each IGBT is a gate resistor Rg and before a Endverstärke - stage of a gate driver GD (see FIG. 6) arranged. One possible embodiment of such an amplifier stage is the in 9 shown emitter follower circuit (Figure 2, Figure 3, circuit part 9). The gate driver GD for its part - as shown in Figure 6 - connected to a microcontroller. 5 The output amplifier stage 9 of the gate driver GD is connected to a positive supply voltage V_G+ And a negative supply voltage V_G- And supplied via a common control terminal with a control signal V_{ga edrive} (See. Also fi gure 4) acted upon. This control signal V_ga EDRI _e is ultimately connected through to the gate of the IGBT.

The half-bridge 6 is üblicherwe ise designed as an integrated component. By component internal stray inductances Lσ. int and due to circuitry external leakage inductances L_σ, E may be larger than the supply voltage V during the decay of a disease caused when switching peak reverse current surges (voltage_cc the half-bridge 6) are induced on the free-wheeling diode 7 which u. U. exceeds the maximum allowable voltage components used. This surge (or a current spike with a large current gradient di / dt or a steep voltage rise du / dt), which are induced during switching of the load driver 3 due to bef in circuit indlichen inductances, it is now time to reduce.

According to the invention the load drive in parallel with the freewheeling diode 7 is located 3 (IGBT) as a limiter for the over-voltage at the freewheeling diode 7 utilized during switching. The limitation of the surge through the plane parallel load driver 3 is achieved by the fact that this load driver 3 briefly during the decay of a reverse current spike, just turned specifically to the extent that an additi ver, falling current in the load driving freewheel diode pair is created. The additive, falling current counteracts the induced current spike counter, thus reducing the current part - unit in the leakage inductance L_σ. _int and L_σ._ext. Thus, the induced overvoltage will be lower. The gate and emitter of the IGBT are connected with each other a high impedance in the off state of the IGBT. This causes the desired momentary turning on the IGBT when the freewheeling diode in parallel lying 7 commutated because the tearing of the freewheeling diode current in addition to the high rate of current rise di / dt also causes a steep rise in voltage dv / dt at the freewheeling diode 7 and the parallel IGBT.

The voltage rise du / dt via the so-called Miller capacitance in the IGBT to an increase of the gate voltage V_GEWhich is only slowly degraded due to the strength hochoh gate-emitter connection. Thereby, the IGBT for the duration of the voltage rise du / dt is partially turned on. This in turn reduces the voltage rise du / dt, the IGBT switches off and is only marginally with current loading (see. Associated current / voltage profiles in Figures 1A and IB).

The results of the influence of overvoltage and reverse current peak by briefly switching of the freewheeling diode 7 '¹ lying parallel IGBT (load driver 3¹') Is represented by the current / voltage profiles in Figures 1A and IB. Here are the current / voltage profiles at a switching-on IGBT (load driver 3 ') in Figure 1A and the current / voltage profiles on one, the other, blocking IGBT

(Load driver 3 '') parallel freewheeling diode 7¹'Shown in Figure IB.

If the gate voltage V_GE turned on the non-switched (of certain sermaßen locked) IGBT briefly, thus increasing the gate voltage V_GE within a short period of time i pulse ör ig. This leads to a significant reduction of the voltage overshoot of the intermediate circuit voltage or supply voltage V_cc compared to the prior art (Stro - / voltage conditions of the prior art are shown in dotted lines; see FIGS. 8A and 8B). Moreover, the Current rise di / dt of the switched load current I_G smaller, but this leads to more switching losses.

As is clear from the comparison of the dotted and the solid current curves of the load current I in Figures 1A and IB, the switching losses are, however, this increases only marginally. The losses in the momentarily switched IGBT are negligible due to the low current connected.

In order to reduce the current gradient or the overvoltage clearer, for each pair of IGBT freewheeling diode 3, 7, a feedback path 11 (Figure 2) may be provided that the voltage at the cathode of the freewheeling diode 7 '' (or at the collector of IGBT) to control input of the IGBT feeds back (the control signal V_gated_r-._ve which is present at the output amplifier stage 9 of the gate driver GD, is affected), whereby a briefly switching the locked se IGBT is effected.

In the feedback path 11 advantageously Zener diodes can be arranged (in the embodiment of Figure 2 is a 30V zener diode 12, which is connected in antiparallel with a clamp Zener diode 13 in series). These have the effect that only upon reaching of a defined over-voltage at the freewheeling diode 7 '¹ turning on the parallel thereto lying, is triggered locked themselves IGBTs (a so-called over-voltage clamping).

With the help of the Zener diodes 12, 13 can overvoltage greater than the clamp voltage by feedback in the output amplifier stage 9 of the gate driver GD and thus an influence of the control signal upon reaching a Ü V_gated_rιe a brief at least partially switch the IGBTs can be effected.

Unlike an immediate feedback to the Ga te terminal of the IGBT, the Zener diodes in the reports will be Circuit loaded with much lower currents. In this way, a similar switching behavior sought, as is already represented by the current / voltage profiles in Figures 1A and IB.

Instead of the two Zener diodes 12, 13 can in the feedback path 11, an external capacitor C_ext be used, as shown in Figure 3A. It is the rise in the cathode voltage of the freewheeling diode 7 '' on the control terminal of the parallel with the freewheeling diode 7 '' connected IGBT fed back, whereby the IGBT is momentarily and at least partially turned on, which counteracts the voltage overshoot and the peak reverse recovery current.

The combination of those two measures may be used to advantage: For using an external capacitor C_ext in the feedback path 11 of the control terminal of the power driver during the voltage rise at the freewheeling inrichtung be biased at a voltage below the turn-on voltage. In this case, the parallel feedback (eg via the described Zenerdiodenschal- direction) occurs much trägheitsf eggs at a threshold voltage is exceeded, as to turn-of the load driver a lesser voltage excursion has to be overcome.

Similarly, a control signal may be proportional to the amount of the current gradient in the freewheeling device to the control terminal of the load driver via a feedback to development branch 14 is fed back (for example by means of a transformer U, an amplifier AMP and an evaluation circuit CC) (Figure 3B). The current gradient at the fall of the current in the freewheel device after exceeding the maximum of the peak reverse recovery current is fed back to the control terminal of the load driver, so that it momentarily at least partially aufsteuert and limits the current drop in the peak reverse recovery current. This measure can with the up or one of the measures described below are combined to bias the control terminal of the load driver. In this way can be reduced to the free-wheeling means in simple surge.

With suitable dimensioning of the current gradient can be used in the freewheel device before reaching the maximum peak reverse recovery current to bias the control terminal of the load driver vorzu- to a voltage below the threshold. This method offers the advantage that the increase in the control voltage at the load driver is automatically synchronized to the commutation process.

The detection of the current gradient in the simplest case as a voltage drop across the stray inductances _σi_π L_σext done. Similarly a transformational acquisition is o- of the detection and evaluation of the current with appropriate sensors possible. The change of sign of the current gradient makes this possibly a rectification or evaluation of the signal in a case distinction dl / dt> <0 is necessary.

In this way the slope of the voltage rise at the freewheeling diode 7 '' specifically controlled and continuity of return current peak and consequently the surge can be set. The size of the set voltage rise du / dt is set by the systematic selection of the capacitance of the capacitor Ce and does not depend - as in the first embodiment - of which saturated during Schaltvorgan- modified value of the Miller capacitance from. In this way, a similar switching characteristics are effected as it is already represented by the current / voltage profiles in Figures 1A and IB.

Instead of the feedback branch 11 or in addition to the feedback path 11 between (cathode of the) freewheeling diode 7 and control terminal of the IGBT can also targeted, the time point of the shifting operation made control of the IGBT via the control signal V_gatedrive be made. Here is the parallel with the freewheeling diode 7 '¹ lying IGBT specifically dependent through a diode from the date of the peak reverse recovery current of the freewheeling 7 '' control signal V_gated_rive (Figure 4) within a certain time window during commutation turned on (at least partially turned on) and then switched off again (blocked), and this at a time when the actual load-controlled control signal Vgatedrive is not intended for this IGBT.

The timed to the peak reverse recovery current control signal V_gatedrive raises the gate voltage _GE at the freewheeling diode 7 '' in parallel and to be locked for IGBT during the commutation process targeted at briefly. Thus, the IGBT controls at least by some.

The current / voltage ratios with such control are exemplary in the upper part of Fig 4 in comparison to the 4 dots illustrated history of 8A and 8B reproduced. The load current I_c decreases during blocking of the IGBT and has a peak in the negative range or elevation with a smaller current steepness di / dt compared to the current profile on, as viewed from the prior art (dotted line shown current waveform of the load current I_c) Results. The intermediate circuit voltage V_Cc shows a significantly smaller overshoot or lower overvoltage (smaller peak), since the supervisors steering of IGBT overvoltage counteracts targeted.

Instead of a stepped control signal V_{gate drive} and an associated temporary additional increase of the gate voltage V_GE can also be a square-wave control signal Vgatedrive and an associated, higher mean gate voltage V_GE (Figure 5) compared to the gate voltage V_GE be provided in FIG. 4 The gate voltage V_GE the parallel to freewheeling diode 7 '¹ lying, actually blocking IGBT during the commutation of the current from the freewheeling diode 7 in a predetermined, short time window during a peak reverse recovery current of the load current I_c occurs at the free-wheeling diode 7, to a value smaller than a threshold voltage raised.

Similar to the first embodiment may here also without high-ohmic gate-emitter junction of a switching as a result of the voltage rise du / dt at the free-wheeling diode 7 and the feedback capacitance (capacitor C_ext) Done.

Advantage of this method is that the voltage interval is reduced by turning on the IGBT and thus the delay time can be minimized by the action of the feedback voltage rises dv / dt.

The timed to the peak reverse recovery current control signal Vgaed_rive raises the voltage at the IGBT in parallel with the freewheeling diode 7 '' in a certain time window. This example, an at least partial supervisors steering of IGBT is made possible by the feedback capacitance. The resulting switching behavior (current / voltage curve) is in the upper part of Figure 5 in comparison with the behavior of the figures 8A and 8B shown.

The gate voltage V_GE will tion less increased than the previous exemplary embodiment, the constant large actuation. The collector voltage V_ceHowever / voltage at the freewheeling direction behaves similarly to the previous exemplary embodiment, while the load current I_c in its current steepness di / dt is slightly less reduced, but the reverse current spike is still significantly less steep than the comparable reverse current spike in the prior art (see. dotted line for the load current I_c in Figure 5). To limit the voltage at the freewheeling diode 7 during switching operations, also circuits with feedback path 11 with additional, specific control of the gate voltage V can_GE the IGBT can be combined. Also, the high-resistance connection of the gate and the emitter of the IGBT in the off - as explained in the first embodiment - a feedback path 11 or a targeted, short-term control of the gate voltage V_GE be combined. Thus, a simple and effective reduction of overvoltage and / or the reverse current spike at the freewheeling diode 7 is given. The freewheeling diodes 7 are thus largely protected against destruction.

Freewheeling diode for limiting an inductive overvoltage are well known. Also equivalent sprags are functionally known which are not necessarily effective with only a single, a circuit breaker connected in parallel diode. For the invention it is immaterial whether a freewheeling diode or a equiva- lentes agent is used as a freewheeling device. Much, however, is that an inductive surge and a large current rise can be reduced according to the inventive method or the inventive circuit arrangement.