Method for the correction of phase and amplitude errors in COFDM signals

The invention relates to a method for correcting the phase and amplitude of a broadband received signal by means of reference signals according to the preamble of patent claim 1 and a circuit arrangement for carrying out the method.

From the publication "Trends in television broadcasting", published in telekom practice, issue 12/92, pages 32 to 38, it is known that work is being done on a digital terrestrial television system worldwide. For digital transmission, the COFDM method (Coded Orthogonal Frequency Division Multiplex) is to be used in this television system. This achieves a higher transmission quality with more efficient use of the frequency ranges used for transmission.

The COFDM transfer method is known from WO 88/00417. The information to be transmitted is first converted analog-to-digital if the information is analog. Subsequently, the digital information is encoded with a channel encoder, whereby an error protection code is added to the information. Finally, the channel coded information of a plurality of single carrier frequencies is modulated, scrambling the temporal order of the information. This achieves increased protection against time- and frequency-selective interference.

From WO 92/10043 a method for frequency control in local oscillators of a receiver is known. The received digital television data are transmitted using the OFDM (Orthogonal Frequency Division Multiplex) method. In addition to the digital television data, two pilot signals are still transmitted in each transmission frame, which have a constant frequency difference. In order to control the frequency of the local oscillators in the receiver, the beginning of a transmission time slot is detected, then the position of the two pilot signals in the transmission frame is determined and then the phase difference between the two pilot signals is calculated as a function of time. From this, a control variable for the local oscillators can then be derived.

From the not previously published application P 42 40 609.9 of the Applicant a method for frequency response correction is known. It is used for terrestrial transmission of digital signals (DAB). In this case, frequency-selective amplitude and phase distortions of a differential-phase-modulated COFDM transmission signal are detected by a control unit, a corresponding correction characteristic is calculated and the spectrum of the transmission signal is weighted with the correction characteristic. It is irrelevant whether the detected distortions have arisen on the transmission link or in transmitter or receiver modules. The method also allows the reliability of the received signals to be estimated.

A disadvantage of this method is that a high computational effort is required to calculate the correction characteristic with time and frequency strongly fluctuating distortions.

Another disadvantage is that the method can correct only a portion of the possible distortions in non-differential phase modulated signals.

EP 0 365 431 A1 discloses a method for correcting the phase and amplitude of a broadband received signal. In this method, a symbol-wise transmission of the broadcast signal. The signal transmission is added a guard interval.

It is therefore the object of the invention to specify a method for equalizing a non-differential-phase-modulated COFDM transmission signal. The required computational effort should be as low as possible.

This object is achieved by the characterizing features of claim 1.

D method according to claim 1 has the advantage that due to the reference signal transmitted in each transmission frame, the amplitude and phase distortions of the data signals can be interpolated as a function of time and frequency for the entire transmission frame. This calculation can be for each Single carrier frequency or for a group of several single carrier frequencies of a transmission frame are performed. Another advantage is that a maximum likelihood decision is made regarding the amplitude and phase of each single carrier frequency. As a result, small phase or amplitude distortions can be rectified immediately, with large distortions it is recognized that the reliability of the received signal decreases.

Furthermore, the method has the advantage that the amplitude correction adapts to changed reception conditions. By taking into account a number of recently received signals, a steady course of the equalized received signal amplitude is achieved.

The method according to claim 2 has the advantage that due to the known time profile of the amplitude and phase distortion and amplitude and phase correction can be calculated according to a correction characteristic of higher order.

The method according to claim 3 has the advantage that the amplitude correction and the phase correction can be performed by a single multiplication of the short-term spectrum of the received signal with the correction characteristic.

The method according to claim 4 has the advantage that no Fourier transform and Inverse transformation is required especially for the frequency-selective correction of amplitude and phase of the received signal. As a result, considerable computing and circuit complexity can be avoided.

The use of the method according to claim 5 has the advantage that an equalization of television broadcasts is made possible. It is furthermore advantageous that the correction values ​​for the amplitude correction can be easily calculated, since the amplitude of the spectrum ideally assumes discrete values ​​in the case of a transmission of digital television data according to the COFDM method.

The circuit arrangement according to claim 6 has the advantage that the calculation of the correction characteristic, the correction of the individual carrier frequencies and the maximum likelihood decision can be performed by a control unit, which must be provided anyway for controlling the receiver. The additional circuitry required to correct the received signal is thus limited to a relatively inexpensive and highly integrated memory.

The invention will be described and explained in more detail below. It shows:<dl id="dl0001"><dt>Fig. 1</dt><dd>a possible course of the single carrier frequencies used for transmission according to the COFDM method as complex pointers,</dd><dt>Fig. 2</dt><dd>a possible phase distortion of a single carrier frequency,</dd><dt>Fig. 3</dt><dd>possible phase differences of the phases of received individual carriers and of the phases originally modulated on the individual carrier frequencies,</dd><dt>Figure 4</dt><dd>possible phase differences of the phases of received individual carriers and the phases originally modulated on the single carrier frequencies after a phase correction and</dd><dt>Fig. 5</dt><dd>a circuit arrangement for carrying out the correction.</dd></dl>

In the following embodiment, the COFDM method is used to transmit digital television data. The single carrier frequencies are modulated with the TV data 4-PSK (Phase Shift Keying). In the receiver, a reference signal REF with a reference phase PR and then a data block with n data signals D1 to Dn which are phase-modulated with respect to the reference phase PR in each transmission frame after a synchronization signal SYNC for timing synchronization of the receiver, in which the amplitudes of all individual carrier frequencies are zero , received. Due to the 4-PSK modulation of each single carrier frequency can with each single carrier frequency two binary states, eg 01, so a dibit be transferred.

Fig. 1 shows the transmission signal, wherein the transmitted single carrier frequencies are shown as complex pointers with amplitude and phase as a function of time t and frequency f. The frequencies f₁ to f_m are the single carrier frequencies of the COFDM transmission method, which are arranged equidistantly on the frequency axis f. The signals SYNC for the receiver synchronization, REF for the transmission of the reference phase PR and D1 to Dn for the transmission of the television data are arranged equidistantly on the time axis t and form a transmission frame. Due to the phase modulation of the data signals D1 to Dn, the individual carrier frequencies can be arranged out of phase with the modulation-free reference phase PR of the reference signal REF, which is expressed by corresponding rotations of the complex hands. The reference phase PR is reset for each transmission frame by the reference signal REF. Therefore, the reference signal REF may also have no phase distortion. It is not necessary that all individual carrier frequencies of the reference signal REF have the same reference phase PR. The data signals D1 to Dn are shown in FIG. 1 mostly phase distortions. These may be caused by the transmission link and / or by transmitter and receiver assemblies. For the sake of clarity, the representation of possible amplitude distortions has been dispensed with.

For demodulation in the receiver, the phase difference between the transmitted reference phase PR and the phases of the single carrier frequencies of the same transmission frame is calculated. If this phase difference is then decoded into binary values, the transmitted information is returned.

Due to distortions, the phase difference in a 4-PSK modulation is not exactly an even multiple of 90 ° but, for example, 100 °. In order to be able to decode binary states from the phase differences even in the case of distortions that are so low, a maximum likelihood decision is made.

This corrects small phase deviations and the transmission system has some margin of error. For example, phase values ​​in an interval between 45 ° and 135 ° in the receiver can be assigned a phase of 90 °. FIG. 2 illustrates this graphically. The ideal phase Pi, which is detected in the maximum likelihood decision, does not correspond to the phase position Pe of the received single carrier frequency. Dashed lines indicate possible limits for the maximum likelihood decision.

FIG. 3 shows the phase deviation P of the received single carrier frequencies from one of the four possible phase angles at 4-PSK module. By definition, the reference signal REF of a first transmission frame received at time t1 has no phase distortion. The phase distortions of the data signals D1 to Dn are shown frequency-dependent. Since during the synchronization signals SYNC all single carrier frequencies have the amplitude zero, no signal can be detected. The reference signal REF of the second transmission frame defines the reference phase PR for the following data signals D1 to Dn, assuming that the reference phase PR is time-invariant. Nevertheless, the reference phases PR (t1) and PR (t2) received in different transmission frames may have a phase difference P due to phase distortions. This phase difference is shown in FIG. 3 for the reference signal REF, which is received at the time t2 in the second transmission frame, shown frequency-dependent.

If deviations ΔP of the received phases occur, in particular due to time-dependent parameter fluctuations, as shown in FIG. 3, which are not due to modulation and are not within the correctable error tolerance described above, errors occur during decoding despite fault tolerance. To avoid this, a phase equalization is performed on the received signal. In this phase equalization, a control unit calculates the phase difference d_PR between at least two reference phases PR (t1) and PR (t2) of the same single carrier frequency. As a rule, reference signals REF of temporally successive transmission frames according to (1) are evaluated for this purpose.<maths id="math0001" num="(1)"><math display="block"><mrow><msub><mrow><mtext>d</mtext></mrow><mrow><mtext>PR</mtext></mrow></msub><mtext> = PR (t1) - PR (t2)</mtext></mrow></math><img file="EP0618697B1_D0001.tif" /></maths>

Is this phase difference d_PR amount greater than a threshold value, it is recognized that a phase equalization is required, and a quotient Q of the phase difference d_PR and the time dTR which lies between the reception of the two reference signals REF (t1) and REF (t2), according to (2).<maths id="math0002" num="(2)"><math display="block"><mrow><mtext>Q = </mtext><mfrac><mrow><msub><mrow><mtext>d</mtext></mrow><mrow><mtext>PR</mtext></mrow></msub></mrow><mrow><mtext>dTR</mtext></mrow></mfrac></mrow></math><img file="EP0618697B1_D0002.tif" /></maths>

This quotient Q now indicates the linear time-dependent change of the reference phase PR. Thus, an expected linear phase error Pfi according to (3) can be calculated for each data signal Di as a function of the time of its transmission.<maths id="math0003" num="(3)"><math display="block"><mrow><mtext>Pfi = Q • (T (REF, t1) - T (Di))</mtext></mrow></math><img file="EP0618697B1_D0003.tif" /></maths>

If the thus calculated phase errors Pfi are subtracted from the phases of the Einzelträgerfrequenzn of Fig. 3, linear time-dependent phase errors can be eliminated.

FIG. 4 shows the phase distortions Pn adjusted by the calculated fraction, which can be corrected on the basis of the subsequent must-likelihood decision, so that no transmission error occurs. Here, the phase correction can have quite different effects. As an example, consider the single carrier frequency f1. As shown in FIG. 3 this single carrier frequency has no phase distortion ΔP for the data signal D1 and a low phase distortion ΔP for the data signal Dn. The phase distortion .DELTA.P of the reference signal of the second Übertragungsrahmnens REF (t2) already has an average phase distortion compared to the reference signal of the first transmission frame REF (t1). The phase distortion ΔP of the single carrier frequency f1 thus increases with time t, but not linearly. Due to the linearly calculated phase correction Pfi, a phase distortion Pn therefore results for the single carrier frequency f1 even after the phase correction, which, however, is lower, as shown in FIG. 4 shown. For the data signal D1, a phase distortion Pn arises, which lies within the error tolerance and therefore does not lead to a transmission error. For the data signal Dn, there is no phase distortion Pn of the single carrier frequency f1, since the originally present phase distortion P can be eliminated by the linear correction.

The time dependence of the phase distortion P can be analyzed by a control unit MP, so that an optimal mathematical model for the calculation of the phase correction values ​​Pfi for interpolating the phases of the data signals D1 to Dn can be found, which of course need not be linear.

It is also possible to include reference phases PR of previously received transmission frames in the calculation and to perform a time-dependent weighting of these reference phases PR. This makes it possible to take into account temporally nonlinear phase distortions. The equations (1) to (3) are then to be modified accordingly.

In order not for each single carrier frequency f₁ to f_m To be able to perform the calculation described above, it is conceivable, for example, to carry out the above calculation only for every second or every 16 th individual carrier frequency and to interpolate missing quotients Q from the calculated quotients Q as a function of the single carrier frequency. As a result, channel capacity can be made free in the reference signal REF, which can be used to transmit additional data.

In accordance with the above-described phase correction, the amplitudes of the single carrier frequencies are also corrected. The amplitudes of the reference signals are buffered in a memory SPE, the amplitudes of the single carrier frequencies are reset due to the interpolated amplitude correction values. The interpolation of the amplitude can take place both as a function of time and of the frequency.

From the frequency-dependent amplitude and phase correction, a complex correction characteristic is then calculated, with which the short-term spectrum SP of the received signal is multiplied. The correction characteristic is composed of the amplification value for the amplitude and the phase correction value Pfi for each individual carrier frequency. The multiplication can be carried out at a point in the signal path at which the short-term spectrum already exists, for example after the Fourier transformation FFT.

Alternatively, a correction pointer can also be calculated for a single carrier frequency, which results from the difference pointer between two reference pointers of a single carrier frequency. This is multiplied by a scale factor in response to the time of reception of the signal to be corrected and added to the originally received pointer to obtain the phase and amplitude corrected single carrier frequency hand.

In order not to have to recalculate this correction characteristic for each transmission frame, it is stored in a memory SPE and can be used to correct data signals of several transmission frames. A recalculation of the correction characteristic can be initiated by the user or the control unit MP. For this purpose, the control unit MP evaluates the equalized amplitudes and the equalized phases of the single carrier frequencies. As soon as the amplitudes or phases of the reference signals REF fluctuate greatly, a recalculation of the correction characteristic is required. FIG. 5 shows a circuit arrangement for performing the correction of phase and amplitude of the received signal. After the digital received signal of a first transmission frame is present as a short-term spectrum SP by a fast Fourier transformation FFT in the spectral range, a control unit MP evaluates this short-term spectrum SP. The reference signal REF and the data signals D1 to Dn are latched in a memory SPE. Likewise, the signals of a second transmission frame transmitted after the first are latched in the memory SPE. Subsequently, the complex correction characteristic curve is calculated by the control unit MP from the reference signals REF of the two transmission frames, which is likewise stored in the memory SPE. After the single carrier frequencies of the first transmission frame have been multiplied by the correction characteristic, the single carrier frequencies of the maximum likelihood decision are supplied by the control unit MP and finally further processed in the demodulator DMOD. If it is determined in the maximum likelihood decision maximum likelihood decision that the phases of the received single carrier frequencies deviate greatly from the possible modulation phases (with 4-PSK modulation: 90 °, 180 °, 270 ° and 360 °), the Information recognized as unreliable.

Subsequently, in the memory SPE, the received data of the first transmission frame is overwritten by the reception data of the following third transmission frame. The received data of the second transmission frame are then multiplied by the previously calculated correction characteristic and fed to the maximum likelihood decision. If it is determined by the control unit MP that the amplitude fluctuations or the phase distortions are large in this decision, a new correction characteristic is calculated by the control unit MP. This new correction characteristic is calculated as a function of the reference signals REF of the second and third transmission frames. The new correction characteristic replaces the previously stored data in the memory SPE and is used to correct the received data after the first transmission frame. As soon as required, a new correction characteristic is calculated again.

If nonlinear distortions of the phase are also to be corrected, previously transmitted reference signals REF are to be stored in the memory SPE and taken into account by the control unit MP in the calculation, in particular the phase correction values.