Device for distance measurement

The invention relates to a device for measuring the distance of the generic type specified in the preamble of claim 1.

In such devices, the coil is typically the resonant circuit coil of a high frequency oscillator, it provides the signal dependent on their damping high-frequency signal. The damping is based on the effective resistance of the coil, which is partially caused by the eddy current losses in the object, the back acting on the coil, and in the other by the DC resistance of the coil, which depends on the temperature of the coil.

The advantage of the distance measurement devices of the present kind essentially consists in that it is sufficient to arrange only the coil at the measurement location, while the oscillator and the evaluation circuit can be operated at a certain distance from the measuring point, in particular at room temperature, while the coil at the measuring location of a higher (or lower) temperature is exposed, in which the other parts of the device could not be operated, especially if they are fitted with semiconductor elements.

At considerably different from the room temperature temperatures, however, the influence of temperature on the proportion of the attenuation which is caused by the direct current resistance of the coil, no longer negligible, so that the high frequency signal to both the distance of the object from the coil as well as the temperature thereof depends. For example, the DC resistance of the coil increases with an increase in temperature of 250^OC to about twice.

In order to avoid the dependence of the obtained from the high frequency signal interval signal from the temperature, has been proposed, by means of a coil supplying direct current source to generate by the coil temperature dependent DC signal and use for compensating the influence of temperature on the distance signal.

The direct voltage signal can be obtained as a voltage drop of a current flowing through the coil constant DC current.

From EP-A-0049304 an apparatus of this kind is known in which an image obtained from the high frequency signal and a signal obtained from the DC signal are fed to a summing element which supplies a sum signal, the distance signal. When in this known apparatus the temperature of the coil changes, the direct current signal and the signal obtained therefrom, the summing supplied change. The result is a corresponding change of the obtained sum signal from the summing element distance signal, even if the distance of the coil remains unchanged from the object, that is, the distance signal is changed at a constant distance with temperature.

The invention, as characterized in claim 1, the object is achieved, to provide a device for distance measurement with improved temperature compensation.

In the inventive device, the lost in the temperature-dependent direct current resistance of the coil going energy is always covered from the corresponding temperature-controlled high-frequency current source so that the light generated by the coil alternating magnetic field is independent of the temperature.

From the Japanese patent application no. 67819/1985 an inductive distance measuring device is known, which has a four-pole distance sensor and a drive circuit having a temperature compensation. The distance sensor is designed in the manner of a small transformer, which has on the primary side and an excitation coil on the secondary side of two differentially-connected detector coils. The meter has two operating states. In a distance is detected and in the other carried out the temperature compensation. In the distance measurement is given to the exciting coil a high-frequency alternating current. the distance-dependent coupling degree can then remove and evaluate with a circuit on the secondary side coil. To compensate for the temperature to which the primary side stimulating oscillator is decoupled and the secondary side connected to a resistance measuring circuit having appropriate switches. The determined resistance value is compared with a reference value and the difference signal is used to adjust the voltage of the oscillator. For slow temperature changes must be rare switched and strong temperatures correspondingly often.

As mentioned above, in the inventive device, the magnetic, hochfrequenzte alternating field of the coil is independent of the temperature. The attenuation of the coil through the object always under the same conditions will be affected, and there is a Hochfreqeuenzsignal that depends only on the distance of the object from the coil.

In the following the invention including specific embodiments will be explained in detail with reference to only one possible embodiment representing, accompanying drawings. Show it:<dl id="dl0001"><dt>Fig. 1</dt><dd>a block diagram,</dd><dt>FIG. 2</dt><dd>the detailed circuit diagram of an inventive device, and</dd><dt>FIGS. 3-5</dt><dd>each one measuring head.</dd></dl>

With the device a distance a from an object 1 of a high-frequency excited coil L1-dependent signal is generated. For this purpose, the object 1 is arranged on one end side of the coil L1 and, if it is not self-electric (or magnetic) conductive, provided these properties with a (not shown) lining. As in principle of Fig. 1 and in detail in Fig. 2, the coil L1 is the oscillating circuit coil of a high-frequency oscillator 5 whose high frequency voltage is located on a line 7 is connected to a diode D1 with subsequent smoothing member 12. Since the reaction of the induced in the object 1 eddy currents increases the coil L1, the damping of the oscillator resonant circuit in the distance a dependent mass, thereby reducing the high-frequency voltage, the output voltage of the smoothing stage 12 is a dependent of the distance a signal.

The damping of the resonant circuit also depends on the DC resistance of the coil L1, which is temperature dependent. So that additionally, the output voltage of the screening member 12 also depends on the temperature of the coil L1.

To compensate for the influence of temperature on the output voltage of the screening member 12, the current flowing through the coil L1 high-frequency current, a direct current of a constant current source 9 is superimposed, and the voltage drop of the DC current to the coil L1, is as an input variable to a control circuit 10, 14, 6 which controls the oscillator supplying the direct current depending on the DC voltage drop and thereby counteract temperature-induced changes in the high frequency voltage.

For this purpose, the constant current source 9 is connected to the line 7 through a high-frequency choke coil L2, holds off the high frequency from the constant current source 9 and the control circuit has an inverting amplifier 10, a resistor network 14 and a controllable current source 6, which the oscillator 5 supplying DC supplies. The input of the amplifier 10 is through a high frequency lock-forming RC circuit R1, C1 connected to the line 7, whose output voltage is at the input of the resistor network 14 whose output is connected to the control input of the current source. 6 If the coil temperature increases, the DC voltage drop across the coil L1, which is located at the input of the amplifier 10, too. In this case, the output voltage and the control voltage of the controllable current source 6 increases, and an amplifier O4 (Fig. 2) of the power source 6, which supplies the supply direct current of the oscillator 5, causes an increase of the supply current and thus the excitation of the coil L1. With a decrease in coil temperature, the processes are correspondingly reversed. Through the network 14 of the circuit parts 5, 6, 10 is achieved at given in the rest of the data that the amount by which the control circuit 10, 14, 6, the high frequency excitation of the coil L1 with a change will change the temperature of which is equal to the amount by to the excitement would change if the control circuit 10, 14, 6 were not present. As a result, the influence of the temperature of the coil L1 is compensated on the radio frequency excitation, so that the high frequency voltage and the voltage corresponding to this signal at the output of the screening member 12 depends only on the distance of the object 1 from the coil L1.

In this way, the oscillator 5 supplying power source 6 to compensate for the temperature dependence of the DC resistance of the coil L1 by the dependent of this signal at the output of the amplifier 10 is controlled.

If the distance a becomes larger at a constant temperature, increases the output voltage of the smoothing member 12 that the input of the amplifier 13 is positive, the output voltage of amplifier 13 increases, whereby the oscillator 5 supplying current is weaker. As a result, an increasing at an increasing distance a signal on the output line 17 is obtained which is independent of the temperature.

The dependence of the high frequency voltage to the coil L1 on the distance a is not linear. This high-frequency voltage decreases with increasing distance a first steeply and then gradually less steep to.

A signal which depends practically linearly on the distance a can be obtained by a non-linear transmission member out of the way of the diode D1 rectified and smoothed in the smoothing element 12 high frequency voltage, the output voltage of first less steeply and then more steeply increases with increasing input voltage. Semiconductor devices with barrier layer, in particular diodes, are known to have a smaller range voltages with increasing voltage first less steep and then more steeply rising current-voltage characteristic. Thus, a non-linear transmission member 15 to receive the rectified and smoothed high-frequency voltage, which supplies a signal whose function of the distance a is linear.

The characteristic of a diode is subject to fluctuations in temperature and in the range of room temperatures and significant changes. Therefore, in the circuit of Fig. 1 and 2, a non-linear element 15 of special, in more detail hereinafter described type, in which the temperature dependence of the diode characteristic is compensated for, provided that works together with an amplifier 16, of the desired signal at the output 17 of the circuit arrangement supplies. Here, the dependence of this signal from the distance a linear, and the signal is, as explained previously, is independent of the temperature of the coil L1, and also the influence of temperature on the decisive for the linearization of a diode characteristic, as explained further below, offset ,

In the following, details of the circuit portions of the circuit arrangement will be explained in detail with reference to FIG. 2.

The current supply of circuit parts 5, 6, 9, 10, 12, 13, 14, 15 and 16 carried out on a line 19 to, for example, to approximately 17.2 volts and a conduit 20 with, for example 6.3 volts with respect to ground 22. This voltages are stabilized in principle a conventional manner by means of a Darlington circuit T1, T2, a differential amplifier O1 and two zener diodes Z1 and Z2.

The high-frequency oscillator 5 is a Colpitts oscillator in base circuit comprising a transistor T4 and the through the coil L1 and two capacitors C2 and C3 formed resonant circuit that is on the one hand connected to the collector of the transistor T4 and the high-frequency conductive line 7 and on the other hand to ground 22 is.

The smoothing element 12 comprises for smoothing rectified high frequency in a known manner two capacitors C6, C7 and a resistor K1, which is performed for the compensation of influences of temperature on the circuit configuration (without coil L1), in particular of the semiconductor diode D1 as a PTC thermistor.

The constant current source 9 is designed in a known manner with an operational amplifier O2, a transistor T5 in the collector circuit and five resistors R9 to R13. It provides the constant direct current to the high-frequency lead, connected to the line 7 the end of the coil L1. This current is, for example, such that the DC voltage drop across the coil L1 at room temperature for about 0.2 volts. Here is also flowing through the coil L1, regardless of the distance a necessarily changing collector current of transistor T4. Also by resistors R1, R16, R19, R20 / 21 and the coil L1 current flowing here is not significant, it is also stabilized in the rest because of the stabilized voltage on the line 19, albeit with greater deviations than the current of constant-current source 9, which in turn is supplied with this voltage.

The controllable current source 6 contains as an actuator for the DC power supply of the oscillator 5 comprises an operational amplifier O4, and resistors R24 to R29. Here, the output voltage of the resistor network 14, the manipulated variable at the input of the operational amplifier O4.

The amplifier 13 is an operational amplifier O5 whose non-inverting input connected by a resistor R 36 to the output of the screening member 12 and the inverting input of a voltage divider R 32, R 33 is energized.

The resistance network 14 includes a resistor 23 connecting the output of the amplifier 10 to the control input of the current source 6, a resistor R24 ​​is still connected upstream. Through the resistors 28 and 42, the required bias voltage at the inverting input of the amplifier O4 is obtained. The resistors R23 and R42 together with the effect resistor R28 that is dependent on the temperature of the coil L1 voltage at the control input of the current source 6 is as large as required for a complete compensation possible.

The described compensation of the influence of temperature on the high-frequency voltage of the coil L1 through the temperature-dependent direct current resistance of the coil L1 is sufficient, as has been shown, no more, when the coil L1 is exposed to extreme temperatures, for example 24 ^OK or 300 ^OC. Then, additional, other temperature factors come to the attenuation and thus the high-frequency voltage of the coil L1 to the effect. For example, it could be to dielectric losses in the insulation of the coil wire or by eddy current losses in the (not shown) spool. To take into account these additional effects at the intended temperature compensation of the measuring head containing the coil 23 can be equipped according to Fig. 4 with a network D3, R60, C12. The appropriate network and its elements, in particular the nature of the diode D1 and the resistance value and the (temperature-dependent) material of the resistor R60 or R61 can be determined by tests. However, the resistance value of R60 or R61 is not more than 5%, not exceed 10% of the DC resistance of the coil L1. The capacitor C12 and C13 should be such that is its impedance is negligibly small at the frequency in question compared to the resistance of the series circuit D1, R60 or the resistor R61. Its capacity must be a multiple of the capacitance of C2 and C3. As shown in Fig. 4 and 5, is the additional network D5, R60, C12 and R61 C13 connected in series with the coil L1, so that the measuring head also in these embodiments has only two terminals, and the compensation with a single DC voltage takes place, ie the direct current source 6 of the oscillator is controlled DC voltage 5 contained only by an all temperature-related influences. In principle could of course also be subjected to a only dependent on the DC voltage and a certain resistance through the network voltage on the DC power source. 6

The non-linear element 15 comprises two parallel-connected branches, each of which consists of a diode-connected transistor T6 or T7 in series with a resistor R49 or R50. Transistors are used, because such a pair, is commercially available in a housing on a semiconductor substrate, which keeps both transistors at the same temperature, and as described further below, one such pair for temperature compensation of an electrical signal is required. The branches T6, T7 and R49, R50 are in Fig. 2 at the output of an operational amplifier O6 whose inverting input is connected to the line 20, and whose inverting input is connected to the base and the collector of the transistor T7. This operational amplifier 06 is basically not necessary, it causes only a further improvement of the effect of the nonlinear element 15. Therefore, the operation of an embodiment is described in which the branches T6, R49 and T7, R50 connected directly (without operational amplifiers) to the line 20 are.

Both versions have the transistors T6 and T7 are the same characteristics and at rest are at the resistors R 49 and 50 identical voltage drops so that at rest are mutually equal voltages across the diode-connected transistors T6 and T7. The output of amplifier 13 is connected through a resistor 44 to the connected to the base and collector of the transistor T6 end of the resistor 49, so that the output voltage of this amplifier 13 increases the voltage across this resistor 49th This increases the voltage to the transistor T6, so that its operating point moves on its current-voltage characteristic and the current supplied by it to the resistor R49 current corresponding to the characteristic decreases more than proportional to the output voltage of the differential amplifier 05th This smaller changes in the output voltage of the differential amplifier 05 (circuit part 13) is relatively less, larger changes counteracted relatively stronger. The degree of this effect, contrary depends on the position of the working point in the idle state and from the resistor 49 and can be dimensioned such that the dependence of the voltage across the resistor R49 by the distance a is substantially linear.

The current-voltage characteristic of the diode-connected transistors T6 and T7 is already a function of temperature in the range of ordinary room temperatures. This temperature effect is compensated by the fact that the transistor T6 facing end of the resistor 49 through a resistor R53 to the noninverting input and the transistor T7 facing end of the resistor R50 through a cold head K2 and a resistor R52 to the inverting input of an operational amplifier O7 is connected, the output of which the desired, provides the distance a linearly dependent signal at the output 17 of the circuit arrangement. In this case this signal is determined by the difference of the voltages across the resistors R49 and R50. The voltage-to R49 is emerged from the high-frequency voltage across the coil L1 and linearized by means of the transistor T6 in terms of distance a. The voltage across R50 is a (the distance a independent) auxiliary voltage. Both voltages associated with a small deviation in agreement on the temperature of the transistors T6 and T7, so that their difference, which is the linearized output voltage to 17 (except for the small deviation) is independent of temperature. This deviation arises because the two voltages are not equal, and therefore the operating points of the diode-connected transistors T6 and T7 are in different places whose characteristic. To compensate for this deviation, the PTC K2 serves.