One frequency repeater for a digital microwave radio system with cancellation of transmitter-to-receiver interference

Background of the Invention

1. Field of the Invention

The present invention relates to a repeating system for a digital radio wave transmission, and in particular, to cancellation of a transmitter-to-receiver interference at a repeater in a repeating system wherein the same frequency carrier pair is used for both transmitter and receiver.

2. Description of the Prior Art

In a digital radio wave transmission system, a carrier, usually a microwave modulated by a digital baseband signal to be transmitted is transmitted as a digital microwave radio signal. A repeater used in the digital microwave transmission system has at least one, usually two repeating paths. Each repeating path comprises a pair of receiving and transmitting antennas. The digital microwave is received at the receiving antenna and is demodulated. The demodulated signal is applied to a decision circuit, which decides and regenerates the digital baseband signal. A carrier is modulated by the regenerated digital baseband signal at a transmitter, and the digital microwave signal is radiated from the transmitting antenna.

The receiving antenna of one repeating path and the transmitting antenna of the other path are mounted side by side on a front face of the repeater, while the transmitting antenna of the one repeating path and the receiving antenna of the other being arranged side by side on the opposite back face of the repeater.

In the repeater, transmitting signals from the two transmitting antennas may interfere the desired signal received at each one receiving antenna, due to the side-to-side and front-to-back antenna couplings.

In order to avoid such a transmitter-to-receiver interference, a conventional repeating system uses a pair of different transmitting and receiving carrier frequencies in all hops over and over again. This system is called a two frequency plan repeating system.

In the view point of the frequency utilization efficiency, the two frequency plan repeating system is inferior to a one-frequency repeating system where the same frequency carrier pair is used for both transmitting and receiving carriers in common for all hops, although a cochannel dual polarization system has already been used for various digital microwave radio.

In order to realize the one frequency repeating system, the transmitter-to-receiver interference must be eliminated. Then, twice radio channel can be utilized within the usual radio band.

Summary of the Invention

It is an object of the present invention to provide a digital radio wave repeater of a one frequency type wherein the same carrier frequency is used for the transmitting and receiving carriers to thereby improve the frequency utilization efficiency.

It is another object of the present invention to provide a one frequency repeater where any transmitter-to-receiver interference signal can be effectively eliminated.

It is still another object of the present invention to provide a one frequency repeater where the transmitting carrier frequency can be equalized to the receiving carrier frequency thereby to improve the reliability of cancellation of the transmitter-to-receiver interference signal.

A one frequency repeater according to the present invention is used in a digital radio wave transmission system wherein a carrier is modulated by a digital baseband signal and is transmitted as a digital radio wave. The repeater comprises transmitter circuit means for producing a first digital radio wave signal modulated by a first digital baseband signal. The first digital radio wave has a first carrier frequency. The first digital radio frequency signal is radiated from the transmitting antenna means. The repeater further comprises receiving antenna means for receiving a second digital radio wave signal modulated by a second digital baseband signal. The second digital radio wave signal has a second carrier frequency substantially equal to the first carrier frequency. The receiving antenna means receives the first digital radio wave signal as a transmitter-to-receiver interference signal due to coupling between the first transmitting antenna means and the first receiving antenna means. The repeater comprises receiver circuit means for extracting the second digital baseband signal by demodulating the second radio wave signal received at the receiving antenna. The demodulated second digital baseband signal is accompanied with a baseband interference signal derived from the transmitter-to-receiver interference signal. The repeater comprises means for producing an estimated-interference signal corresponding to the baseband interference signal from the first digital baseband signal and interference cancelling means for cancelling the baseband interference signal from an output signal of the receiver circuit means by said estimated interference signal. The first subtracting means outputs the demodulated second digital baseband signal without the baseband interference signal.

The estimated-interference signal producing means is a transversal filter means comprising a delay line with a predetermined number of tap gains.

The one frequency repeater further comprises decision circuit means for deciding the second digital baseband signal from an output of the interference cancelling means and regenerating the second digital baseband signal, and tap gain control means. The tap gain control means comprises first subtracting means for subtracting the regenerated second digital baseband signal from the output signal of the interference cancelling means and providing an error signal, multiplier means for multiplying the first digital baseband signal by the error signal and providing a multiplied signal, and low-pass filter means for smoothing the multiplied signal. The smoothed multiplied signal is supplied to the transversal filter means to control the tap gains so that the error signal is zero.

In another aspect of the present invention, the transmitter circuit means includes variable frequency oscillator means for fine adjusting the first carrier frequency. The variable frequency oscillator means is controlled by an output signal of the first multiplier means so that the first carrier frequency is made equal to the second carrier frequency.

In still another aspect of the present invention, the estimated-interference signal is produced by frequency shifting the first digital baseband signal or a digital intermediate frequency signal by an amount corresponding to a frequency difference between the transmitter and the receiver carrier frequencies.

In another aspect, the transmitter circuit means has first variable intermediate frequency oscillator means for providing a first intermediate frequency signal, modulating means for modulating the first intermediate frequency signal by the first digital baseband signal, and frequency-up converter means for converting the first intermediate frequency of the modulated signal to the first carrier frequency and producing the first digital radio wave signal. The receiver circuit means has frequency-down converter means for converting a second digital radio wave signal received at the receiving antenna means to a down-converted signal having a second intermediate frequency, second variable intermediate frequency oscillator means for providing a second intermediate frequency signal, and demodulating means for demodulating the down-converted signal by the second intermediate frequency signal and providing a demodulated signal. The repeater further comprises means for detecting from the demodulated signal a phase difference between the down-converted signal and the second intermediate signal, the second variable intermediate frequency oscillator means being controlled by the phase difference, and phase comparator means for phase-comparing the second intermediate signal and the first intermediate signal and providing an error signal. The error signal being applied to the first variable intermediate oscillator means so that the first intermediate frequency signal is phase-locked to the second intermediate signal.

Further objects, features and other aspects will be understood from the following detailed description of preferred embodiments of the present invention with respect to the accompanying drawings.

Brief Description of the Drawings

<ul id="ul0001" list-style="none"><li>Fig. 1 is a schematic view for illustrating a conventional two frequency plan repeating system;</li><li>Fig. 2 is a schematic view for illustrating a one frequency repeating system;</li><li>Fig. 3 is a schematic view for illustrating various transmitter-to-receiver interferences of a repeater in the one frequency repeating system;</li><li>Fig. 4 is a view illustrating an equivalent circuit for the baseband of the repeater of Fig. 3;</li><li>Fig. 5 is a block circuit diagram of a repeater according to an embodiment of the present invention;</li><li>Fig. 6 is a block circuit diagram view of an example of an estimated-interference signal producing circuit;</li><li>Fig. 7 is a block circuit diagram view of a repeater according to another embodiment of the present invention;</li><li>Fig. 8 is a block circuit diagram view of a repeater according to still another embodiment of the present invention;</li><li>Fig. 9 is a block circuit diagram view of a part according to an embodiment of the present invention;</li><li>Fig. 10 is a block circuit diagram view of a repeater according to another embodiment of the present invention;</li><li>Fig. 11 is a block circuit diagram view of an example of a frequency shifter used in the embodiment of Fig. 10; and</li><li>Fig. 12 is a diagram view of a control circuit used for the frequency shifter of Fig. 11.</li></ul>

Detailed Description of Preferred Embodiments

Prior to description of embodiments of the present invention, a description will be made to the conventional two frequency plan repeating system as described hereinafter and a fundamental concept of the present invention.

Referring to Fig. 1, the conventional two frequency plan repeating system is depicted for a multi-hop two-way microwave radio. In the system, a carrier frequency of a digital microwave signal received at a repeater is different from that of a digital microwave signal transmitted therefrom so that the digital microwave signal transmitted from the repeater cannot be received at any receiver in the same repeater, as illustrated by a received carrier frequency f₂ and a transmitting carrier frequency f₁ at a repeater 101 but a received carrier frequency f₁ and a transmitting carrier frequency f₂ at a next repeater 100 or 102.

Therefore, the two frequency plan repeating system is advantageous to reduction of the transmitter-to-receiver interference. However, it is disadvantageous in the frequency utilization efficiency because the allowable frequency band is divided into the transmitter band and the receiver band, although a cochannel dual polarization system has already been used for various digital microwave radio system.

Referring to Fig. 2, the one frequency repeating system uses the same frequency f₁ for the transmitting carrier and the receiving carrier in each repeater 100-102. However, the frequencies of the transmitting and the carrier frequencies do not coincide with each other but have an inevitable small frequency differences in the actual use by various reasons why, for example, those frequency sources are different. Therefore, those carrier frequencies are represented by different notations f₁₁, f₁₂, f₁₃ and f₁₄ in Fig. 3.

Referring to Fig. 3, each repeater (represented at 101) in the one frequency repeating system includes a first repeating path comprising a receiving antenna 201, a regenerative repeating circuit 110, and a transmitting antenna 202. Thus, a digital microwave signal 301 of a carrier frequency f₁₁ received at the receiving antenna 201 is repeated by the first repeating path and transmitted from the transmitting antenna 202 as a digital microwave signal 302 of a carrier frequency f₁₂ which is substantially equal to the frequency f₁₁. Another second opposite directional repeating path of the repeater 101 similarly comprises a receiver antenna 203, regenerative repeating circuit 111, and a transmitting antenna 204, but the receiving antenna 203 and the transmitting antenna 204 are arranged adjacent the transmitting antenna 202 and the receiving antenna 201, respectively. The second repeating path repeats a digital microwave signal 303 of a carrier frequency f₁₃ which is substantially equal to the carrier frequency f₁₂ as a signal 304 of a carrier frequency f₁₄ which is substantially equal to the frequency f₁₃.

Since the microwave signals 302 and 304 transmitted from transmitting antennas 202 and 204 have a substantially equal carrier frequency f₁ as the desired signal 303 received at receiving antenna 203, a part of the signal 302 is received as an interference signal 401 at repeating circuit 111 due to the side-to-side coupling between transmitting antenna 202 and receiving antenna 203, while a part of the signal 304 being applied to repeating circuit 111 as another interference signal 402 due to the front-to-back coupling between transmitting antenna 204 and receiving antenna 203. When the desired signal 303 is demodulated to obtain the baseband signal at a demodulator in the repeating circuit 111, those interference signals 401 and 402 are also demodulated to generate respective baseband signals, which interfere the desired baseband signal.

Another part 403 of the signal 302 and another part 404 of the signal 304 are also interference signals with the desired signal 301 received at antenna 201, due to the front-to-back coupling between antennas 202 and 201 and the side-to-side coupling between antennas 204 and 201.

Referring to Fig. 4, an equivalent circuit of the repeater is illustrated in respect to the baseband signals under a condition of the transmitter-to-receiver interferences due to the side-to-side coupling and the front-to-back coupling.

In a condition of no interference signal, a desired baseband signal BS₁ is applied to a decision circuit 1101 through a terminal 1001 corresponding to the receiving antenna (201 in Fig. 3) and is regenerated or reproduced as a correct baseband signal BS₁. The regenerated baseband signal BS₂ is supplied to a terminal 1002 corresponding to the transmitting antenna (202 in Fig. 3). Similarly, another desired baseband signal BS₃ is regenerated as a baseband signal BS₄ at a decision circuit 1111, which is supplied to a terminal 1004 corresponding to the transmitting antenna (204 in Fig. 3). Another terminal 1003 is corresponding to the receiving antenna (203 in Fig. 3).

However, in the actual use, an interference signal 401 is added to the desired signal 303 due to the side-to-side coupling, as described above. Therefore, baseband signal BS₂ is added to the desired baseband signal BS₃ as a baseband interference signal BS₂' with a delay time corresponding to a transferring time period of the interference signal 401 from antenna 202 to antenna 203 and with a phase rotation corresponding to a small frequency difference between the transmitting carrier frequency f₁₂ and the receiving carrier frequency f₁₃. An angular velocity of the phase rotation is represented by Δω₁.

Those delay time and phase rotation are illustrated by a delay circuit 131 and a multiplier 132 in Fig. 4.

Similarly, the baseband signal BS₄ is also added to the desired baseband signal BS₃ as another baseband interference signal BS₄' due to the front-to-back coupling between antennas 203 and 204. The similar delay and phase rotation are illustrated by a delay circuit 133 and a multiplier 134.

Addition of BS₂' and BS₄' to BS₃ is illustrated by an adder 141 in Fig. 4.

It will be appreciated that those baseband signals BS₂ and BS₄ are added to another desired baseband signal BS₁ at an adder 142 as baseband interference signals BS₂'' and BS₄'' through similar equivalent circuits comprising delay circuits 135 and 137 and multipliers 136 and 138, as shown in Fig. 4.

Since the frequency differences between the transmitting carrier frequencies f₁₂ and f₁₄ and the receiving carrier frequencies f₁₁ and f₁₃ are different from one another, the angular velocities of the phase rotations of the baseband interference signal are different from one another. Therefore, respective angular velocities are represented by Δω₁ - Δω₄ in Fig. 4.

Providing that Δω₁-Δω₄ are represented by Δω₁ and that the symbol rate of the baseband signal is represented by S.R., a following formula (1) is usually given: <maths id="math0001" num=""><math display="inline"><mrow><mtext>Δω₁ « S.R.   (1)</mtext></mrow></math><img file="EP0187672B1_D0001.tif" /></maths>

From the above description in connection with Fig. 4, it will be appreciated that the transmitter-to-receiver interference in a repeater of one frequency repeating system can be cancelled by producing a quasi baseband interference signal similar to a baseband interference signal, for example, BS₂' from a baseband signal BS₂ which is an interference signal source and by subtracting the estimated-interference baseband interference signal from an output signal of a demodulator in a receiver.

This is a fundamental concept of the present invention.

Referring to Fig. 5, a repeater 101 according to an embodiment of the present invention comprises two opposite directional repeating paths. Each repeating path comprises a receiving antenna of 201 and 203, a receiver circuit of 1 and 2, a decision circuit of 3 and 4, transmitter circuit of 5 and 6, and a transmitting antenna of 202 and 204.

Receiver circuits 1 and 2 comprise demodulators for demodulating signals received at antennas 201 and 203 to obtain baseband signals which are carried by the microwave radio signals 301 and 302, respectively.

For each receiver circuit of 1 and 2, a known digital radio receiver circuit is used, which usually includes two frequency-down converters, one being a down converter from a radio frequency (RF) to an intermediate frequency (IF) and the other being one from IF to the baseband.

Decision circuits 3 and 4 are for deciding the baseband signals applied thereto and regenerating or reproducing the baseband signals, respectively. The decision circuit is also known in the prior art.

Transmitter circuits 5 and 6 are for generating digital microwave signals from the carrier and the baseband signals, and comprises a digital demodulator. The transmitter circuit is also known in the prior art, and usually includes two frequency-up converter, one being a baseband to IF converter and the other being an IF to RF converter.

According to the present invention, the repeater 101 of Fig. 5 further comprises two interference cancellers 7 and 8.

The interference canceller 7 is for removing the baseband interference signals (BS₂'' and BS₄'' in Fig. 4) due to the interference signals (403 and 404 in Fig. 3) from the demodulated output signal of receiver circuit 1, while the other interference canceller 8 is for removing the baseband interference signals (BS₂' and BS₄' in Fig. 4) from the demodulated output signal of receiver circuit 2.

The interference canceller 7 comprises two estimated-interference signal producing circuits 70 and 71 and a subtractor 72. The estimated-interference signal producing circuit 70 produces an estimated-interference signal of the baseband interference signal BS₄'' from the output baseband signal BS₄ of decision circuit 4. Another estimated-interference signal producing circuit 71 is for producing an estimated-interference signal corresponding to the baseband interference signal BS₂'' from the output baseband signal BS₂ of decision circuit 3. The subtractor 72 is for subtracting the two estimated-interference signals from the output signal of receiver circuit 1. Thus, the baseband interference signals BS₄'' and BS₂'' are removed from the output signal of receiver circuit 1, and the desired baseband signal BS₁ is applied to the decision circuit 3.

The other interference canceller 8 also comprises two estimated-interference signal producing circuits 80 and 81 for producing estimated-interference signals corresponding to the baseband interference signals BS₂' and BS₄' from the output baseband signals BS₂ and BS₄ of decision circuits 3 and 4, respectively. The estimated-interference signals are applied to a subtractor 82 and are subtracted thereat from the output signal of receiver circuit 2. Thus, the desired baseband signal BS₃ is applied to decision circuit 4 without the baseband interference signals BS₂' and BS₄'.

The estimated-interference signal producing circuit 80 is a circuit having a characteristic equal to an equivalent circuit of delay circuit 131 and multiplier 132 in Fig. 4.

Referring to Fig. 4 again, the multiplier 132 can be replaced by a complex coefficient multiplier of a type wherein the coefficient changes slowly, because Δω₁ meets the formula (1), and can be considered as a complex constant coefficient multiplier for a short time period.

Therefore, the estimated-interference signal producing circuit 80 can be composed of a filter such as a transversal filter.

Fig. 6 shows an embodiment of the estimated-interference signal producing circuit 80 composed of a transversal filter or a delay line with tap gains. The transversal filter 80 comprises four T-second delay circuits 801-804 and five tap gains 805-809. The delay time T is determined to be a value of 1/R.S. or less (R.S. is the symbolic rate as described above). The determination of the number of taps is made on a case-by-case calculation. The baseband signal BS₂ is applied to an input terminal 1101 of the transversal filter 80.

Generally speaking, the filtering characteristic of transversal filter can be changed by control of the tap gains. In the shown embodiment, the transversal filter 80 is accompanied with a control circuit 9 in order to control the tap gains 805-809 in a manner of the adaptive control technique.

The control circuit 9 comprises a subtractor 91, a multiplier 92 and a low-pass filter (LPF) 93. The subtractor 91 subtracts the output signal of decision circuit 4 from the input signal of the decision circuit and gives an error signal e between the both output and input signals. The error signal e is multiplied at multiplier 92 by the baseband signal BS₂ from the output of the decision circuit 3. The multiplied output passes through LPF 93 and is thereafter applied to tap gains 805-809 to control the gains.

The error signal e represents the baseband interference signal BS₂' caused by the coupling between antennas 202 and 203 and has a strong or close correlation with the baseband signal BS₂. The correlation between the error signal and the baseband signal is detected by the multiplier 92. The tap gains 805-809 are controlled by the detected correlation through LPF 93 so that the correlation is made zero. Thus, the estimated-interference signal similar to the baseband interference signal BS₂' can be obtained as the output signal of the transversal filter 80. The estimated-interference signal is subtracted at subtractor 82 from the demodulated output signal of receiver circuit 2. Thus, the baseband interference signal BS₂' is removed from the demodulated output signal of receiver circuit 2. Terminals 1100, 1101, 1103 and 1104 in Fig. 6 are the terminals represented by the same reference numerals in Fig. 5.

Each of the other estimated-interference signal producing circuits 70, 71 and 81 is also arranged by a transversal filter and a control circuit similar to those in Fig. 6.

In Fig. 6, delay circuits 73 and 83 are for merely adjusting the relative times between the output signal of receiver circuit 1 and the output signals of transversal filters 70 and 71 and between the output signal of receiver circuit 2 and the output signals of transversal filters 80 and 81.

The embodiment of Fig. 5 is arranged to cancel the all of the transmitter-to-receiver interferences (401-404 in Fig. 3) due to the side-to-side and front-to-back antenna couplings. However, if any one of the interference signals 401 and 402 has a low power so that it can be ignored in comparison with the desired received signal 303, the estimated-interference signal producing circuit 80 or 81 corresponding to the low-power interference signal can be omitted. The estimated-interference signal producing circuit 70 or 71 can also be omitted in the similar condition.

Generally speaking, the front-to-back antenna coupling is lower than the side-to-side antenna coupling and therefore, estimated-interference signal producing circuits 71 and 81 can be omitted, as shown in Fig. 8.

In a usual microwave radio transmission system, the cochannel dual polarization transmission is used wherein two digital baseband signals are carried on a pair of orthogonally related polarized waves by a, so-called, quadrature-amplitude modulation (QAM) technique.

The present invention can be applied to a one frequency repeater in the cochannel dual polarization transmission system.

Referring to Fig. 7, a repeater 101 shown therein is used in the cochannel dual polarization transmission system and comprises two pairs of receiving and transmitting antennas 201-202 and 203-204.

A microwave radio signal 301 of a frequency of f₁₁ comprises a vertically polarized (V-pol) signal 301a and a horizontally polarized (H-pol) signal 301b. After the radio signal 301 is received at receiving antenna 201, the received signal is separated at a polarization discriminator 500 into the V-pol signal and the H-pol signal. The V-pol signal is applied to a repeating path comprising a receiver 1a, a decision circuit 3a and a transmitter circuit 5a. A baseband signal BS_1a is extracted from the V-pol signal and is regenerated as a baseband signal BS_2a, which is carried on a carrier of a frequency of f₁₂ to reproduce a V-pol signal.

In a similar repeating path of a receiver circuit 1b, a decision circuit 3b and a transmitter circuit 5b, a baseband signal BS_1b is extracted from the H-pol signal and a fresh H-pol signal of a carrier frequency of f₁₂ is reproduced. The reproduced H-pol signal carries a regenerated baseband signal BS_2b.

Those reproduced V-pol signal and H-pol signal are combined at a cross-polarization combiner 501, and the combined signal is radiated from the transmitting antenna as a microwave radio signal 302 comprising a V-pol component 302a and an H-pol component 302b.

Another microwave radio signal 303 of a frequency of f₁₃ comprising cross-polarized waves 303a and 303b is received at antenna 23 and is processed at a polarization discriminator 502, two repeating paths, and a combiner 503 in a manner similar to the above-described process of the microwave radio signal 301. That is, the V-pol signal is processed at a repeating path comprising a receiver circuit 2a, a decision circuit 4a and a transmitter circuit 6a so that a baseband signal BS_3a carried on the V-pol signal is regenerated as a baseband signal BS_4a and is carried on a V-pol wave signal of a carrier frequency of f₁₄. On the other hand, a baseband signal BS_3b is extracted from the H-pol signal and is regenerated as a baseband signal BS_4b which is carried on an H-pol wave signal of the same frequency carrier. The V-pol signal and the H-pol signal carrying the baseband signals BS_4a and BS_4b are combined at combiner 12 and are radiated from transmitter antenna 204 as a microwave radio signal 304 comprising a V-pol and an H-pol component 304a and 304b.

In the repeater, the microwave radio signals 302 and 304 are inputted to both of receiver circuits 2a and 2b through receiver antenna 203 and discriminator 11 as interference signals (401 and 402 in Fig. 3) due to the side-to-side and front-to-back antenna couplings. Accordingly, baseband signals BS_2a, BS_2b, BS_4a and BS_4b are interference signal sources against the desired baseband signal BS_3a.

A transmitter-to-receiver interference canceller 8'a is provided between the receiver circuit 2a and decision circuit 4a. The canceller 8'a comprises four estimated-interference signal producing circuits 80a, 80b, 81a and 81b and a subtractor 82. Those estimated-interference signal producing circuits 80a-81b are ones for producing estimated-interference signals similar to baseband interference signals BS'_2a, BS'_2b, BS'_4a and BS'_4b based on baseband signals BS_2a, BS_2b, BS_4a and BS_4b, respectively. Each estimated-interference signal producing circuit of 80a-81b can be arranged by a transversal filter and a control circuit in the manner of Fig. 6.

Similar transmitter-to-receiver interference cancellers 7'a, 7'b, and 8'b are provided between receiver circuits and decision circuits 1a-3a, 1b-3b and 2b-4b, respectively.

Thus, desired baseband signals BS_1a, BS_1b, BS_3a and BS_3b are applied to decision circuits 3a, 3b, 4a and 4b, without baseband interference signals.

In this embodiment, it is also possible to omit any one of estimated-interference signal producing circuits corresponding to an interference signal having an ignorable low power.

In the above-described embodiments, it is assumed that the carrier frequencies f₁₁-f₁₄ are substantially equal to one another so as to meet the formula (1). If the formula (1) is not met, the transmitter-to-receiver signal cannot be cancelled.

Referring to Fig. 8, another embodiment shown therein is similar to the embodiment of Fig. 5 except the provision of a circuit means for adjusting the transmitting carrier frequency so as to the formula (1). Further two estimated-interference signal producing circuits 71 and 81 are omitted assuming that the front-to-back antenna coupling is ignorable.

The similar parts are represented by the same reference numbers in Fig. 5, and the description is omitted.

Transmitter circuits 5 and 6 have carrier sources of variable frequency oscillators 51 and 61, respectively. Modulators are shown at 52 and 62.

Subtractors 60 and 30, which are corresponding to subtractor 91 in Fig. 6, are connected across decision circuits 3 and 4, respectively. The output error signal e of each subtractor 60 and 30 is supplied to each of multiplier 96 and 99 which are corresponding to 92 in Fig. 6. 97 and 98 represent complex output terminals of multipliers 96 and 99. Output signals from the complex output terminals 97 and 98 are applied to variable frequency oscillators 51 and 61, respectively.

Now, providing that the delay circuits 131 and 137 in Fig. 4 are ignorable, the error signal e of each subtractor of 30 and 60 is given by the following equation (2): <maths id="math0002" num=""><math display="inline"><mrow><mtext>e = α·D·exp(j·Δω)   (2),</mtext></mrow></math><img file="EP0187672B1_D0002.tif" /></maths> where α is an amount of the transmitter-to-receiver interference, D being a baseband signal which is a source of the interference signal, and Δω is a phase difference between the transmitting and the receiving carrier frequencies f₁₁-f₁₄ or f₁₂-f₁₃.

Therefore, the complex output T of each multiplier of 96 and 99 is given by:<maths id="math0003" num=""><img file="EP0187672B1_D0003.tif" /></maths> where D* is a complex conjugate.

It is appreciated from equation (3) that output signals from terminals 97 and 98 represent phase differences between transmitting carriers f₁₂ and f₁₄ and receiving carriers f₁₃ and f₁₁, respectively.

Therefore, variable frequency oscillators 51 and 61 are adjusted by the output signals from terminals 97 and 98, so that the transmitting carrier frequencies f₁₂ and f₁₄ are locked to the receiving carrier frequencies f₁₃ and f₁₄. Accordingly, Δω is made nearly equal to zero, that is, formula (1) is fulfilled. Thus, cancellation of the transmitter-to-receiver interference can be carried out.

The subtractors 30 and 60 and multipliers 96 and 99 can be commonly used for those in control circuits (9 in Fig. 6) of the transversal filters 71 and 80, but can be additionally provided.

Fig. 9 shows another circuit arrangement for adjusting the transmitting carrier frequency to the receiving carrier frequency.

Referring to Fig. 9, similar parts are represented by the same reference numerals as in Fig. 5. The receiver circuit 2 includes a frequency-down converter comprising a multiplier 21 and a fixed frequency oscillator 10. The microwave signal received at antenna 203 is frequency converted to an intermediate frequency (IF) by the frequency-down converter. The frequency converted signal is demodulated at a demodulator 22. The demodulator 22 comprises a multiplier 221 and a variable frequency oscillator 222 generating IF signal.

The frequency-converted signal is multiplied by the IF signal at multiplier 222 and a demodulated signal of the baseband is outputted therefrom.

The demodulator 22 has a phase difference detector 223 for detecting a phase difference between the output signals of the frequency-down converter and variable frequency oscillator 222. The phase difference detector 223 detects the phase difference component on the output signal of the multiplier 221 and controls the oscillating frequency of the variable frequency oscillator 222. Thus, the oscillating frequency is locked to the frequency of the output signal from the frequency-down converter.

Transmitter circuit 5 has a modulator 53 comprising a multiplier 531 and a variable frequency oscillator 532 providing an IF signal.

The digital baseband signal from decision circuit 3 is multiplied by the IF signal to modulate the IF signal. The modulated signal is applied to a frequency-up converter comprising a multiplier 54 and fixed oscillator converter 10 and is frequency converted to a transmitting carrier frequency. The resulted digital microwave signal is radiated from antenna 202.

In the embodiment, the frequency-up converter in the transmitter circuit 5 and the frequency-down converter in the receiver circuit 2 commonly use the fixed frequency oscillator 10. Therefore, if the oscillating frequencies of IF oscillators 532 and 212 are coincident with each other, the transmitting carrier frequency f₁₂ coincide with the receiving carrier frequency f₁₃.

The variable frequency oscillator 532 is controlled by a phase-lock circuit 55 which comprises a multiplier 551 for detecting a phase difference between oscillating frequencies of IF oscillators 532 and 222. The output signal is applied to oscillator 532 through a low-pass filter 552 and controls the oscillating frequency of the variable frequency oscillator 532. Accordingly, the IF signal in the transmitter circuit 5 is locked to the IF signal in the receiver circuit 2. Thus, the transmitting carrier frequency is coincident with the receiving carrier frequency. This means that formula (1) is maintained, and the transmitter-to-receiver interference can be readily cancelled in the embodiments described above.

The circuit of Fig. 9 can be employed in embodiments as described hereinafter. Further, the circuit of Fig. 9 is directed to a pair of transmitter circuit 5 and receiver circuit 2, but the similar circuit can be used in various pair of transmitter circuit and receiver circuit in the repeater.

Referring to Fig. 10, another embodiment is directed to cancel the transmitter-to-receiver interference signals at the IF band. Similar parts are represented by the same reference numerals as in Fig. 5.

The transmitters 5 and 6 comprise modulators 56 and 61 for modulating the IF signal by the digital intermediate signals from decision circuits 3 and 4, and frequency-up converters 57 and 62 for converting the modulated signal to the carrier frequencies, respectively.

The receivers 1 and 2 comprise frequency-down converters 11 and 23 for down-converting the radio signals received at respective antennas to the IF frequency, and demodulators 12 and 24 for demodulating the frequency-down converted signals to extract the digital baseband signals, respectively.

In this embodiment, the IF band signal in the transmitter is used for producing a signal for removing the transmitter-to-receiver interference signal.

With respect to cancellation of the transmitter-to-receiver signal through transmitting antenna 201 and receiving antenna 203, the IF band signal is taken out at an intermediate point from modulator 56 and frequency-up converter 57. The IF band signal is applied to a frequency shifter 84. The frequency shifter 84 is for frequency shifting the IF signal by a frequency difference between the intermediate frequencies in the transmitter 5 and the receiver 2. The frequency difference is corresponding to the frequency difference of the transmitting and the receiving carrier frequencies.

Referring to Fig. 11, an example of the frequency shifter 84 is illustrated. The frequency shifter 84 comprises an oscillator 841 for providing a signal having an angular frequency ω_V and a phase shifter 842 for phase shifting the frequency signal from oscillator 841 by V/2. The output signals from oscillator 841 and phase shifter 842 are multiplied by the IF band signal from modulator 56 at multipliers 843 and 844, respectively. The multiplied signals are filtered at low-pass filters 845 and 846, respectively. These oscillator 841, phase shifter 842, multipliers 843 and 844, and low-pass filters 845 and 846 makes a coherent detector 84a.

Providing that the IF band signal p(t) is given by:<maths id="math0004" num=""><img file="EP0187672B1_D0004.tif" /></maths> where ω_IF is an angular frequency of the IF signal. The output signals from low-pass filters 845 and 846 are represented by the following single equation: <maths id="math0005" num=""><math display="inline"><mrow><msup><mrow><mtext>γ(t) = B(t)e</mtext></mrow><mrow><mtext>j(ω</mtext></mrow></msup><msup><mrow><mtext>IF - </mtext></mrow><mrow><mtext>ω</mtext></mrow></msup><msup><mrow><mtext>V</mtext></mrow><mrow><mtext>)t</mtext></mrow></msup><mtext>.   (5)</mtext></mrow></math><img file="EP0187672B1_D0005.tif" /></maths>

The output signals from low-pass filters 845 and 846 are applied to a modulator 84b. The modulator 84b comprises an oscillator 847 for oscillating a signal having an angular frequency ω_IF, a phase shifter 848 for shifting a phase of the output signal of oscillator 847 by π/2, multipliers 849 and 850 for multiplying the output signals from low-pass filters 845 and 846 by the output signals from oscillator 847 and phase shifter 850, respectively. The output signals of multipliers 849 and 850 are added to each other at an adder 851. The output (q(t)) of adder 851 is represented by:<maths id="math0006" num=""><img file="EP0187672B1_D0006.tif" /></maths> Therefore, if ω_V is selected to be the intermediate angular frequency in receiver 2, the IF band signal p(t) is frequency shifted at the frequency shifter 84 by the frequency difference between the IF signals of the transmitter 5 and receiver 2.

Referring to Fig. 10 again, the output of the frequency shifter 84 is multiplied by a coefficient α at a multiplier 85. The coefficient α represents an amount of the transmitter-to-receiver interference and is preset in a coefficient generator 86. The output multiplied signal αq(t) from multiplier 85 is applied to a subtractor 82 which is connected to an output side of frequency-down converter 23 in the receiver 2. Subtractor 82 subtracts the multiplied signal αq(t) from the output IF band signal of frequency-down converter 23.

If the transferring delay of the transmitter-to-receiver interference signal from antenna 201 to antenna 203 is ignored, the interference signal of the IF band in the receiver is expressed by αp(t)exp{j(ω_IF - ω_V)}. Therefore, the IF band signal r(t) in the receiver 2 is given by: <maths id="math0007" num=""><math display="inline"><mrow><msub><mrow><mtext>r(t) = αp(t)exp{j(ω</mtext></mrow><mrow><mtext>IF</mtext></mrow></msub><msub><mrow><mtext> - ω</mtext></mrow><mrow><mtext>V</mtext></mrow></msub><mtext>)} + R(t)   (7)</mtext></mrow></math><img file="EP0187672B1_D0007.tif" /></maths> where R(t) is a desired IF band signal at a time when the transmitter-to-receiver interference is not caused.

The output A(t) of subtractor 82 is: <maths id="math0008" num=""><math display="inline"><mrow><mtext>A(t) = r(t) - α·q(t).</mtext></mrow></math><img file="EP0187672B1_D0008.tif" /></maths> From equations (6) and (7),<maths id="math0009" num=""><img file="EP0187672B1_D0009.tif" /></maths> Therefore, the transmitter-to-receiver interference signal is removed in the IF band.

The interference signal from transmitter 6 to receiver 1 is, in the similar manner, removed by circuits of a frequency shifter 74 like 84, a coefficient generator 75, a multiplier 76, and a subtractor 72.

Similarly, transmitter-to-receiver interference signals from antennas 201 and 204 to antennas 201 and 203 are also cancelled by circuit arrangements of frequency shifters 84 and 74, multipliers 78 and 88, coefficient generators, delay circuits 79 and 89, and subtractors 72 and 83, respectively.

Delay circuits 79 and 89 are corresponding to the equivalent delay circuits 135 and 133 in Fig. 4. If the delays corresponding to the equivalent delay circuits 131 and 137 in Fig. 4 are not ignored, delay circuits should be connected between multiplier 85 and subtractor 82 and between multiplier 76 and subtractor 72, respectively.

In the shown embodiment, each of frequency shifters 84 and 74 is commonly used for cancellation of the transmitter-to-receiver interference signals due to the front-to-back coupling and the side-to-side coupling. However, separate frequency shifters can be used for cancellation of the interference signals due to different couplings.

If any one of transmitter-to-receiver interference signals can be ignored, the corresponding circuits for removing the one can be omitted.

In the embodiment shown in Fig. 10, the transmitter-to-receiver interference signal is cancelled in the RF band. However, if the input signal of each frequency shifter of 74 and 84 is taken out from the baseband signal as illustrated by dotted lines in Fig. 10, the transmitter-to-receiver interference signal can be cancelled in the baseband. In the case, subtractors 72 and 83 must be moved to output sides of modulators 12 and 24, and the oscillating frequency difference of the oscillators 841 and 847 in Fig. 11 should be selected to be a frequency difference in the baseband corresponding to that of the microwave carrier frequencies.

In the embodiment of Figs. 10 and 11, it is assumed that the frequency difference ( <maths id="math0010" num=""><math display="inline"><mrow><msub><mrow><mtext>Δω = ω</mtext></mrow><mrow><mtext>IF</mtext></mrow></msub><msub><mrow><mtext> - ω</mtext></mrow><mrow><mtext>V</mtext></mrow></msub></mrow></math><img file="EP0187672B1_D0010.tif" /></maths> ) is constant. However, ω_IF and ω_V vary in the microwave transmission system, so that the Δω is not maintained constant.

Fig. 12 shows a circuit for controlling the frequency difference between the oscillators 841 and 842 in response to variation of an actual frequency difference between IF signals in the transmitter and the receiver.

Referring to Fig. 12, an error e of the output signal and the input signal of decision circuit is detected at a subtractor 40 similar to subtractor 60 in Fig. 8 or 91 in Fig. 6.

According to equation (2), <maths id="math0011" num=""><math display="inline"><mrow><mtext>e = α·D·exp(jΔω₁)</mtext></mrow></math><img file="EP0187672B1_D0011.tif" /></maths> when no interference cancelling signal α·q(t) is used. In case the interference cancelling signal α·q(t) is used, e = 0 as will be noted from equation (8) when the phase difference (Δω₂) of the oscillators 841 and 847 is equal to the phase difference (ω_IF - ω_V) between the IF band signals in the receiver and the transmitter. If Δω₂ <img file="EP0187672B1_D0012.tif" /> ω_IF - ω_V, the interference signal (α·exp(jΔω)) is not cancelled, the error e is given by: <maths id="math0012" num=""><math display="inline"><mrow><mtext>e = D{α·exp(jΔω) - α'·exp(jΔω₂t)}</mtext></mrow></math><img file="EP0187672B1_D0013.tif" /></maths> where (α'·D·exp(jΔω₂t)) is a component of the interference cancelling signal.

The error signal is applied to a multiplier 852 and is multiplied by a complex conjugate {α'·q(t)}* of the multiplier 85. The complex component of the result is expressed by <maths id="math0013" num=""><math display="inline"><mrow><msub><mrow><mtext>ϑ</mtext></mrow><mrow><mtext>e</mtext></mrow></msub><mtext>(t) = |p(t)|²[α·α'·exp{jΔωt - Δω₁t}]</mtext></mrow></math><img file="EP0187672B1_D0014.tif" /></maths> . Now, providing <maths id="math0014" num=""><math display="inline"><mrow><msub><mrow><mtext>α = a·exp(jϑ</mtext></mrow><mrow><mtext>I</mtext></mrow></msub><mtext>)</mtext></mrow></math><img file="EP0187672B1_D0015.tif" /></maths> and <maths id="math0015" num=""><math display="inline"><mrow><msub><mrow><mtext>α' = a'·exp(jϑ</mtext></mrow><mrow><mtext>C</mtext></mrow></msub><mtext>)</mtext></mrow></math><img file="EP0187672B1_D0016.tif" /></maths> , <maths id="math0016" num=""><math display="inline"><mrow><msub><mrow><mtext>ϑ</mtext></mrow><mrow><mtext>e</mtext></mrow></msub><msub><mrow><mtext>(t) = |p(t)|²·a·a'·sin[ϑ</mtext></mrow><mrow><mtext>I</mtext></mrow></msub><msub><mrow><mtext> + Δωt - (ϑ</mtext></mrow><mrow><mtext>C</mtext></mrow></msub><msub><mrow><mtext> + Δω</mtext></mrow><mrow><mtext>e</mtext></mrow></msub><mtext>t)]</mtext></mrow></math><img file="EP0187672B1_D0017.tif" /></maths> . Namely, the multiplier 852 outputs the signal ϑ_e(t), and the ϑ_e(t) represents an actual difference between the phase difference of the oscillators 841 and 842 and the phase difference between the IF band signals in the transmitter and the receiver.

The multiplier output ϑ_e(t) is used for control of the oscillating frequency of one of the oscillators 841 and 842. Thus, the interference cancelling signal α·q(t) is equalized to the interference signal.

The control circuit can be applied to control of the frequency shifter 74.