Nonfrequency-converting microwave radio repeater using a low powerumption amplifier

L-~60 l 106~5~7 1 ¦ A NONFREQUENCY-CONVERTING MICROWAVE RADIO REPEAT~R 2 ¦ USING A LOW POWER CONSUMPTION AMPLIFIER 3 l ¦ BACKGROUND OF THE INVENTION 5 ¦ This invention relates to microwave radio repeaters for 6 ¦ telecommunications, and more specifically to an inexpensive nonfrequency7 ¦ converting radio repeater system having very low power requirements.

8 l 9 ¦ Microwave radio repeater systems are used to amplify and redirect the RF carriers where the distance between radio terminals is 11 extreme and where geographical and manmade obstacles block the line-of12 sight transmission path. Generally, microwave repeaters may be classified 13 as either active or passive. The more common type of passive repeater 14 is the flat "billboard" type metal reflector which acts as a simple microwave mirror. The passive repeater is very useful, where the 16 topography permits, and where it is suitable in other respects for the 17 particular system application. The principal reasons for using a passive 18 repeater are that it provides a fixed signal gain, it is very reliable, 19 and it requires virtually no maintenance. Thus, it can be used on short microwave hops which are inaccessible or too expensive for the typical 21 active repeater. However, for a variety of reasons (cost being the 22 biggest factor), certain applications of passive repeaters are even more 2 impractical than active repeaters both to install and operate. In 2 frequency-congested areas, billboard reflectors can cause interference 2 with adjacent systems since a billboard reflects all electromagnetic 2 energy independent of the source. Furthermore, passive repeaters cannot 27 be used effectively, for reasons of low RF gain, in the 2 GHz band 31 below or wherc the ircluded ~ exceeds 140.

32 j - i : . . . ~ .: ; . . .

1069S~7 L-460 1 Currently, the active repeaters in most radio relay systems 2 are either of the demodulating-remodulating "baseband" type or the IF or RF heterodyne type. In each case, the individual information-carrying channels are amplified and otherwise processed at frequencies usually S lower than the microwave carrier frequency. For such active repeaters, 6 the overall costs can be prohibitive for inaccessible repeater locations.

7 Aside from the initial radio equipment cost, access roads for routine 8 maintenance must be built and maintained, utility power lines or local 9 power generators must be installed, and repeater housings also constructed. The invention disclosed herein offers an economical and 11 reliable alternative to the traditional active microwave repeater for 12 applications in remote locations where passive repeaters might otherwise 13 be used.

14 An inexpensive microwave radio repeater providing 25 to 50 dB 1~ of gain (not including antenna gain) and capable of being powered from 17 solar cells and/or primary cells could offer a very practical alternative 18 to the passive repeater in the application described above. Since solar 19 cells are low-voltage devices and are capable of supplying only very little power, a radio repeater powered from such cells must be extremely 21 uncomplicated and have very low power requirements. A nonfrequency22 converting radio repeater using a single stage of amplification can 2 fulfill all these requirements.

24 Nonfrequency-converting duplex (bidirectional) repeaters, 26 similar in configuration to the type used herein, have been used 27 commercially for years in the communication industry on submarine 28 cable systems and on CATV cable systems. See "System Considerations 29 the Design of a Two-l~ay Transmission Syste~", by H. B.

Marron and '2 1069587 L-460 l A. W.

Barnhart in the l9th Annual NCTA Convention Transcript 1970, pp.

2 547 - s71.

For reasons of poor amplifier response and high echo 3 distortion, such repeaters have been limited to cable TV systems and 4 low frequency communication systems. A recent discussion of a single channel, nonfrequency-converting duplex repeater in the VHF/UHF 6 band (225 MHz to 468 MHz) may be found in "Short-Hop Radio-Relay Systems7 Using Tunnel-Diode Repeaters" by D. L.

Hedderly et al, Proceedings IEE, 8 vol. 114, April 1967, pp. 435 - 442.

This experimental system for use 9 on electrical transmission high-voltage towers used tunnel diode amplifiers and yagi aerials with curtain reflectors, both of which ll are very impractical for high-quality reliable microwave radio 12 telecommunication operation of the type discussed herein.

13 1~ A multichannel nonfrequency-converting repeater has never been used on a commercial basis for microwave radio communications in 16 the common carrier bands above 1.7 GHz.

The reason for this is that the 17 many technical problems involved have been thought to be insurmountable.18 A high-level echo having a long time delay resulting from transmitting 19 and receiving on the same carrier frequency can cause severe echo distortion in the radio baseband.

And, this is true independent of the 21 form of carrier modulation used.

The invention herein minimize~s the 22 echo distortion to acceptable levels, making the nonfrequency-converting23 repeater a practical concept.

Also, by using a reliable repeater design 24 having absolute minimum power requirements, the entire repeater can be used in remote locations without the necessity of locally generated 26 utility-supplied AC power.

27 28 2g 1069587 1 SUMMARY OF INVENTIO~ 2 A non-frequency converting microwave radio re~eater, having 3 first and second input terminals, employes a single microwave amplifier 4 to provide duplex operation for angle-modulated carrier signals Fl and ~2' where Fl ~ F2, and their, respective non-overlapping modulation bands 6 ~Fl and QF2 ~ which are received, respectively, at the first and second 7 input terminals. The two modulation bands, which are travelling in 8 opposite directions, are coupled from said input terminals into bandpass 9 filters which channel the recei~ed signals to the input of the single microwave amplifier. Once amplified, the signals are separated with two more bandpass filters. Coupling devices are used to combine and 12 separate the signals for the two directions of transmission and to inter 13 connect the first and second input terminals with radio antennas.

14 By using two microwave amplifiers,additional coupling de vices and bandpass filters, the filters having appropriate passbands, 16 frequency diversity operation may be obtained. For this case, two addi 17 tional angle-modulated carrier signals F3 and F4 are employed and the, 18 respective, modulation bands are ~F3 and ~F4. The modulation bands ~F 9 and ~F2 pass through one microwave amplifier and the modulation bands ~F3 and ~F4 pass through the other microwave amplifier. Additional coup 21 ling devices are also required to effect the desired interconnection be 22 tween the input and output antennas and between the proper filters and 23 their associated amplifier.

2g 5 BRIEF DESCRIPTION OF THE DRAWINGS _ .

26 The principles of this invention will be best understood by 27 reference to the detailed description of the invention in conjunction 28 with the drawings.

~9 ~0 31 - 5 32 L 4~0 ~C~695~37 1 FIG. 1 illustrates a two-hop microwave telecommunication system 2 employing a single repeater of the type discussed herein.

3 FIG. 2 is a block diagram illustrating one embodiment of this 4 invention.

~ FIG. 3 is a block diagram illustrating a frequency diversity 6 repeater in accordance with this invention.

DESCRIPTION OF THE PREFERRED EMBODIMENT 9 The microwave radio repeater described herein would typically 0 be used on a two-hop microwave system or on most multihop systems.

11 The only restriction is that the path length to and from the particular 12 repeater discussed herein should be limited so as to avoid excessive 13 path loss. A basic two-hop application of this repeater is as shown 14 in FIG. 1.

The nonfrequency-converting repeater 11 transmits and receives to and from the west terminal 10 over the same two frequencies 16 that are used for transmission and reception to and from the east terminal 17 12. For example, the Fl west-to-east RF carrier would be 2.l0 GHz and 18 the F2 east-to-west RF carrier would be 2.15 GHz. The corresponding 9 modulation bands are respectively aFl and ~F2, however, for simplicity, only the carrier frequencies Fl and F2 will be used in the following dis21 cussion. It will be understood that, where appropriate, the modulation 22 bands ~Fl and ~F2 are also to be included. Such a system configuration 23 is designed for use with the "high-low" or CCIR type frequency plans which 24 divide the frequency bands into equal halves.

The preferred configuration of the radio repeater for narrow26 band operation is shown diagrammatically in FIG. 2. The basic concept 27 of this configuration is to combine the west-to-east carrier signal, Fl, 28 and the east-to-west carrier signal, F2, at the in~ut to microwave 31 ~ - 6 L-460 10~9587 1 amplifier 26. The amplifier then simultaneously amplifies the two 2 RE' channels being transmitted in both directions through the repeater.

3 The signals are then split at the output of the amplifier, filtered, 4 and directed to the proper transmitting antenna.

The Fl carrier is received at the west microwave parabolic 6 dish antenna 20 and fed to circulator 22 using either coaxial cable or 7 an appropriate waveguide run. The rotation of circulator 22 is such that 8 Fl is passed to bandpass filter 23, centered at the Fl frequency, and 9 then on to circulator 25. At this circulator, the Fl carrier is directed 0 to the microwave amplifier 26 wherein the signal is amplified. the ¦ t~ ical ga~ required for most applications is in the 40 - S0 d~ range, ~si 22246i 228 230 31 - 6A 32 ~i L-460 1 1069~i87 l ¦ as will be discussed more fully later. At the output of amplifier 26, 2 ¦ Fl is directed through circulator 27 with the orientation as shown, and 3 ¦ on to a second bandpass filter 29 having a passband centered at the Fl ¦ frequency. Bandpass filter 29 rejects any unwanted signals outside the 5 ¦ Pl band, which would include the F2 carrier, and also undesired 6 ¦ intermodulation products. After passing through filter 29, the Fl carrier 7 ¦ is directed by circulator 30 to the east microwave antenna 21. At the 8 ¦ same time, the F2 carrier is received by microwave antenna 21 and fed 9 ¦ through a feeder system consisting of either a coaxial run or a lO ¦ waveguide system to circulator 30. Due to the rotation of this circulator, 11 ¦ F2 is directed to bandpass filter 28 whose passband is centeret at the 12 ¦ F2 frequency. After filtering, the F2 carrier is passed through 13 ¦ circulator 25 and on to bandpass filter 23. Sinca the bandpass filter 14 23 offers a nonterminating impedance to the F2 band of frequencies, the F2 carrier is reflected and directed back to circulator 25 and on 16 to microwave amplifier 26. Since the amplifier is not into saturation, 17 the F2 carrier is amplified along with carrier Fl by the same amount 18 in microwave amplifier 26 and directed to circulator 27. Due to the 19 orientation of circulator 27, the F2 carrier is reflected off of filter 29 back through circulator 27 and to the bandpass filter whose ~l passband is centered at the F2 frequency and is transmitted by 22 antenna 20.

23 2~ The microwave amplifier used in the specific embodiment tested was a low-noise linear microwave transistor using discrete 26 microstripline circuit design. The important characteristics of the 27 amplifier are high gain over the frequencies of interest, high output 28 power, low noise figure, good frequency response, high DC efficiency, 29 minimum delay distortion, minimum AM/PM conversion, and reliable 31 operation. It must also be capable of operating over a wide range of 3~ ~ - 7 L-460 ~o69587 l environmental conditions since the typical repeater site will be 2 unattended and subject to wide temperature variation. The particular ,~ amplifier used in the specific embodiment tested had a maximum delay of 2 ns variation over 20 MHz, an available gain of + 50 dB, an output power capability (1 dB gain-compression point) of + 15 dBm in the 2 GHz and 4 GHz band and an AM/PM conversion of 1/dB. Of secondary 7 importance is nonlinear amplitude distortion in the amplifier since 8 the RF carriers are frequency or phase modulated. Theory and measurement 9 indicate that a very low amount of intermodulation distortion will be produced with only two RF angle-modulated signals in an amplifier ll operating near or into saturation and having a low value of AM/PM 12 conversion. This is important since an upfade condition, which would 13 cause the amplifier to go into saturation, must not cause noticeable l~ distortion. The only result of an upfade condition should be a reduction in available gain in both directions. The preferable 16 operating point of the amplifier should be at the 1 dB gain compression 17 point, or at the edge of saturation, since it is at this point that the 18 amplifier is operating at maximum efficiency. The actual amplifier 19 used in the specific embodiment tested was manufactured by Varian Associates, Palo Alto, California, commercially available as part 21 number VSL-7441. Since this amplifier had a small signal gain of 22 25 dB, two such amplifiers were used in series to obtain the requisite 23 amplification. Certainly, there are other amplifiers commercially 2~ available having greater RF gain and output power which would operate equally well as the amplifier tested in the specific embodiment.

26 27 A possible variation of the single amplifier configuration 2 would be to have two such amplifiers operate in parallel, with both 2 input and output coupled together with a suitable impedance-matching 3 - 8 L 460 1 10tj9S87 1 ¦ device such as an RF hybrid. The advantage of this configuration would 2 ¦ be that the maximum output level capability would be increased and a 3 ¦ certain degree of system protection would be provided. However, the 4 ¦ doubling of DC power consumption would probably outweigh the benefit 5 ¦ derived from having two such amplifiers operate in parallel. This may 6 ¦ not be true, however, in high channel capacity systems where reliability ¦ is critisal. The increased current drain may not be as important in 8 ¦ every case as reliability.

g I Equally as important as the amplifier design is the proper choice 11 of antenna. The standard parabolic antenna consists of a parabolic dish 12 illuminated by a feed horn at the focus. These antennas are commercially 13 available in most frequency bands with diameters of 2, 4, 6, 8, 10, and 14 sometimes 12 feet.

The standard parabolic dish antenna may be used in this system, providing certain precautions are taken to avoid side-to-side 16 coupling from one antenna to the other at the same repeater location.

17 Recently, metallic cylindrical build-out shields (called shrouds) have 18 been attached to these standard parabolic dish antennas to reduce the 19 far-field spurious radiation (side lobes).

It has also been found that these high-performance shrouded parabolic antennas can also significantly 21 improve the side-to-side coupling.

22 23 Shrouded antennas are normally fabricated using two-piece 24 construction with the cylindrical shield being added to the dish antenna with rivets or similar devices.

It has now been found that the preferable 26 construction technique is to have a single unit construction so that the 2~ leakage between the shroud and the dish reflector is eliminated.

As one 28 might expect, this improvement to the parabolic antenna not only reduces 29 side-to-side coupling but also aids in directivity. This permits fuller 31 dish illumination resulting in grea;er gain efficiency. Using this 32 L-460 1 1069S87 ]. ¦ preferred type of construc~ion on two 6 foot parabolic dish antennas, ¦ and using cross-polarization between the transmit and receive signals, 31 the side-to-side isolation was measured, in the specific embodiment ¦ tested, to be better than 65 dB at 2 GHz, with a gain efficiency of 5 ¦ about 60%. At 4 G~z, better than 85 dB isolation between similar 61 antennas was obtained and better than 90 dB at 6 GHz.

~1 8 ¦ The horn reflector antenna probably has sufficient performance 9 ¦ characteristics to be used for certain duplex repeater applications.

10 ¦ The horn antenna has high gain and provides sufficient side-to-side 11 ¦ ratios to allow operation in two directions at the same frequency. The 12 ¦ disadvantages of this antenna for this particular application are 13 ¦ numerous. It is not commercially available in all frequency ranges 14 ¦ (e.g., 2 GHz antennas are not made), and it is very large, heavy, and 15 ¦ difficult to mount on a microwave tower. ~ot only is the antenna itself 16 ¦ expensive, but the mounting and tower costs are also higher than the 17 ¦ recommended parabolic antenna discussed earlier.

18 l 19 ¦ Since the transmitting and receiving signals are at the same 20 ¦ frequency, very low antenna side-to-side coupling is a necessity for low 21 ¦ dis~ortion operation. The L3 path shown in FIG. 2 illustrates the 22 ¦ coupling path from the transmitting antenna to the receiving antenna.

23 ¦ Any crosscoupling between these two antennas will introduce an echolike 24 ¦ distortion having a long time delay. As has been predicted in the prior 25 ¦ art, the antenna-to-antenna coupling produces the largest single source 2~ ¦ of intermodulation distortion in the repeater, and it is particularly 27 ¦ damaging to high-quality performance. See "Echo Distortion in Frequency_ 28 ¦ Modulation" by R. G.

Medhurst, in Electronic and Radio Engineer, 29 ¦ July 1959, pp. 253 - 259. In general, the distortion magnitude is a 301 complex function of the relative magnitude of the echo signal, the echo 31 l 32 I - 10 L-460 11~6S S87 l ¦ delay with respect to the main signal, the peak deviation, 2 ¦ the width of the baseband, and the top modulating frequency. Specifically, 3 ¦ Medhurst has shown for frequency division multiplex (FDM) systems that ¦ the maximum echo distortion in the top baseband channel is proportional 5 ¦ to the fourth power of the echo delay and to the square of the echo 6 ¦ amplitude. The precautions suggested above should be exercised to the 7 ¦ fullest extent possible to reduce the echo distortion for high-quality 8 ¦ telecommunication operation, whether the baseband infom ation is FDM 9 ¦ channels, TV, or some other information.

10 l 11 ¦ Another possible source of echo distortion is the coupling , 12 between the amplifier input and output via the filters and circulators.

13 This is shown in FIG. 2 by the two loop paths Ll and L2.

The level of14 this echo can be made very small by proper filter design and by the optional use of the circulators as shown in FIG. 2.

16 17 The bandpass filters shown in FIG. 2 as 23, 24, 28, and 29 18 provide improved performance for duplex service in this application 19 over the use of lowpass/highpass filters which have been used exclusively in CATV cable systems.

The basic function of the bandpass filters, 21 as explained earlier, is to properly channel the carrier signals from the 22 receive antenna, through the amplifier, and on to the transmit antenna.

23 The receive filters 23 and 28 also limit the amount of outside interference 24 entering amplifier 26.

Similarly, the transmit filters 24 and 29 attenuate the level of unwanted intermodulation products and higher-order 26 harmonics, generated by the amplifier, that would otherwise be put out27 with the desired signals.

The echo distortion introduced by feedback 28 paths Ll and L2 can be reduced to acceptable levels by proper design 29 of the bandpass filters.

In Ll, the amount of attenuation to the F2 band of frequencies is equal to the isolation of circulator 22 plus the 31 ~2 ~ L-460 1 1~69S87 l ¦ out-of-band rejection of bandpass filter 23 less the gain of amplifier 26.

2 ¦ The isolation of circulator 22 between ports 3 and 2 can be no better 3¦ than the antenna match on port 1. In the 2 GHz frequency band, an 4¦ antenna return loss of 20 to 25 dB is typical and therefore the isolation 5 ¦ from port 3 to port 2 can be no better than this amount. With a maximum 6 ¦ amplifier gain of 50 dB and assuming a circulator isolation of 20 dB, 7 ¦ the out-of-band rejection of bandpass filter 23 should exceed 70 dB if 8 ¦ acceptable toll quality noise requirements are to be maintained. A 9 ¦ minimum rejection of 80 dB would imply a worst case echo in the F2 band lO ¦ of 40 dB below the desired F2 signal. In a radio system tested having ll ¦ a capacity of 36 FDM channels, the 60 dB echo produced negligible amounts 12 ¦ of intermodulation (IM) noise. Of course, the IM noise depends not only on 13 the echo amplitude but also the amount of echo delay - the longer the 14 echo delay, the greater the IM noise.

The amount of attenuation to the Fl band of frequencies in loop 16 Ll is greater than the F2 isolation by the amount of isolation loss 17 given by circulator 27. Specifically, the rejection in the Fl band is 18 the isolation of circulator 27, circulator 22, and the out-of-band l9 rejection of bandpass 24, less the gain of amplifier 26. Using the same rejection figures as before, the amount of echo in the Fl band 21 caused by loop Ll would be over 60 dB below the desired signals. A 22 similar analysis for loop L2 produces a corresponding level of echo.

23 Aside from echo considerations, the gain around the various loops (Ll, 24 L2 or L3) must be less than 1 to prevent possible oscillation or instability.

If circulators are used as shown in FIG. 2, then care should be 26 taken to properly match each port with a proper terminating impedance 27 for optimum operation. For example, circulator 27 at port 2 must see 28 a good termination looking into filter 29, otherwise a return signal will 29 be conducted across circulator 27 and around loop Ll. For certain repeater applications, however, the use of circulators may not be 31 3 - 12 1 10~958~ ~-46~ 1 necessary. If the bandpass filters are designed with adequate out-of-band 2 rejection, then the added isolation provided by the circulators would not 3 be required, and their use would be an unnecessary added expense.

There are two other related factors to consider in the design of the bandpass filters for this application: group delay (also called 6 delay distortion) and filter return loss. As is well known in the art, 7 delay distortion may be caused by the microwave path, the waveguide system, ~ or the radio equipment itself. The end product is intermodulation distortion 9 in the baseband. In the propagation path between antennas, delay distortion is caused by the reflected energy or multipath interference which reaches 11 the receiving antenna but is delayed as compared to the direct signal.

12 This is a path consideration, and there is nothing that can be done 13 in the repeater equipment which will effect this type of delay dist6rtion.

14 Waveguide echoes are another source of delay distortion, and they result from impedance mismatches, i.e., poor return loss. To minimize this 16 parameter, excessive waveguide lengths should be avoided and care should 17 be taken in properly terminating the equipment. In FIG. 2, since the 18 circulators 22 and 30 will reflect the impedance offered by *he bandpass 19 filters, the design of the filters must provide adequate return loss to minimize the delay distortion problem. For many applications, a return 21 loss of 26 dB is minimum. Typically 30 to 40 dB is adequate.

22 23 Another source of delay distortion is in the bandpass filters 24 themselves. If the filters have a nonlinear phase response, the transmission time through the filter will vary with frequency~ This 26 introduces a nonlinearity in the carrier signal which results in distortion.

27 With higher channel capacity systems, this is even more of a problem 28 since higher capacity systems cannot tolerate as much group delay as 29 narrow-band systems. Whereas a 36 channel system could probably tolerate as much as lO nano-seconds ~ns) across the RF band, a 600 31 313 - 13 L-460 1 1069S~ 1 ¦ channel system cannot tolerate much more than 2 ns across the 20 MHz band.

2 ¦ Therefore, in higher channel capacity systems, a delay equalizer used 3 ¦ in conjunction with each path may be a desirable modification to the 4 ¦ configuration shown in FIG. 2. Such equalizers are well known in the 5 ¦ art and should be placed in the Fl path between filter 23 and circulator 6 ¦ 25 and in the F2 path between filter 28 and circulator 2S. An equalizer .~ ¦ here minimizes the distortion associated with phase distortion and with 8 ¦ possible AM/PM conversion in the amplifier 26. Alternatively, the delay 9 ¦ equalizers could be placed in the receive terminal equipment, but due 10 ¦ to AMtPM conversion it is probably preferable to equalize this distortion ll ¦ at the repeater equipment where it occurs.

12 l l~ ¦ There are varied system considerations in applying the duplex 14 ¦ repeater configuration illustrated in FIG. 2 to an actual microwave 15 ¦ system. For example, in a 2 GHz system as illustrated in FIG. l with 16 ¦ the following performance specifications, a fade margin of appro~imately 17 ¦ 35 dB can be expected for a total terminal-to-terminal path length of 18 ¦ up to 30 miles. ~Terminal output power +30 dBm, receiver threshold l9 -90 dBm, 6 foot antenna gain l30.4 dB, bandpass filter insertion loss l dB, and an available amplifier gain of 50 dB). Any greater distance 21 and the fade margin will drop proportionately with the increase in path 22 loss. One surprising result of this FIG. l configuration is that the 23 repeater may be placed anywhere in the path with virtually no change 24 of the system fade margin if the amplifier is always operated at its peak efficiency point, i.e., at the l dB gain compression point. This 26 is certainly contrary to a microwave system using either a passive or 27 typical active repeater. In the design of a microwave system using 28 a passive repeater over the same terminal-to-terminal 30 mile distance, 29 the optimum repeater location is not at the path midpoint. For a ~0 passive repeater, it is highly desirable to keep one of the paths very 31 32 - 14 L-460 10~;~587 1 short since its effectiveness is an inverse function of the product of 2 the length of the two paths, rather than the sum of their length. For 3 the duplex repeater described herein, the optimum location is at the midpoint of the path since statistically the two equal path lengths will fade less than a very short length path and one very long length path.

6 This unexpected result simplifies most path layout problems.

8 Occasionally a two-hop system cannot be configured for midpath 9 operation, and one path must necessarily be longer than the other (e.g., one mile and 29 miles).

An improvement can be made to the 11 arrangement shown in FIG. 2 to improve system performance over the 12 long-to-short path direction.

Since the short-to-long signal will drive 13 the repeater amplifier into saturation at some gain less than the full 14 amount available, a reduction of this signal level into the amplifier will allow increased gain for the path in the opposite direction.

lG Attenuating the strong signal to that of the weaker signal is preferably 17 accomplished by using a fixed RF attenuator between bandpass filter 28 18 and circulator 30 (assuming that the F2 path is the stronger signal).

19 In addition to the two-hop microwave path as shown in FIG. 1, 21 the duplex repeater discussed herein may also be used in a multihop 22 system.

Due to the topography of a microwave path or for a variety 23 of other reasons, it is frequently impossible to lay out a system using 24 a single repeater.

The duplex repeater configuration shown in FIG. 2 may be used as any other active repeater in a tandem repeater mode.

26 When several duplex repeaters are used one after another, the duplex 27 repeater spacing should be limited to a distance such that the free 2 space attenuation between these duplex repeaters does not exceed the 29 effective repeater gain, which o course would include the antenna gains.

For example, in a 2 GHz system, using 6 foot parabolic dish ~1 32 - l5 ,,,.. ' ~. '. .' L-460 l()f~i~S8~ 1 antennas and an amplifier gain of 50 dB, the repeater spacings should 2 be limited to distances less than 2.3 miles. Repeater spacings in excess 3 of this amount will cause the path loss to exceed the repeater gain.

4 The end result will be that the fade margin will be reduced to the point that eventually the signal cannot be recovered from the thermal noise.

6 In other words, the radio receiver threshold will be reached and the 7 system will be inoperable. In a lO GHz tandem system, using the same 8 6 foot diameter parabolic dish antennas and an amplifier gain of 50 dB, 9 the duplex repeater spacing should be limited to less than 11 miles 0 between successive duplex repeaters of the type discussed herein.

FIG. 3 illustrates an application of the duplex repeater to a 12 frequency diversity microwave system. Frequencies F1 and F3 are used 13 to carry traffic in the west-to-east direction, and frequencies F2 and F4 14 are used to carry traffic in the east-to-west direction. As with the F and F2 carriers, the F3 and F4 frequencies have associated modulation 1~ bands ~F3 abd ~F4. As noted hereinabove, for simplicity, only the F3 17 and F4 designations will be used in the following discussion. It will be 18 understood that, where appropriate, the modulation bands ~F3 and ~F4 are 19 also to be included. This repeater configuration is basically composed of two complete duplex repeaters coupled together with circulators.

21 Functionally, this configuration operates much the same as the basic 22 duplex repeater.

Carriers Fl and F2 are channeled through circulators 23 45 and 47 and through amplifier 46 via the bandpass filters 43, 48, 44, 24 and 49. These bandpass filters operate in a similar fashion as those illustrated in FIG. 2. The diversity frequencies F3 and F4 have a somewhat 26 longer path through the repeater than F1 and F2. Frequency F3 enters 27 the repeater via the west antenna 40 and is conducted through circulator 28 29 31 - 16 32 L-460 1~ 95~37 1 41 to bandpass filter 43. Since it is outside the transmission band of 2 handpass filter 43, the signal is reflected back to circulator 41 and then 3 to circulator 42. Due to the rotation of circulator 42, the signal 4 .is conducted to circulator 52 and then finally to bandpass filter 53.

Since bandpass filter 53 has a center frequency at F3, the signal 6 is allowed to pass through to circulator 55 and on to microwave ampli 10 ¦ i r 56. Once ampliried, the signal is conducted through circulator 57 la lq 21 22 24 26 , 28 29 31 - 16A 32 ~j;

~ L-460 ~069587 1 and through bandpass filter 59 where it again travels through three more 2 circulators 60, 51, and 50 to eventually be retransmitted at the east 3 antenna 61.

The east-to-west path carrier F4 is similarly conducted 4 through the repeater and through amplifier 56 to be retransmitted at 5 the west antenna 40.

As an obvious modification, the circulators could 6 all be replaced with an electrical connection.

Putting the bandpass 7 filter directly together with no added isolation between them presents 8 no serious problem if the out-of-band rejection of each is sufficient 9 to keep the unwanted echo to an acceptable level.

ll It is to be understood that the above-described embodiments 12 are only illustrative of the application of the principles of the present 13 invention. Numerous other modifications and alternative arrangements 14 may be devised by those skilled in the art without departing from the spirit and scope of the invention.

el ¦ 2~ 26 29 31 32 - 17 :. . . . .