Method and device for optoelectronic distance measurement

The invention relates to a method for optoelectronic distance measurement in which

- an intensity-modulated main light beam emitted by a main light emitter is directed, on the one hand, to a distant measurement object, the distance of which is to be measured from an observation location, and the light scattered there reaches a main photo receiver via receiving optics and, on the other hand, a branched part of the main light beam at the same time is directed to a photoreference receiver over a first known reference distance;

- An intensity-modulated reference light beam emitted by a reference light emitter, on the one hand, reaches the photoreference receiver over a second known reference distance and, on the other hand, part of the reference light beam reaches the main receiver over a third reference distance, and in which

- The signals supplied by the main and reference receivers are fed to a comparative signal evaluation to obtain an error-compensated measurement signal. The invention also relates to a device for optoelectronic distance measurement

- two light transmitters, whose respective light beam is intensity-modulated, whereby the light beam of the first light transmitter, referred to as the "main emitter", can be directed on the one hand to a distant measurement object whose distance from an observation location is to be measured, and on the other hand a separated beam portion over a first reference distance to a comes from two photoreceivers, which is referred to as the "reference receiver" and where the light beam of the second light transmitter, referred to as the "reference emitter", reaches the reference receiver over a second reference distance and a beam portion separated therefrom reaches the second photoreceiver called the "main receiver" over a third reference distance, which is also connected to the The portion of the light25 beam from the main emitter that is controlled back by the measurement object is acted upon,

- In each case a signal mixer assigned to the reference receiver or the main receiver, which converts the receiver signal mixtures into an intermediate frequency range, as well as with

- An evaluation device for determining the measuring distance from the output signals of the 30 two mixers.

The invention also relates to a device for optoelectronic distance measurement

- Two light transmitters, whose respective light beam is intensity-modulated, the light beam 35 of the first light transmitter, referred to as the “main emitter”, on the one hand being alignable to a distant measurement object whose distance from an observation location is to be measured, and on the other hand a separated beam portion over a first reference distance get one of two photo receivers, which is referred to as the reference receiver, and wherein the light beam of the second light transmitter, referred to as the “reference emitter”, reaches the reference receiver via a second reference distance and a beam portion separated therefrom reaches the second photo receiver called the “main receiver” via a third reference distance Main emitter is applied.

- In each case a signal 45 mixer assigned to the reference receiver or the main receiver, which converts the receiver signal mixtures into an intermediate frequency range, as well as with

- An evaluation device for determining the measuring distance from the output signals of the two mixers.

so The optoelectronic measurement of distances up to 100 m with accuracies of a few millimeters has become important for numerous applications, especially in the construction industry and in plant engineering. The dynamics of such distance measuring systems should be as high as possible in order to be able to process both very weak and strong signals. This makes the use of defined target marks on the object, the distance of which is to be determined from an observation point, superfluous. The possibility of direct distance measurement

AT 413 451 Β certain surfaces, ie without the use of target marks, enables reduced production times and cost savings with a simultaneous reduction in production tolerances, particularly in the sectors mentioned.

Methods and devices for precise optoelectronic distance measurement are known. In most cases, as in the case of the invention, a preferably sinusoidal intensity-modulated beam from a light source, in particular a laser diode, is directed onto an object to be measured. The intensity-modulated light scattered back from the measurement object is detected by a photodiode. The distance to be measured results from the phase shift of the sinusoidally modulated light intensity scattered back from the object to be measured in relation to the emitted light intensity

Light intensity of the light source.

A main problem with high-precision distance or phase measurement systems of the type mentioned is the elimination of temperature and age-dependent parasitic phase changes of the

Light source, in particular in the laser diode transmitter and / or in the photodiode receiver. Various methods are known to deal with this problem.

One possibility described in the publication EP 0 701 702 B1 is the use of a mechanically switchable reference path. In this case, an intensity-modulated laser beam is first directed onto the measurement object during a first measurement and directly onto the photoreceiver via a tiltable mirror during a second reference distance measurement. By subtracting the measured phases, the temperature and aging effects of the components are to be eliminated. Since, however, with the alternating distance and reference distance measurement, very different optical reception powers are to be expected, the measurement error resulting from this25 is not eliminated. A major disadvantage of this concept is the use of moving mechanical components, which limits the reliability and service life of the entire measuring system.

Other known distance measuring devices of the type in question described in DE 196 43 287 A1 work with a reference photo receiver and a main photo receiver. A part of the intensity-modulated laser light is directed to the measurement object and from there to the main photo receiver and another part, separated from the laser light beam, is directed directly to the reference photo receiver. Since the reference photoreceiver is constantly illuminated during a measurement, no movable mechanical switch is required. With this concept, the phase response of the laser diode transmitter is eliminated, but not the time-varying phase response of the receiving components, which is generally different for the measurement and reference measurement branches. In addition, with distance measuring devices of this type, very different reception powers are to be expected in the two branches, which results in further phase errors.

In a further type of known optoelectronic distance measuring device (cf. US Pat. No. 4,403,857), which forms the starting point for the invention to be described here, two laser emitters and two photodiode receivers are used in order to eliminate the phase errors mentioned. As specified in the preambles of claims 1 to 12 in detail, in this device part of the intensity-modulated power of a main light emitter is directed directly onto the measurement object, from where it reaches a main photoreceiver as scattered light. Another part of this transmission power is sent to a reference photo receiver over a precisely known first reference distance. There is also a reference light emitter, the output power of which is also intensity-modulated and of which such a part reaches the main photoreceiver over a second reference distance, while another part is fed directly to the reference photoreceiver over a third reference distance.

The main and reference light emitters are activated one after the other via an electronic switch. This measuring principle does not require any mechanical switch. In addition

AT 413 451 Β, temperature and age-related phase changes are completely eliminated in both the transmitter and the receiver unit. However, since the measurements with the signals from the main and reference light emitters will result in considerable differences in the received power, the resulting phase errors are not eliminated even with the concept on which this known distance measuring device is based. Phase errors dependent on reception power are particularly noticeable in avalanche photodiodes (APD), which are preferred as the main receiver because of other advantages. At high amplifications, saturation effects gradually occur with increasing power, whereby the avalanche gain becomes dependent on the received power. This also results in a power-dependent phase shift when receiving high-frequency modulated optical radiation. In addition, the charge generated in the barrier layer of the APD varies with the received power, which influences the barrier layer width and thus also the barrier layer capacitance. The phase behavior of the low-pass filter formed by it changes with the junction capacitance. In the case of high APD amplification factors, a phase shift of usually greater than 5 ° can be caused by a received power variation of two orders of magnitude.

Another optoelectronic distance measuring device is known from EP 500 600 A. The device disclosed in this reference uses two different ones

IF frequencies, but only with one modulation frequency, which means that phase errors cannot be detected or eliminated with the desired sensitivity.

A much simpler and less precise device is known from EP 475 326 A, which has only one emitter and only one receiver.

DE 43 28 553 A discloses the alternating deactivation of a measuring receiver and a reference receiver by switching photodiodes. The respectively activated received signal is mixed with a local oscillator signal of its own frequency, which results in an intermediate frequency signal. Since ambiguities can occur at the high modulation frequencies, the respective activated received signal is also changed in its frequency, so that a further intermediate frequency signal is produced. In order to only have to tune the IF amplifier to one frequency, the local oscillator signal is tuned in such a way that the same intermediate frequency is always obtained. However, this cannot result in all phase errors due to the different frequencies of the

Signals in the IF stage are eliminated.

Another system, which is not concerned with eliminating the phase errors, is known from US Pat. No. 5,082,364 A. Two optical measurement signals are used, one for the coarse scale and one for the fine scale and an optical LO signal. This device manages without an optical reference signal for the accuracy it is aiming for.

The invention is therefore based on the object of specifying a distance measuring method and a device of the type defined in the introduction which works according to this method, with which a highly accurate distance measurement is possible and which of temperature, Phase errors dependent on aging and received power are completely independent. Mechanical or electronic changeover switches are to be dispensed with and the total measurement time for obtaining the measurement results is to be significantly shortened compared to the prior art.

According to the invention, the method of the type mentioned is characterized in that

- The light intensities of the main and the reference emitter are simultaneously intensity-modulated with different frequencies, wherein

- The mixed signals supplied by the main receiver and the reference receiver, each with a signal component with the intensity modulation frequency of the main emitter as well as one

AT 413 451 Β

Signal component with the intensity modulation frequency of the reference emitter are each converted into an intermediate frequency range that contains two frequency components, one frequency component being formed with the signal from the reference emitter and the other frequency component being formed with the signal from the main emitter,

- for comparative signal evaluation, the separation of the two occurring simultaneously

Phase information contained in intermediate frequency signals takes place due to the different frequencies in the intermediate frequency range and the different modulation frequencies for the intensity modulation of the main and reference light beam, and that

- To achieve a clear distance measurement and to improve the measurement result, time-sequential measurement processes are carried out, with the modulation frequencies for the intensity modulation being interchanged on the one hand and being changed equally on the other hand according to a fixed scheme.

In one embodiment it is provided that the output power of the main and / or the reference emitter is / are varied. This enables adaptation to the different dynamic requirements of the respective measuring process.

In a development it is provided that lasers are used as the main light emitter and for the reference light emitter. These light sources are ideally suited for measuring processes.

In a further embodiment it is provided that the fundamental wavelengths of the two lasers are selected to be different. This reduces the problems of crosstalk.

In one embodiment of this variant, it is provided that crosstalk between the light signal paths assigned to the main light beam or the reference light beam is reduced by optical filtering.

In a further development of the basic idea of ​​the invention it is provided that for comparative purposes

Signal evaluation the phases of the signal components of the signal mixtures in the intermediate frequency range by digital Fourier transformation with evaluation of the real and imaginary parts, which are determined in the frequency range Fourier-transformed signal mixtures at the respective intermediate frequencies. In this way, four phase relationships and the temperature-aging and received power-dependent phase shifts are obtained, which are caused in the main35 and in the reference receiver.

In one embodiment it is provided that the signal mixtures to be fed to the digital Fourier transformation are first subjected to a low-pass filtering. This eliminates signal components that result from the non-linear mixing process and reduces the noise level.

In one embodiment, it is provided that the intermediate frequencies and the sampling times of an analog-to-digital conversion preceding the Fourier transformation, i.e. the measurement window of the respective digital sampling, are selected so that an integer number of periods of both the signal component with the first intermediate frequency as well as the signal component with the second intermediate frequency lie in the measuring window of the digital sampling. This avoids the so-called leakage effect that occurs with the digital Fourier transformation when the frequency components are not in the frequency grid.

so In two further variants it is provided that the portion of the reference light beam or the reference and main light beam reaching the main receiver or the reference receiver is initially reflected or scattered and only scattered light portions reach the respective receiver. In this way it is avoided that only a small part of the respective beam cross-section is detected, which as a rule does not have the mean modulation phase, so that phase errors would occur.

AT 413 451 Β

The device according to the invention is marked

- By means of a device having a frequency synthesizer, through which the light beams emitted by the main emitter or the reference emitter simultaneously with respectively different ones

Frequencies are intensity modulable, and

a device present in the intensity modulation device for the time-sequential mutual interchanging of the frequencies of the intensity modulation signals arriving at the respective light emitters. With this device, the method according to the invention can be carried out reliably and precisely.

The advantageous designs and developments of the device according to the various variants of the method and advantageous structural details are listed in the subclaims.

The basic principles of the invention are as follows:

Similar to the distance measuring method described in US Pat. No. 4,403,857, two light transmitters, in particular lasers and two photodiode receivers, are also used in the subject matter of the invention. Deviating from this known method, however, according to the invention, the light, preferably sinusoidally intensity-modulated with a first modulation frequency fi, of the first light transmitter, referred to as the main emitter, is directed onto the surface of a measurement object. The light that is backscattered from there, likewise intensity-modulated, reaches the second photoreceiver, referred to as the main receiver, for example via receiving optics. At the same time, part of the modulated light from the main emitter is guided directly via a first reference path to the second photoreceiver called the reference receiver. The reference emitter is also preferably sinusoidally intensity-modulated with a second modulation frequency. Part of its modulated optical radiation reaches the main receiver via a second known reference path and in particular via a scattering medium, while another portion of its modulated optical radiation reaches the reference receiver via a third reference path.

Both receivers are acted upon by both emitter signals at the same time, so that, in contrast to the distance measuring method described in the above-mentioned US patent specification, no changeover switch is required and the measuring time is significantly shortened. The photoreceivers convert the detected modulated optical power into photocurrents, which are then preferably converted into voltages with transimpedance amplifiers.

The two signal voltages obtained in this way are then converted into suitable intermediate frequency ranges by assigned mixers using a locally generated frequency and then, after analog-to-digital conversion, a signal evaluation is carried out for error-free determination of the signal delay-related phase shift and thus the distance.

The invention and advantageous details are explained in more detail below with reference to the drawings.

Fig. 1 shows a basic, preferred structure of a distance measuring device based on the method according to the invention, and

FIG. 2 illustrates an intermediate frequency signal mixture obtained at the output of the amplifier 16 in FIG. 1 in the time domain (left) or in the frequency domain (right).

The basic arrangement of a distance measuring device working according to the method according to the invention contains a first light transmitter designated as main emitter 1, in particular a laser, and a second light transmitter designated as reference emitter 2, preferably a laser, and a first photoreceiver designated as reference receiver 4

AT 413 451 Β or a designated as the main receiver 5 second photo receiver. The main emitter 1 can be a comparatively powerful edge-emitting laser diode (EEL: Edge Emitting Laser Diode), the emission wavelength of which, as indicated in the drawing, for example! = 650 nm. As reference emitter 2, one is preferably on top of another

Wavelength, for example λ₂ = 850 nm radiating laser diode z. B. a VCSEL (Vertical Cavity Emitting Laser Diode) is used. The choice of different wavelengths for the main and reference emitters enables optical filtering, which is explained in more detail below, so that possible problems caused by optical crosstalk can be reduced. A PIN photodiode is preferably used as the reference receiver 4, while an avalanche photodiode is preferably provided as the main receiver 5.

Of decisive importance for the invention is the idea of ​​modulating the intensity of the radiation power emitted by the main emitter 1 or the reference emitter 2 at a specific measurement time at different frequencies, preferably sinusoidally. For example, the main emitter 1 is (initially) with the frequency T and the reference emitter 2 (initially) with the frequency f₂ intensity modulated. The two modulation frequencies T, f₂ are obtained via two frequency synthesizers 21 and 22 controlled by a common oscillator 20. The feeding of the modulation signals with the frequencies fi and f₂ in the (not shown) excitation circuit for the main emitter 1 or the

Reference emitter 2 takes place, for example, via high-frequency amplifiers 23 or 24.

The bundled main light beam 30 emitted by the main emitter 1 arrives, for example guided in a tube 40, first of all on a beam splitter 3, which divides the main light beam 30 into two parts, namely a main part 32, which is directed onto the measurement object, and a branched part 33 which reaches the reference receiver 4 via a first known reference distance Di, with the interposition of a diffuser 51 or scatterer. Through the diffuser 51 resp. the scatter ensures a homogeneous modulation phase distribution in the beam cross section in front of the reference receiver 4. A semitransparent mirror, a prism, a glass plate, an integrated optical beam splitter, a diffractive mirror can be used as the beam splitter 3

Element, e.g. B. a hologram or the like. Be provided. It should be emphasized, however, that the beams of the reference sections do not necessarily have to be collimated or deflected by beam splitters. Alternatively, for example, a volume scattering or direct illumination of the receiver z. B. be provided via a diffuser. The portion 36 of the measuring beam 32 that is backscattered from the distant measurement object reaches the main receiver 5 via a collecting optics 37. The generally collimated reference light beam 31 emitted by the reference emitter 2 is likewise divided into two parts by means of a beam splitter 6, a first part 34 over a second reference distance D₂ and reaches the reference receiver 4 via a diffuser 51 or scatter, while a second component 35 passes a third reference distance D₃, preferably via an optical filter 41 tuned to the wavelength of the reference emitter, first a scattering medium 11 and then acted upon the main receiver 5 as a scattered portion together with the portion 36 of the main light beam backscattered from the measurement object. In principle, any scattering material can be used as the scatter 11. Even the housing wall would be suitable. To control the scattered light output of the reference emitter light, however, the degree of scattering of the scatter material should be matched to the receiving system. Since a strong power attenuation of the reference emitter beam 35 is sought due to the generally weak measurement signals, a scatter material with a low degree of scatter is advantageous at least in the case of the main receiver 5, e.g. B. black paper, black velvet or the like.

The use of the scatter 11, which is very advantageous in connection with the invention, is based on the following observation: In the beam cross-section of laser diodes, the modulation phase is not homogeneous, that is, different points of the beam cross-section have different phases with respect to the modulated light intensity. In the case of homogeneous backscattering, a phase averaged over the beam cross-section is measured. In the case, however, that certain

Areas of the beam are masked from the measurement object surface, z. B. if a part

AT 413 451 B of the light spot hits black, absorbing areas and another part hits white, strongly backscattering areas of the test object surface, the mean phase changes and a measurement error occurs which depends on the unknown test object surface. This error cannot be compensated. In most cases, however, all points of the

Light spots scattered back almost to the same degree.

In the case of a reference emitter beam 35 guided directly onto the main receiver 5, only a small part of the beam cross section would be detected due to the small APD area. The phase of the beam measured in this way then generally does not represent the mean modulation phase, io In addition, the phase distribution in the beam cross-section is not constant over time and moreover depends on the temperature. The use of the scatter 11 ensures that signal components are detected from all points of the beam cross-section. Thus, a middle phase is measured, which has a more constant behavior than a point phase. Errors that arise through punctual measurement of the phase of the reference emitter beam 35 are reduced by the scatter 11.

To avoid such phase errors, it can also be advantageous to route the beams 33 or 34 via a scatter, e.g. B. the diffuser 51, before the modulated light is detected by the reference receiver 4.

Another task of the scatter 11 is a strong power attenuation of the reference emitter light. Because of the very weak measurement signals, the receiving system is very sensitive. The power attenuation prevents the APD in the main receiver 5 from being overdriven. In addition, this reduces the shot density, which increases with the detected light output.

It is important to adjust the scattering power of the reference emitter light to the system so that an optimized signal-to-noise ratio (SNR = signal-to-noise ratio) is achieved. If the scattered light output is too strong, large shot noise is to be expected, and thus a poor SNR in the case of weak measurement signals when determining the phase of the measurement signal. On the other hand, if the scattered light power is too low, the SNR when determining the phase of the reference emitter signal is poor. Since this phase is also included in the distance measurement, this can lead to a measurement error. It is therefore advisable to find a suitable compromise. For a distance measuring range of up to 100 m, it has been found in experiments that the detected scattered light power of the reference emitter beam 35 should be approximately as great as a power of the measuring beam 36 that is backscattered from a distance of 15 m. A backscatter coefficient of the test object surface of 0.5 and a homogeneous backscatter is assumed. With a power of the measuring beam 32 of 1 mW and a diameter of the receiving optics 37 of 50 mm, this corresponds to a detected scattered light power of the reference emitter beam of 0.7 nW.

The photocurrents supplied by the reference receiver 4 or the main receiver 5 are first converted into corresponding measurement voltages x via transimpedance amplifiers 9 and 10, respectively₂ or Xi converted.

In principle, impedance-controlled (50 Ohm) HF power amplifiers can also be used instead of the transimpedance amplifiers. As a rule, however, they have poorer noise properties and lower amplifications. For the reference branch, however, instead of the transimpedance amplifier 9, it can be sensible, for reasons of cost, to use an impedance-controlled RF power amplifier, with a strong signal being able to be achieved with a lower power consumption.

The signal x coming from the reference receiver 4₂ then acts on a measurement signal input of a first mixer 7, while the signal x₁ reaches the measurement signal input of a second mixer 8. The two mixers 7 and 8 are from the same local frequency9

AT 413 451 Β

Oscillator 20 (master oscillator) via frequency synthesizer 12 with a frequency f_L0 controlled, which is selected so that a signal mixture with the frequency components f_ZF1 and f_ZF2 is present. It is important here that the modulation frequencies of the main emitter 1 and the reference emitter 2 are different and the main receiver 5 and the reference receiver 4 both supply a signal mixture consisting of two signals with the frequencies and f₂ consists. This composite signal is generated in the two signal branches with the mixers 7 and 8, which are usually of the same construction, and the local oscillator signal of frequency f_L.o converted into the mentioned intermediate frequency range. Direct mixing via avalanche photodiodes (APD) is also conceivable in this context, with the signal of the local oscillator 12 being superimposed directly on the APD operating high voltage. The resulting modulation of the avalanche amplification causes this local oscillator signal to be mixed with the received signal, so that the APD output current comprises the two IF signal components f_ZF1 and f_ZF2 contains. There is thus no need for high-frequency amplifiers and high-frequency mixers. In this case, however, avalanche photodiodes are to be used both for the reference branch and for the measuring branch. The first intermediate frequency signal with the frequency f_ZFi is then derived from the received signal with the first modulation frequency fi and the intermediate frequency signal with the second intermediate frequency f_ZF2 from the received signal with the second modulation frequency f₂ derived. After a low-pass filtering 13 or 14 to eliminate the signal components with the sum frequencies + f_L.o and f₂ + f_LO and for noise signal reduction and amplification 15 or 16, the suitably amplified IF signals x₄ or x₃ scanned with analog-to-digital converters 17 and 18, respectively. The first and the second intermediate frequency f_ZF1 or f_ZF2 and the sampling times of the analog-to-digital converters 17, 18, that is to say the measurement window, are expediently chosen so that in each case an integral number of periods of the two intermediate frequency signals f_ZF1, f_ZF2 lie in the measuring window of the digital sampling. This avoids a so-called leakage effect that occurs with the digital, discrete Fourier transformation (DFT) when the frequency components are not in the frequency grid of the DFT; z. B. 40 kHz and 60 kHz for a 1 kHz frequency grid, that is, the distance between the discrete frequency values ​​is 1 kHz and the associated measuring window 1/1 kHz = 1 ms.

A digital Fourier transform 19 of the sampled signal mixture x₃ of the main receiver branch 42 and independently of the sampled signal mixture Χ4 of the reference receiver branch 43, e.g. B. for a certain measurement window as indicated above, provides the phases of the light signal components 33 and 36 of the main emitter 1, the phases of the light signal components 34 and 35 of the reference emitter 2 and temperature, aging and received power dependent

Phase shifts that are caused in the main and reference receiver branches. So four phase relationships are determined.

Since the phase can only be measured unambiguously in an interval from 0 to 2π, but the measuring distance produces much larger phase shifts in most cases, according to a significant improvement of the basic idea of ​​the invention - to achieve a clear distance measurement and to improve the measurement results - The modulation frequencies changed in a second measurement pass, ie the main emitter 1 is now with the frequency f₂ and the reference emitter 2 is sinusoidally intensity-modulated with the frequency. The measurement process described above is repeated with these new settings, so that for these new settings there are four further phases for the sampled signal mixtures in the main and reference receiver branches.

Since the signal of the main emitter 1 and the signal of the reference emitter 2 in the branch of the main receiver 5 or in the branch of the reference receiver 4 - as shown - pass through the same components, the formation of phase differences with respect to the signal phases measured in the main receiving branch 42 and in the reference receiving branch 43 the phase behavior of the respective receiver components is completely eliminated. In addition, the reception conditions are constant, since the main and reference emitter signals pass through the respective reception branches at the same time. The signals are separated - as also shown 55 - via the different modulation frequencies fi and f₂. Reception power dependent ο

AT 413 451 Β

Phase errors are thus also eliminated.

Through further formation of phase differences, the phase behavior of the main emitter 1 and the phase behavior of the reference emitter 2 are also eliminated, so that ultimately only a constant phase difference remains, which is due to the specified device-internal path differences of the reference distances D ^ D₂ and D₃ and by the measurement distance D (not shown in greater detail in FIG. 1)_O Both emitter signals arise outside the two laser emitters. The measuring distance D_O influences the phase of the modulated and detected by the main receiver 5 light component 32 of the main emitter 1, the reference distance Di the phase of the modulated and detected by the reference receiver light component 33 of the main emitter 1, the reference distance D.₂ the phase of the modulated light component 34 of the reference emitter 2 detected by the reference receiver and the reference distance D₃ the phase of the modulated and detected by the reference receiver light component 35 of the reference emitter 2. Since the device-internal transit times over the reference distances D ^ D₂ and D₃ outside the emitter 1, 2 are known and constant, the distance to be measured can be determined. It is therefore determined with the method according to the invention completely independently of the phase behavior of the transmitting and receiving units.

The phase difference determination by means of Fourier transformation is explained below. The basic form of the IF signals which are sampled by the analog-to-digital converters 17 and 18 can be seen in FIG. The composite signal is also shown there in the diagram on the right

Frequency range shown.

First, the main emitter 1 emits radiation with the modulation frequency ff and the reference emitter 2 radiation with the modulation frequency f₂. The mixer input signals Χϊ and x₂each have two sinusoidal signal components with the frequencies ff and f₂ the modulation.

It applies

Xi = x_{} A} cos (2TTfJ + <Phs (A) + «Mfi) - 2πή2Ο₀/ c) (1) + x, ₂ cos (2nf₂f + (p_R.s (f2) + <Ρηε (4) - 2π /₂Ο₃ / c) x₂ = x₂1 cos (2TTf, t + <Phs (A) + <Pre (/ i) - 2irfiDi / c) (2) + x_{2 2} cos (2TTf₂f + <PR_S.(f₂) + (p_{R £}(/ i) - 2nf₂D.₂ / c) with <Phs (ff) <Prs (f₂)

Φηε (fi) 45 φ_ΗΕ (fz) φβΕ (fl) φβΕ (f2) 50 □ o

Tuesday

D.₂ d₃

C.

Sum of the initial phase of the synthesizer (21) and the phase shift of the driver (23) and main emitter (1) at the frequency ff; temperature and age dependent;

Sum of the initial phase of the synthesizer (22) and the phase shift of the driver (24) and reference emitter (2) at the frequency f₂; temperature and age dependent;

Sum of the phase shift of the main receiver (5) and the transimpedance amplifier (10) at the frequency ff; temperature and age dependent; Sum of the phase shift of the main receiver (5) and the transimpedance amplifier (10) at the frequency f₂; temperature and age dependent; Sum of the phase shift of the reference receiver (4) and the transimpedance amplifier (9) at the frequency ff; temperature and age dependent;

Sum of the phase shift of the reference receiver (4) and the transimpedance amplifier (9) at the frequency f₂; temperature and age dependent; Measuring distance;

constant and known device-internal distance;

constant and known device-internal distance;

constant and known device-internal distance;

Speed ​​of light in air.

1

AT 413 451 B

By mixing (multiplying) the signals from Eq. (1) and (2) with the local oscillator signal of frequency f_L0 and subsequent low-pass filtering follows x₃ = X₃J C0S (2ttW + <p_HS(fi) + (PheW + <Pzf₃(/ zfi) - 2πή2Ο₀ / c) 5 + x_{3 2} cos (2nf_ZF2f + + (PheVi) + "iOzraifz") - 2π7₂Ο₃ / c) x₄ = x₄₁ cos (2ttW + <Ph <M) + <p_{R £}(fi) + <Pzf4 (/ zfi) - 2ττί ^ / c) + x_{2 2} cos (2TTf_ZF2t + <jO_RS(f₂) + <p_{R £}(f₂) + <PzF₄(fzF₂) - 2-nf₂D.₂ / c) io with the intermediate frequencies

7zfi = | fi - 4o | and fzF2⁼ I4 * ή_θ | (3) (4) (5) (6)

Due to the low-pass filtering, the signal components with the sum frequencies ή + f_LO and f₂ + f \ _o, which also arise from the non-linear mixing process, is eliminated. Also the

This reduces the noise level.

It means:

<PzF3 (fzFi) Sum of the initial phase of the synthesizer (12), initial sampling phase of the ADC (18) and phase shift of the low-pass filter (14) and amplifier (16) at frequency f_ZFi;

<PzF3 (fzF₂) Sum of the initial phase of the synthesizer (12), initial sampling phase of the ADC (18) and phase shift of the low-pass filter (14) and amplifier (16) at the frequency f_ZF2;

<PzF₄(fzFi) Sum of the initial phase of the synthesizer (12), initial sampling phase of the ADC (17) and phase shift of the low-pass filter (13) and amplifier (15) at frequency f_ZFi;

<PzF4 (fzF₂) Sum of the initial phase of the synthesizer (12), initial sampling phase of the ADC (17) and phase shift of the low-pass filter (13) and amplifier (15) at the frequency f_ZF2.

The IF signals x₃ and x₄ are now sampled synchronously with the A / D converters 17, 18. By discrete Fourier transformation of the sampled signals x₃ and x₄ in block 19 the signal components with the various intermediate frequencies f_ZF1 and f_ZF2 separate.

In FIG. 2, a composite signal x is shown on the left as an example₃ in the time domain with the frequency components f_ZFi = 40 kHz and f_ZF2 = 60 kHz shown. On the right is the amount of the Fourier-transformed signal mixture over the intermediate frequency f_ZF applied. Two sharp signal peaks can be seen at the respective intermediate frequencies. At other frequencies the values ​​of the spectrum are almost zero. The different heights of the peaks result from different amplitudes of the signal components in the time domain (0.7 V, 0.4 V).

The values ​​of the transformed signal mixture in the frequency domain are complex, ie it is set according to

X₃(fzF) = Re »)} + j lm {X₃(f_ZF)} from a real and an imaginary part. In Fig. 2 is the amount

X₃(4f) | = ^²{X3 (fZF)} + Re²{X3 (f_ZF)}

2

AT 413 451 Β shown. From the complex values ​​at the respective frequencies f_ZF1 and f_ZF2 can by means of the arctan function according to = arctan (lm {X₃(f_ZF1)} / Re {X₃(f_ZFi)}) and φ₂(ί₂) = arctan (lm {X₃(f_ZF2)} / Re {X₃(f_ZF2)}) io the phases of the separated signal components at the intermediate frequencies to be considered f_ZF1and f_ZF2 be determined. For x₃ from Eq. (3) they are ¢ 1 (/ 1)⁼ (PhsUi ') ⁺ Φηε (/ ι) ⁺ Φζρ₃(/ ζα) - 2ττΙ \ 2ΰο / C (7) and ¢ 2 (/ 2) ⁼ ¢^5(/2) ⁺ Ψη ^ 2) ⁺ Φζώ (/ ζώ) - 2π /₂Ο₃ / c. (8th)

For the phases of X4 from Eq. (4) follows correspondingly ¢ 3 (/ 1)⁼ <Phs (/ i) ^{+ +} Φζ / μ (/ ζρι) - 2lTf, 2D, / c (9) and ¢ 4 (/ 2) ⁼ ¢^(/2) ⁺ ¢/^/2) ⁺ Φζμ ^ ζώ) ​​- 2Trf₂D.₂1 c. (10)

It is particularly advantageous if, in the next step, the modulation frequencies f and f₂are interchanged, so that now the main emitter 1 radiation with the modulation frequency f₂and the reference emitter 2 emits radiation at the modulation frequency. As will be described further below, this measurement with interchanged modulation frequencies improves the clear distance measurement quite considerably. According to the sequence described above, the phases ¢ 1 (/ 2)⁼ φΗδ (/ 2) ⁺ ΦΗΕ ^) ⁺ ΦζΡ3 (/ ζΡ2) * 2lTf₂2Do / C (11) ¢ 2 (/ 1) = φκδ (/ ΐ) ⁺ Φηε (/ ι) ⁺ <PZFi (fzF \) - 2πήΟ₃ / c (12) ¢ 3 (/ 2) - <PhsU2) ⁺ Φκε (/ ^) ⁺ Φζρ4 (/ ζρ2) - 2πί₂20ι / c (13) ⁴⁰ ¢4(/1) ⁼ φρδ (/ ι) ⁺ Φ «ε (/ ι) ⁺ Φζμ (/ ζρι) 2πήΟ₂ / c (14) measured. By forming the differences, the gin follows. (7-14) ¢ 1 (/ 1) - ¢ 3 (/ 1)⁼ Φηε (/ ι) - Φκε (/ ι) ⁺ φζ / = 3 (/ ζπ) Φζρ4 (/ ζρι) 2tt / i2Do / c + 2πή20ι / c (15) ¢ 2 (/ 2) - ¢ 4 (/ 2) ⁼ Φηε ^) * Φ «ε (/ 2) ⁺ <PzFi (fzF2} - φζ & Μζρΐ) - 2πΓ₂Ο₃ / c + 2irf₂D.₂1 c (16) ¢ 1 (/ 2) ¢ 3 (/ 2) ⁼ ¢/-/5(/2) - ¢55(/2) ⁺ (PzF3 (fzF2) - (PzFttfzFz) - 2TTf₂2Do / c + 2nf₂2Di / c (17) ¢ 2 (/ 1) - ¢ 4 (/ 1) ⁼ ¢55(/1) - ¢55(/1) ⁺ <PzF3 (fzFi) - (PzfaUzfi) - 2πήΟ₃ / c + 2πήΟ₂ / c, (18) and by subtracting the gin. (15) and (18) or (16) and (17) finally result

Δφ (/ ι) = 2πί₁2Ο₁ / c - 2πήΟ₂ / c + 2ττήΟ₃ / c - 2rrf, 2D₀ / c + 2πη (19)

3

AT 413 451 Β

Δφ (/₂) = -2Trf₂2D ^ I c + 2nf₂D.₂ / c - 2πί₂Ο₃ / c + 2πί₂2Ο₀ / c - 2πη. (20)

Since the phase can only be measured unambiguously in an interval from 0 to 2π, but the measuring distance produces much larger phase shifts in most cases, the Gin. (19) and (20) introduced the integer number n of full periods which, in addition to the residual phase term, determines the total phase rotation. From the latter two gin. (19) and (20) can now determine the distance D to be measured_O and the number of periods n can be clearly determined, because the device-internal distances Di, D₂ and D₃ are constant and can be measured in advance. The two modulation frequencies f | and f₂ should be so close together that both gin. (19) and (20) result in the same number of periods n. This ambiguity of the measurement distance is the reason for the interchanging of the modulation frequencies according to the teaching of claim 2, because the additional measurement with interchanged frequencies provides the additional measurement from Eq. (19) independent Eq. (20). These two independent equations also provide D for large measuring distances_O unique values ​​for n and D_O.

Another advantage of the additional measurement with frequency swapping is that, like the Gin. (17) and (18) reveal - a complete elimination of the main and reference receiver phases (q>_HE (fi), Φηε ^), <PHE (fi), (Ψηε ^)) and the phases of the IF range. This elimination is done by subtracting the gin. (15) and (18) or the gin. (16) and (17).

The phase differences on the left side of the gin. (19) and (20) result from the phase measurement.

A smaller frequency difference - f₂ (e.g. a few 100 kHz, at ft = 900 MHz) is desirable on the one hand to clearly determine the integer number of periods n in the case of large measuring distances (e.g.> 100 m) (the same n in Gin. (19) and 20 )). On the other hand, greater noise-related measurement errors occur with small frequency differences, so that n may be incorrectly determined.

For even more precise measurement of large distances and simultaneous precise determination of the number of periods n, it is advantageous to use a second frequency pair f₃ and f₄ to use for intensity modulation in a further measurement run, which is, for example, 10 MHz from fi and f₂ differs. The above-mentioned procedure is carried out again with this frequency pair, in which case no interchanging is necessary, since the exact distance measurement with the frequency pair L and f₂ is carried out. Due to the larger frequency difference (e.g. f₃ - fi = 10 MHz) a possible measurement error is further reduced, and the integer number of periods n can now be determined unambiguously even with very large measurement distances. This measure of the uniform, small change in the intensity modulation frequencies specified in claim 18 enables an error-free determination of the distance and number of periods by using a further frequency pair, e.g. B. f₄ - 10 MHz, f₂ - Reach 10 MHz. In addition, by changing the measurement frequencies, optimal working points can be found, which lead to optimal signal-to-noise ratios. Due to the tolerances of bandpass filters, these optimal frequencies may differ slightly from device to device.

The method according to the invention and the distance measuring device based on it are characterized primarily by the following advantages:

- All phase errors are eliminated by measuring the reference and main emitter signals at the same time. This completely eliminates all temperature, aging and received power-dependent phase errors in both the transmitter and the receiver unit.

- The measurement accuracy is significantly improved.

AT 413 451 Β

- The reliability of the measurement results is significantly better.

- The measuring device is largely maintenance-free, since no mechanical switches or the like are required.

- By measuring the reference and main emitter signals at the same time, the measuring time is reduced and the measuring accuracy is increased.