Improved magnitude response and temporal alignment in phase vocoder based bandwidth extension for audio signals

By means of voice-operated phase coders [1-3] or other techniques for time or tone modification algorithms such as the Synchronized Overlap-Add algorithm (SOLA, for its acronym in English)) , the audio signals can for example be modified with respect to the proportion of prerecorded sound, preserving the original tone. In addition, these methods can be applied to carry out a transposition of the signal while maintaining the original duration of the prerecorded sound. The latter can be achieved by expanding the audio serial with an integer factor and subsequent adjustment of the prerecorded sound ratio of the expanded audio signal by applying the same factor. For a discrete time signal, the latter corresponds to a subsampling of the audio signal with expanded time over the expansion factor since the sampling rate remains unchanged.

Bandwidth extension methods based on oneo λ oo encoders ^{voice of} F^ase θθηιο [4-5] generate, depending on the total bandwidth required,

ZZÓ.J OO

MLDP _ORa variable amount of limited band subbands (patches) added to form a serial that exhibits the total bandwidth needed.

The temporary alignment of the individual patches that result from the application of the voice-operated phase encoder proves to be a challenge

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specific. In general, these patches have time delays of different durations. This is due to the fact that the synthesis window of the voice-operated phase encoder has a fixed jump size that depends on the expansion factor, and therefore each individual patch has a predefined duration delay. This results in a selective time delay in the frequency of the bandwidth extended sum serial. As this selective frequency delay affects the vertical coherence properties of the total serial, it has a negative impact on the transient response of the bandwidth extension method.

io

Another challenge arises when considering individual patches, where the lack of cross frequency coherence has a negative impact on the magnitude response of the voice-operated phase encoder.

is The objective of the present invention is to provide a concept for generating an extended audio bandwidth serial, which provides better audio quality.

The objective is achieved using an apparatus for generating an extended serial of audio bandwidth according to claim 1, a method for generating an extended serial of audio bandwidth according to claim 19 or a computer program for according to claim 20.

An apparatus for generating an extended serial of audio bandwidth from an input serial comprises a patch generator for generating one or more patch serial from the input serial. The patch generator is configured to perform a time expansion of sub-band signals from an analysis filter bank and comprises a phase adjuster to adjust phases of the sub-band signals using a phase correction that depends on a filter-channel bank.

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Another advantage of the present invention is that negative impacts on the magnitude responses normally introduced by voice-operated phase-encoding structures for bandwidth extensions or other structures for bandwidth extensions are avoided.

Another advantage of the present invention is that an optimized magnitude response of the individual patches is obtained, which, for example, are created by means of voice-operated phase encoders or structures such as voice-operated phase encoders. In another embodiment, the temporary alignment of the individual patches can also be treated, but the phase correction within a patch, that is between the sub-band signals processed using one and the same transposition factor, can be applied with or without valid time correction for all sub-band signals within a joint patch.

An embodiment of the present invention consists of a new method for optimizing the magnitude response and temporary alignment of the individual patches created by voice-operated phase encoders. This method basically consists of phase correction options for the subbands transposed in a complex modulated filter bank implementation and the introduction of additional time delays in the individual patches resulting from the voice-operated phase encoders with

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Different transposition factors. The length of time of the additional delay introduced to a specific patch depends on the transposition factor applied and can be determined theoretically. Alternatively, the delay is adjusted so that, when applying a serial input with Dirac pulse, the temporary center of gravity of the Dirac pulse transposed in each patch is aligned in the same temporary position in a spectrogram representation.

There are many methods that carry out transpositions of audio serials by an individual transposition factor such as the voice-operated phase io encoder. If several transposed serials must be combined, the time delays between the different serial outputs can be corrected. A correct vertical alignment between the patches is useful but not necessarily part of these algorithms. This is not harmful as long as they are not considered transient. The problem of the correct alignment of the different patches is not taken into account in the avant-garde literature.

The transposition of spectra by means of voice-operated phase encoders does not guarantee to preserve the vertical coherence of transients. In addition, the subsequent echoes emerge in the high frequency bands due to the overlay and summation method used in the voice-operated phase encoder as the different time delays of the individual patches that contribute to the sum serial. Therefore it is desirable to align the patches so that parametric postprocessing of the bandwidth extension can exploit a better vertical alignment between the patches. Therefore, the entire period of validity covered by the pre and post echo should be minimized.

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Xjy ρΛχ

A voice-operated phase encoder is typically implemented by means of a multiplicative phase modification of integer of subband samples in the domain of a pair of complex modulated filter banks of synthesis / analysis. This procedure does not automatically guarantee the adequate alignment of the phases of the serial contributions of the sub-band synthesizer output, and thus a magnitude response of the phase encoder operated by non-flat voice is obtained. This artifact results in an amplitude with time variation of a transposed slow sine sweep. In terms of audio quality for general audio, the drawback lies in the coloration of the output signal signal by means of modulation effects.

Preferred embodiments of the present invention are discussed below with respect to the accompanying drawings, where:

Fig. 1 illustrates a DIRAC pulse spectrogram with low pass filter;

Fig. 2 illustrates an impulse vanguard transposition spectrogram

Dirac with transposition factors 2, 3, and 4;

Fig. 3 illustrates an aligned transposition of time spectrum or Dirac pulse with transposition factors 2, 3, and 4;

Fig. 4 illustrates an aligned transposition of Dirac pulse time spectrometer with transposition factors 2, 3, and 4 and delay adjustment;

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Fig. 5 illustrates a time diagram of the slow sinusoidal scan transposition with poorly adjusted phase;

Fig. 6 illustrates a slow sinusoidal scan transposition with better phase correction;

Fig. 7 illustrates a slow sinusoidal scan transposition with another better phase correction;

io Fig. 8 illustrates a bandwidth extension system according to an embodiment;

Fig. 9 illustrates another embodiment of an example of a processing implementation for processing an individual sub-band series;

Fig. 10 illustrates an embodiment where non-linear sub-band processing and subsequent envelope adjustment within a sub-band domain is shown;

Fig. 11 illustrates another embodiment of the non-linear sub-band processing of Fig. 10;

Fig. 12 illustrates different implementations for selecting the phase correction that depends on the sub-band channel;

Fig. 13 illustrates an implementation of the phase adjuster;

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Fig. 14a illustrates implementation details for an analysis bank filter that allows independent phase correction of the transposition factor; Y

Fig. 14b illustrates implementation details for an analysis bank filter that requires a phase correction that depends on the transposition factor.

The present application provides different aspects of computing devices, methods or programs for processing audio signals in the context of bandwidth extension and in the context of other audio applications, which are not related to bandwidth extension. The features of the individual aspects of the following claims may be combined partially or totally, but they may also be used separately from each other, since the individual aspects already provide advantages with respect to perceptual quality, computational complexity and resources of processor / memory when implemented in a computer or micro processor system.

The embodiments employ a time alignment of the different harmonic patches created by the voice-operated phase encoders. Time alignment is done taking into account the center of gravity of a transposed Dirac pulse. The following Fig. 1 shows the spectrogram of a Dirac pulse with a low-pass filter that exhibits a limited bandwidth. This signal serves as the input signal for the transposition.

When transposing this Dirac Pulse by means of a voice-operated phase encoder, frequency selective delays are introduced into the sub

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Resulting bands The duration of these depends on the transposition factor used. Consequently, the transposition of a Dirac Impulse with the transposition factor 2, 3 and 4 is shown as an example in Fig. 2.

Selective frequency delays are compensated by inserting an additional individual time delay in each patch, that is, in each resulting patch. Thus, each individual subband is aligned so that the center of gravity of the Dirac pulse in each patch is in the same temporal position of the center of gravity of the Dirac pulse in the highest patch. The alignment is done taking into account the highest patch since it usually has the highest time delay. By applying the delay compensation of the invention, the center of gravity of the Dirac pulse is in the same temporal position for all patches within a spectrogram. Said representation of the resulting serials would be seen as in Fig. 3. Thus all the expansion of transient energy is minimized.

Finally, it is necessary to additionally compensate for the remaining time delay between the transposed high frequency regions and the original input serial. For this purpose, the input serial can be delayed as well as the centers of gravity of the transposed Dirac pulses, aligned to a certain time position in advance, corresponding to the temporal position of the Dirac pulse with band limit. Consequently, the spectrogram of the resulting serial is shown in Fig. 4.

To apply the described method, it is not taken into account if the voice-operated phase encoder as a fundamental component of the bandwidth extension method is performed in time domain or within a representation

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of filter bank as for example a mirror filter bank in quadrature polyphase (PQMF).

Using “SOLA” ​​techniques, the subjective audio quality of transients 5 is damaged by echo effects due to overlapping and summing while the vertical coherence criterion is met in the transients. Some possible, and slight deviations from the positions of the center of gravity in the individual patches from the actual center of gravity in the highest patch lie in the pre-mask or post-mask range, respectively.

The result of a poorly adjusted voice-operated phase encoder in terms of magnitude response is illustrated by the output signal signal of Fig. 5 which corresponds to a sinewave input of constant amplitude. As you can see, there are strong variations in amplitude and constant cancellations in the output signal signal. The output signal signal of a voice-operated phase encoder as soon as it is set is shown in Fig. 6.

An operation in a complex voice modulated phase coder based on a filter bank is the multiplicative phase modification of sub-band samples. A time domain sinusoidal input turns out to be accurate in the sub-band complex value signals so

Hp_n((ü) exp [z (iy ^ + ^)]

Where ω is the sinusoidal frequency, n is the sub-band index, k is the sub-band time interval index, q_TO is the time offset of

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Analysis filter bank, C is a complex constant, ν_η(ω) is the frequency response of the filter bank prototype filter, and θ_η It is a phase-end characteristic for the filter bank in question, defined by the requirement that ν_η(ώ) is transformed with real value. For typical QMF filter bank designs, they are supposed to be positive. In a phase modification a typical result has the form

Dv_n (ω) exp [i (T coq_sk + Τθ_η )]

Where T is the transposition order and q_s is the time offset of the analysis filter bank. Since the synthesis filter bank is typically chosen as a mirror image of the analysis filter bank, a suitable sinusoidal synthesis requires this last expression to correspond to the analysis subbands of a sinusoidal serial. Nonconformity of the above leads to amplitude modulations as shown in Fig. 5.

An embodiment of the present invention consists in using an additive post-modification of phase correction taking into account

This will indicate the unmodified subband signals until the desired cross subband phase evolution is obtained.

£&gt; v „(&lt;y) exp [ζ (Γ (oq_sk + Τθ_η)] i—&gt; Dv_n(ω) exp [ζ (Γcoq_sk + θ_η)].

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For the specific example of a complex modulated QMF filter bank with irregular stacking, one has ^θη

And the phase correction of the invention is given taking into account

The output signal signal of the voice operated phase encoder with phase set in accordance with this rule is shown in Fig. 7.

If the analysis / synthesis filter bank pair has more asymmetric distribution of phase rotation, there will be a phase correction ψ_η that, when the analysis subbands are crossed, and a minus sign before the synthesis, the situation returns to the previous symmetric case. In this case the phase correction of the invention must be adjusted taking into account

Δθ_η = (1-Τ) (θ_η-ψ_η)

An example of this is given by a pair of QMF filter bank band 64 used in the next MPEG standard (acronym in English for a group of experts on moving images) in Unified Speech and Audio Coding (USAC) in English) taking into account ^ψ· = Η 4)

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Where C is a real number and can have values ​​between 2 and 3.5. The particular values ​​are 321/128 or 385/128.

Therefore for that pair you can use

Δ0 „= ^ σ-ΐ) (« + ΐ).

Furthermore, in a special implementation of the previous situation, it is observed that a phase correction, independent of the transposition order T, could be incorporated in the pass of the analysis filter bank itself. Since a correction before the multiplication of the voice-operated phase encoder corresponds to T times the same correction after the phase multiplication, the following decomposition occurs as an advantage,

The modulation of the analysis filter bank is modified to add the phase yf Æ («+}) compared to the case for the standardized QMF filter bank pair, and the phase correction of the invention becomes equal to the second term only ,

The advantage of phase correction is that a flat response of magnitude is obtained for each order contribution of the voice-operated phase encoder to the output serial.

The processing of the invention is suitable for all audio applications that extend the audio signal bandwidth by applying the time expansion of the voice-operated phase encoder and subsampling or pre-recorded sound in an increased proportion respectively.

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Fig. 8 illustrates a bandwidth extension system according to an aspect of the present invention. The bandwidth extension system comprises a core decoder 80 that generates a core decoded serial. The core decoder 80 is connected to a patch generator 82 discussed below in detail. Patch generator 82 comprises all features of Fig. 8 but the core decoder 80, the low band connection 83 and the low band corrector 84 as the mixer 85. Specifically, the patch generator is configured to generate one or more patch signals from the audio input serial 86, where a patch serial has a different patch frequency than the patch frequency of another patch or the patch frequency of the audio input serial. Specifically, the patch generator comprises a first patch 87a, a second patch 87b and a third patch 87c, where in the embodiment of Fig. 8, each individual patch 87a, 87b, 87c comprises a subsampling apparatus 88a, 88b , 88c, a QMF analysis block 89a, 89b, 89c, a time expansion block 90a, 90b, 90c, and a patch channel patch block 91 a, 91 b, 91 c. The serial serial outputs of blocks 91a to 91c and the low band corrector 84 enter a mixer 85 that emits an extended serial bandwidth. This serial can be processed by processing modules such as the envelope correction module, the tonality correction module or other known modules of the bandwidth extension serial processing.

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Preferably, a patch correction is performed so that the patch generator 82 generates the serial or patch serials so that the time misalignment between the audio input serial and the serial or patch serial or time misalignment between different serials of patches, when compared with uncorrected, reduced or eliminated processing. In the embodiment of Fig. 8, this reduction or elimination of time misalignment is obtained by patch correctors 91 a to 91 c. Alternatively or additionally, patch generator 82 is configured to perform a phase correction that depends on the filter-channel bank with time expansion functionality. This is indicated by the phase correction input.

92a, 92b, 92c.

It should be noted that in the embodiment of Fig. 8 each QMF analysis block 89a emits a plurality of sub-band serials. The time expansion functionality must be performed for each individual subband serial. When, for example, the QMF analysis 89a emits 32 sub-band serials, there may be 32 time expanders 90a. However, an individual patch corrector for all serials with individual time expansion of this patch 87a is sufficient. As discussed later, Fig. 9 illustrates the processing in the time expander to be performed for each individual sub-band serial output serial by means of a QMF analysis bank as the QMF analysis banks 89a, 89b, 89c.

While a delay for the result of all time series processed using the same amount of time expansion is sufficient, an individual phase correction must be applied for each sub-band serial, since

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that the individual phase correction, although independent of the signal, depends on the number of channels of a sub-band filter bank or, otherwise, a sub-band index of a sub-band signal, where A sub-band index means the same as a number of channels in the context of this description.

Fig. 9 illustrates another embodiment of an example processing implementation for processing an individual subband signal. The individual subband serial has been subject to any type of decimation before or after its filtering by means of an analysis filter bank not shown in Fig. 9. Therefore, the length of time of the Individual sub-band signal is shorter than the length of time before forming the decimation. The individual sub-band signal enters a block extractor 1800, which may be identical to block extractor 201, but can be implemented differently. Block extractor 1800 in Fig. 9 operates using a sample feed rate / block exemplary called e. The sample / block advance value can be variable or fixedly determined and is illustrated in Fig. 9 with a float in the block extractor box 1800. In the output signal of block extractor 1800, there are a plurality of extracted blocks.

These blocks are very overlapping, since the sample / block feed value is significantly less than the block length of the block extractor. An example is that the block extractor extracts blocks from 12 samples. The first block comprises samples 0 to 11, the second block comprises samples 1 to 12, the third block comprises samples 2 to 13, etc. In this embodiment, the sample / block feed value is equal to 1, and there is an overlap of 11-folds.

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The individual blocks enter a partition generator in windows 1802 to partition windows of the blocks using a window function for each block. In addition, a phase calculator 1804 is provided, which calculates a phase for each block. Phase calculator 1804 can use the individual block before partitioning into windows or after partitioning into windows. Next, a pxke phase adjustment value is calculated and entered into a 1806 phase adjuster. The phase adjuster applies the adjustment value to each sample in the block. In addition, the k factor is equal to the bandwidth extension factor. When, for example, an extension of bandwidth by a factor 2 must be obtained, the phase p calculated for a block extracted by block extractor 1800 is multiplied by factor 2 and the adjustment value applied to each sample of the block in The 1806 phase adjuster is p multiplied by 2.

In one embodiment, the individual sub-band signal is a complex sub-band signal, and the phase of a block can be calculated in different ways. One way is to turn the sample in the middle or around the middle of the block and calculate the phase of this complex sample.

Although illustrated in Fig. 9 so that a phase adjuster operates after the partition generator in windows. These two blocks can also be exchanged, so that the phase adjustment is carried out on the blocks extracted by the block extractor and the subsequent window partition operation is performed. As both operations, that is, window partition and phase adjustment are multiplications of real value or complex value, these two operations can be summarized in a single operation using

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a complex multiplication factor, which, itself, is the product of a phase adjustment multiplication factor and partition factor in windows g.

The blocks with adjusted phase enter an overlap / sum block and amplitude correction 1808, where the blocks subject to partition in windows and with adjusted phase are superimposed-summed. It is important, however, that the advance value of the sample / block in block 1808 is different from the value used in block extractor 1800. Particularly, the sample / block advance value in block 1808 is io greater than the value e used in block 1800, to obtain a time expansion of the serial emission by a block 1808. Thus, the serial emission of sub-band processed by block 1808 has a length greater than the sub-band serial that enters block 1800. When the bandwidth extension of two must be obtained, the sample / block advance value is used, which is twice the corresponding value in block 1800. Thus, an expansion of time by a factor of two is obtained. When, however, other time expansion factors are needed, other sample advance values ​​/ may be used so that the output serial of block 1808 has a required length of time. In one embodiment, only one sample with index m =

0 It will be modified to have k (or T) times its phase. In this embodiment, this is not valid for the entire block. For other samples, the modification may be different as for example illustrated in Fig. 13 block 143.

With respect to the overlap, an amplitude correction is preferably performed to focus on the subject of different overlays in block 1800 and 1808. This amplitude correction could, however, also be introduced into the multiplication factor of the partition generator adjuster in

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windows / phase, but the amplitude correction can also be done after the overlay / processing.

In the example with a previous block length of 12 and a sample / block feed value in the block extractor of one, the sample / block feed value for overlay / sum block 1808 would be equal to two, when Performs a bandwidth extension by a factor of two. This would result in an overlap of five blocks. When a bandwidth extension must be made by a factor of three, the advance value of the sample / block used by block 1808 would be equal to three, and the overlap would fall to an overlap of three. When a four-fold bandwidth extension must be performed, the overlap / sum block 1808 would have to use a sample lead / block value of four, which would result in an overlap of more than two blocks.

In addition, a phase correction that depends on the filter bank channel enters the phase adjuster. Preferably, an individual phase correction operation is performed, where the phase correction value is a combination of the adjustment phase value that depends on the serial determined by the phase calculator and the phase correction independent of the serial (but depends on the number of channels of the filter bank).

While Fig. 8 illustrates an embodiment of bandwidth extension of an apparatus for generating an extended serial of audio bandwidth with greater bandwidth than the original serial of the core decoder, where several banks of QMF analysis filter 89a to 89c, another embodiment, where only one analysis filter bank is used is described

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with respect to Figs. 10 and 11. Furthermore, it should be defined with respect to Fig. 8 that QMF analysis 89d for the core encoder is only necessary when the mixer 85 comprises a synthesis filter bank. However, when mixing with the low band serial is done in time domain, Item 89d is not required.

Also, the mixer 85 may further comprise an envelope adjuster, or basically a high frequency reconstruction processor to process the serial input in the high frequency reconstructor taking into account the transmitted high frequency reconstruction parameters. These reconstruction parameters may include envelope adjustment parameters, noise aggregate parameters, reverse filtration parameters, missing harmonics parameters or other parameters. The use of these parameters and the parameters themselves and as are applied to make an adjustment of the envelope or, generally, a generation of the extended bandwidth signal is described in ISO / IEC 14496-3: 2005 (E), section 4.6.8 dedicated to the spectral band replication tool (SBR).

Alternatively, however, the mixer 85 may comprise a synthesis filter bank and subsequently to the synthesis filter bank an HFR processor to process the signal using the HFR parameters in the time domain instead of in the bank domain filter, where the HFR processor is placed before the synthesis filter bank.

In addition, in Fig. 8 the decimation functionality can also be applied after the QMF analysis. At the same time, the time expansion functionality illustrated in 92a to 92c, illustrated individually for each branch of

I

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Transposition can also be performed with a single operation for all three branches together.

Fig. 10 illustrates an apparatus for generating an extended audio bandwidth signal from a low band input serial 100 according to another embodiment. The apparatus comprises an analysis filter bank 101, non-line sub-band processors with respect to sub-band 102a, 102b, an envelope adjuster subsequently connected 103 or, in general, a reconstruction processor of high frequency operating in the high frequency reconstruction parameters as for example, entering the parameter line 104. The non-linear sub-band processors 102a, 102b of Fig. 10 or 11 are patch generators similar to block 82 in Fig. 8. The envelope adjuster, or in general, the high frequency reconstruction processor processes individual sub-band signals for each sub-band channel and enters sub signals. -processed band for each sub-band channel in a synthesis filter bank 105. The synthesis filter bank 105 receives, in its input channel of the lower channel s, a sub-band representation of the signal of the low-band core decoder generated, for example, by the analysis bank QMF 89d illustrated in the Fig. 8. Depending on the implementation, the low band may also derive from the output signals of the analysis filter bank 101 in Fig. 10. Transposed subband signals are supplied in higher filter bank channels of the synthesis filter bank for high frequency reconstruction.

Filter bank 105 finally emits a serial output signal from the transponder comprising bandwidth extensions by transposition factors 2, 3, and 4, and the serial output signal by block 105 is no longer

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limited to the bandwidth at the cross frequency, that is, at the higher frequency of the core signal corresponding to the lower frequency of signal components generated by SBR or HFR.

In the embodiment of Fig. 10, the analysis filter bank performs a double sampling overlay and has a certain analysis subband separation 106. The synthesis filter bank 105 has a subband separation of Synthesis 107 which, in this embodiment, duplicates the size of the analysis sub-band separation resulting in a transposition contribution as discussed below in the context of Fig. 11.

Fig. 11 illustrates a detailed implementation of a preferred embodiment of a non-linear sub-band processor 2a in Fig. 10. The circuit illustrated in Fig. 11 receives as input a single sub-band signal 108, which is processed in three “branches”: The upper branch 110a is for a transposition by a transposition factor of 2. The branch in the middle of Fig. 11 indicated as 110b is for a transposition by a transposition factor of 3, and the lower branch in Fig. eleven it is for a transposition by a transposition factor of 4 and is indicated by the reference number 110c. However, the actual transposition obtained by each processing element in Fig. 11 is only 1 (i.e. no transposition) for branch 110a. The actual transposition obtained by the processing element illustrated in Fig. 11 for the middle branch 110b is equal to 1.5 and the actual transposition for the lower branch 110c is equal to 2. This is indicated by the numbers in parentheses to the left of Fig. 11, where transposition factors T are indicated. Transpositions of 1.5 and 2 represent a first transposition contribution obtained by having decimation operations on branches 110b , 110c and a time expansion by the superposition processor

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-sum. The second contribution, that is to say twice the transposition, is obtained by the synthesis filter bank 105, which has a synthesis sub-band separation 107 which is twice the sub-band separation of the analysis filter bank. Therefore, as the synthesis filter bank has a two-fold separation of the synthesis subband, there is no decimation functionality in the branch 110a.

Branch 11 Ob, however, has a decimation functionality to obtain a transposition by 1.5. Because the synthesis filter bank io has twice the physical sub-band separation of the analysis filter bank, a transposition factor of 3 is obtained as indicated in Fig. 11 to the left of the block extractor for the second branch 110b.

Similarly, a third branch has a functionality of 15 decimation corresponding to a transposition factor of 2, and the final contribution of the different sub-band separations in the analysis filter bank and synthesis filter bank finally corresponds to the transposition factor of 4 of the third branch 110c.

Particularly each branch has a block extractor 120a, 120b, 120c and each block extractor can be similar to block extractor 1800 of Fig. 9. In addition, each branch has a phase calculator 122a, 122b and 122c, and the calculator phase may be similar to the phase calculator 1804 of Fig. 9. In addition, each branch has a phase adjuster 124a, 124b, 124c and the phase adjuster may be similar to the phase adjuster 1806 of Fig. 9. In addition, each branch has a window partition generator 126a, 126b, 126c, where each window partition generator can be similar to the partition generator in

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windows 1802 of Fig. 9. However, partition generators in windows 126a, 126b, 126c can be configured to apply a rectangular window together with a "zero padding" (zero padding). The transposition or patch signal from each branch 110a, 110b, 110c, in the embodiment of Fig. 11, enters the adder 128, which adds the contribution of each branch to the current sub-band signal to finally obtain the so-called transposition blocks in the output signal of adder 128. Then, an overlap-sum processing is performed in the overlapping device 130 and the overlapping device 130 may be similar to the overlapping / adding block 1808 of Fig. 9. The superimposition device applies an overlap-sum advance value of 2 e, where e is the superposition-advance value or "displacement value" of block extractors 120a, 120b, 120c, and the superposition-adder device 130 emits the transposed serial which is in the embodiment of Fig. 11, an individual sub-band output signal for channel k, that is, for the currently observed subband channel. The processing illustrated in Fig. eleven it is performed for each sub-band of analysis for a certain group of sub-bands of analysis and, as illustrated in Fig. 10, the transposed sub-band serials enter the synthesis filter bank 105 after being processed by block 103 to finally obtain the transposed output signal signal illustrated in Fig. 10 in the output signal of block 105.

In one embodiment, block extractor 120a of the first branch of transponder 110a extracts 10 sub-band samples and then subsequent conversion of these 10 QMF samples to polar coordinates. The output signal is defined as in Fig. 13, block 143, discussed below. This output signal, generated by the 124a phase adjuster, is

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Forwarded to the partition generator in windows 126a, which extends the serial output by zeros for the first and last value of the block, where this operation is equivalent to (synthesis) the partition into windows with a rectangular window of length 10. The extractor of block 120a in branch 110a does not perform a decimation. Therefore, the samples extracted by the block extractor are mapped into a block extracted in the same sample separation as they were extracted.

However, this is different for branches 110b and 110c. The block extractor 120b preferably extracts a block of 8 sub-band samples and distributes these 8 sub-band samples in the extracted block in a different sub-band sample separation. Subband sample entries without integer for the extracted block are obtained by interpolation, and therefore the QMF samples obtained together with the interpolated samples are converted into polar coordinates and processed by the phase adjuster 124b to obtain an expression similar to the expression in block 143 of Fig. 13. Then, again, the partition into windows is performed in a partition generator in windows 126b is performed to extend the serial output of the block by the phase adjuster 124b by means of zeros for the first two samples and the last two samples, whose operation is equivalent to a (synthesis) partition in windows with a rectangular window of length 8.

The block extractor 120c is configured to extract a block with time extension of 6 sub-band samples and performs a decimation of a decimation factor 2, converts the QMF samples into polar coordinates and again performs an operation on the phase adjuster 124b to obtain an expression similar to what is included in block 143 of the

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Fig. 13, and the output signal is again extended by means of zeros, however now for the first three sub-band samples and for the last three sub-band samples. This operation is equivalent to (synthesis) the partition into windows with a rectangular window of length with a rectangular window of length 6.

The serial transposition outputs of each branch are added to form the QMF output serial combined by the adder 128, and the combined QMF serial outputs finally overlap using overlay -sum in the io block 130, where the feed value or overlap offset -sum is twice the offset value of block extractors 120a, 120b,

120c as analyzed before.

Next, different embodiments are analyzed to determine preferred phase corrections in the context of Fig. 12. In an embodiment indicated in 151, there is a symmetrical situation of pair of analysis / synthesis filter banks and the ZO phase correction_n it has a first term 151a depending on the transposition factor T and a second term

151b which depends on the channel number no, in the notation in Fig. 11, k.

In this embodiment, the phase adjuster is configured to apply a phase correction using the value J0_n indicated as Q (k) in Fig. 11, which not only depends on the channel of the filter bank according to the term 151b, but can also depend on the transposition factor T indicated with the term

151a. It is important, however, that the phase correction does not depend on the actual sub-band serial. This dependence is due to the phase calculator for transposition of the voice-operated phase encoder as analyzed in

<img file="AR080475A1_D0026.tif" />

context with blocks 122a, 122b, 122b, but the phase correction or "complex output signal gain value Q (k)" is independent of the subband signal.

In another embodiment, indicated as horn 152 in Fig. 12, an asymmetric distribution of phase turns occurs. The phase turns are used to move the input samples of a block of the analysis filter bank along the time axis and to shift the output serial values ​​of a synthesis filter bank along the axis of weather. The values ​​of the phase turns are indicated io corno Ψ_η. The phase corrections used in a case with asymmetric distribution of phase turns are indicated as Ώθη, and again there is a term that depends on a transposition factor 152a and sub-band channel 152b.

Another preferred embodiment of the present invention indicated as

153 it has the advantage over embodiments 151 and 152 that the phase correction term ΞΙθη or Q (k) illustrated in Fig. 11 only depends on the sub-band channel, but no longer depends on the transposition factor. This advantageous situation can be obtained by specific application of phase turns to the analysis filter bank to cancel the term dependent on the transposition of the phase correction. In a certain embodiment for a specific implementation of a filter bank this value is equal to Z0_nindicated in Fig. 12. However, for other designs of filter banks the value of ΞΙθη may vary. Fig. 12 illustrates a constant factor of 385/128, but this factor can vary from 2 to 4 depending on the situation. Likewise, the use of other values ​​from 385/128 is described, and the deviation from this value for the specific design of the filter bank, for which this value is optimal will only be considered as

<img file="AR080475A1_D0027.tif" />

The result is a slight dependence on the transposition factor, which can be ignored to a certain extent.

Fig. 13 illustrates a sequence of steps performed by each branch of transponder 110a, 11 Ob, 11 Oc. In a step 140, a sample m for an extracted block is determined by extraction of pure sample as in block 120a, or by decimating as in blocks 120b, 120c and probably also by interpolation as indicated in the context of block 120b . Then, in step 141, the magnitude r and phase Φ of each sample is calculated. In block 142, the phase calculator 122a, 122b, 122c in Fig. 11, calculates a certain magnitude and a certain phase for the block. In the preferred embodiment, the magnitude and phase of the value in the middle of the extracted and potentially declined and interpolated block is calculated as the phase value for the block and as the amplitude value of the block. However, other samples of the block can be turned to determine the phase and magnitude for each block. Alternatively, even an average magnitude or average phase of each block determined by adding the magnitudes and phases of all samples in a block and dividing the resulting values ​​by the number of samples in a block can be used as phase and magnitude of the block. In the embodiment of the

Fig. 13, however, it is preferable to use the magnitude and phase of the sample in the middle of the block at zero index as magnitude and phase for the block. A sample adjusted by the phase adjuster 124a, 124b, 124c is then calculated using the Ω phase correction of the invention (being a complex number) as a first term, using a magnitude modification as a second term (of which however it is possible exempt), using the phase value that depends on the signal by blocks 122a, 122b, 122c corresponding to (T-1)

<img file="AR080475A1_D0028.tif" />

Φ (0) as a third term, and using the reai phase of the sample currently considered Φ (ιτι) as a fourth term as indicated by block 143.

Fig. 14a and Fig. 14b indicate two different modulation functionalities for analysis filter banks for the embodiments of Fig. 12. Fig. 14a illustrates a modulation for an analysis filter bank that requires correction of phase that depends on the transposition factor. This modulation of the filter bank corresponds to the embodiment 153 in Fig. 12.

io

An alternative embodiment is illustrated in Fig. 14b corresponding to embodiment 152, where a phase correction that depends on the transposition factor is applied due to an asymmetric distribution of the phase turns. In particular, Fig. 14b illustrates the specific modulation of the analysis filter bank corresponding to the complex SBR filter bank in ISO / IEC 14496-3, section 4.6.18.4.2, incorporated herein by reference.

When comparing Figs. 14a and 14b it turns out that the amount of phase turns to calculate the cosine and sine values ​​is different in the last two terms of Fig. 14b and last term of Fig. 14a.

One embodiment comprises an apparatus for generating an extended audio bandwidth signal from an input signal, comprising: a patch generator for generating one or more patch signals from the audio input serial, where a signal of patch has a central patch frequency different from the central patch frequency of a

<img file="AR080475A1_D0029.tif" />

different patch or from a central frequency of the audio input serial, where the patch generator is configured to generate the serial or patch serial to reduce or eliminate a time misalignment between the audio input serial and the serial or serial patch or misalignment of time between different patch serials, or where the patch generator is configured to perform a phase correction that depends on the filter-channel bank within a time expansion functionality.

In another embodiment, the patch generator comprises a plurality of patches, each patch has a decimation functionality, a time expansion functionality, and a patch corrector to apply a time correction to the serial patch to reduce or eliminate misalignment of time.

In another embodiment, the patch generator is configured so that the time delay is stored and selected so that when a serial of the impulse type is processed, the centers of gravity of the serials subject to patch obtained by the processing are aligned between each other in time.

In another embodiment, the time delays applied by the patch generator to reduce or eliminate misalignment are stored in a fixed manner and independent of the processed serial.

In another embodiment, the time expander comprises a block extractor using an extraction advance value, a window partition generator / phase adjuster, and an overlay-adder device with a

<img file="AR080475A1_D0030.tif" />

overlap advance value - add different to the extraction advance value.

In another embodiment, a time delay applied to reduce or eliminate misalignment depends on the extraction advance value, overlap advance value -sum or both.

In another embodiment, the time expander comprises the block extractor, the window partition generator / phase adjuster, and the io overlay-adder device for at least two different channels with different channel numbers of a bank of analysis filter, where the window partition generator / phase adjuster for each of the at least two channels is configured to apply a phase adjustment for each channel, The phase setting depends on the channel number.

In another embodiment, where the phase adjuster is configured to apply a phase adjustment in sampling values ​​of a block of sampling values, the phase adjustment is a combination of a phase value that depends on an amount of expansion. of time and of the reai phase of the block, and phase value independent of the serial that depends on the channel number.

Although some aspects have been described in the context of an apparatus, it turns out that these aspects also represent a description of the corresponding method, where a block or device corresponds to a step of the method or feature of the step of the method. Similarly, the aspects described in the context of the passage of a method further represent the

<img file="AR080475A1_D0031.tif" />

description of a corresponding block or item or feature of a corresponding device.

The encoded audio signal of the invention may be stored in a digital storage medium or transmitted in a transmission medium such as wireless transmission medium or via Internet wiring.

Depending on certain implementation requirements, the embodiments of the invention may be implemented in hardware or software, or the implementation may be performed using a digital storage medium for example a floppy disk, DVD, CD, ROM, PROM, EPROM, EEPROM or FLASH memory, with electronically readable control signals stored in them, that cooperate (or can cooperate) with a programmable computing system so that the respective method is applied.

Some embodiments according to the invention comprise data carriers with electronically readable control signals, which can cooperate with a programmable computing system so that one of the methods described herein is applied.

Generally, the embodiments of the present invention may be implemented as a computer program product with a program code, the program code applies one of the methods when the computer program product operates on a computer. The program code may for example be stored in a machine-readable carrier.

<img file="AR080475A1_D0032.tif" />

Other embodiments include the computer program to apply one of the methods herein, stored in a machine-readable carrier.

In other words, an embodiment of the method of invention, therefore, consists of a computer program with a program code to apply one of the methods of the present when the computer program product operates on a computer.

io Another embodiment of the method of invention therefore consists of a data carrier (or digital storage medium or computer readable medium) comprising, the computer program for applying one of the methods of the present engraving on the same.

Another embodiment of the method of the invention therefore consists of a data stream or signal sequence representing the computer program to apply one of the methods of the present. The data stream or signal sequence may, for example, be configured to be transferred via data communication connection, for example via the Internet.

Another embodiment comprises a processing means, for example a computer, or logical programmable device , configured for or adapted to apply one of the methods herein.

Another embodiment comprises a computer with a computer program installed in it to apply one of the methods herein.

<img file="AR080475A1_D0033.tif" />

In some embodiments, a programmable logic device (for example a Field of Programmable Door Matrices) may be used to apply some or all of the functionalities of the methods herein. In some embodiments, a Programmable Door Matrix Field may cooperate with a microprocessor to apply one of the methods herein. Generally, the methods are preferably applied by a hardware apparatus.

The embodiments described above are only illustrative for the principles of the present invention. It is understood that modifications and variations of the provisions and details herein will be apparent to other experts in the art. Therefore, it is intended to be limited only to the scope of the patent claims and not to the specific details presented by way of description and explanation of the embodiments of the present.

Literature:

[1] JL Flanagan and RM Golden, Voice-Operated Phase Encoder, The

Bell System Technical Journal, November 1966, pp 1394-1509 [2] US Patent 6549884 Laroche, J. &amp; Dolson, M .: Voice-coding phase / tone-displacement [3] J. Laroche and M. Dolson, New Voice-operated Phase Encoder Techniques for Tone-Shift, Harmonization and other Exotic Effects,

Proc. IEEE Workshop of Serial Proc. ao Serial Proc. A Audio et Acùs., New Paltz, NY 1999.

<img file="AR080475A1_D0034.tif" />

[4] Frederik Nagel, Sascha Disch, Harmonic Bandwidth Extension Method for Audio Encoders, ICASSP, Taipei, Taiwan, April 2009 [5] Frederik Nagel., Sascha Disch and Nikolaus Rettelbach, Width Extension Method Voice-operated phase encoder band with new transient handling for audio encoders, AES Convention 126, Munich, Germany May 7-10, 2009